Active quadrature demodulation for subsampling / circular ranging optical coherence tomography imaging

Abstract

A method comprises: using an electromagnetic radiation source to scan a sample within a time period, the time period comprising a first time period and a second time period, the sample portion of the electromagnetic radiation source being directed to the sample in a sample arm of an optical interferometer system, and the reference portion of the electromagnetic radiation source being directed to the reference arm of the optical interferometer system; using a phase modulator to apply a phase shift comprising a first phase shift and a second phase shift to at least one of the reference portion or the sample portion of the electromagnetic radiation source, the first phase shift being applied during the first time period and the second phase shift being applied during the second time period, the second phase shift being 90 degrees out of phase with the first phase shift; collecting in-phase data based on a first interference between a first backscattered electromagnetic radiation during the first time period and at least one of the reference portion or the sample portion subjected to the first phase shift; collecting orthogonal data based on a second interference between a second backscattered electromagnetic radiation during the second time period and at least one of the reference portion or the sample portion subjected to the second phase shift; and determining a composite interference signal based on the in-phase data and the orthogonal data.

Description

[0001] CROSS-REFERENCE TO RELATED APPLICATIONS

[0002] This application claims the benefit of and priority to U.S. Provisional Application No. 62 / 711,728, filed on July 30, 2018, and U.S. Provisional Application No. 62 / 799,582, filed on January 31, 2019, the entire contents of which are incorporated herein by reference for all purposes.

[0003] A note about federally funded research

[0004] This invention was made with government support under Grant P41EB015903 from the National Institutes of Health (NIH). The government has certain rights in this invention. Background Art

[0005] Acquisition in the frequency domain has increased the sensitivity and imaging speed of optical coherence tomography techniques by several orders of magnitude, but imposes constraints on the acquisition bandwidth, which limits the imaging range, the optical bandwidth, or both. Coherence circular ranging (CR) allows imaging at high speed and at long distances and high axial resolution simultaneously. By using an optical frequency comb, the depth of the interferometric signal is spatially folded, enabling long imaging distances at greatly reduced RF bandwidth. CR combines the high imaging speed and sensitivity of Fourier domain techniques with the long imaging distances known from time-domain OCT. Unfortunately, the use of frequency combs produces RF errors that can cause artifacts when imaging at several times the primary measurement range of the frequency comb free spectral range (FSR). In order to access the full depth range, it is beneficial to implement an efficient, high-speed, and stable method to acquire resolved (also called composite) interferometric signals. In some previous reports of CR, this has been achieved by passive polarization-based optical orthogonal demodulation. Although this approach has reduced artifacts by more than 50 dB, it uses relatively complex optical circuits and has achieved optimal performance when paired with frequent recalibration routines. More robust methods for performing demodulation may simplify the construction and operation of these CR systems. Summary of the invention

[0006] Accordingly, in various embodiments, the present invention discloses an active phase modulation method that uses a phase modulator (PM) to provide stable I / Q demodulation of the detected signal, so that a composite signal including real in-phase (I) and imaginary quadrature (Q) signal components at a high laser repetition rate of up to or exceeding 1 MHz can be obtained, which makes it feasible to use CR for clinical and industrial applications.

[0007] In one embodiment of the present invention, a method is provided, which includes: using an electromagnetic radiation source to scan a sample within a time period, the time period including a first time period and a second time period, the sample portion of the electromagnetic radiation source is guided to the sample in the sample arm of an optical interference system, and the reference portion of the electromagnetic radiation source is guided to the reference arm of the optical interference system; using a phase modulator to apply a phase shift including a first phase shift and a second phase shift to at least one of the reference portion or the sample portion of the electromagnetic radiation source, the first phase shift is applied during the first time period and the second phase shift is applied during the second time period, and the second phase shift is 90 degrees different from the first phase shift; based on a first interference between a first backscattered electromagnetic radiation during the first time period and at least one of the reference portion or the sample portion subjected to the first phase shift, collecting in-phase data; based on a second interference between a second backscattered electromagnetic radiation during the second time period and at least one of the reference portion or the sample portion subjected to the second phase shift, collecting orthogonal data; and determining a composite interference signal based on the in-phase data and the orthogonal data.

[0008] In some embodiments, the first phase shift is 0 degrees during the first time period and the second phase shift is 90 degrees during the second time period.

[0009] In some embodiments, applying a phase shift further includes: applying a phase shift to a reference portion of an electromagnetic radiation source, collecting in-phase data further includes: collecting in-phase data based on a first interference between first backscattered electromagnetic radiation during a first time period and the reference portion subjected to the first phase shift, collecting orthogonal data further includes: collecting orthogonal data based on a second interference between second backscattered electromagnetic radiation during a second time period and the reference portion subjected to the second phase shift, and determining a composite interference signal further includes: determining the composite interference signal based on the in-phase data and the orthogonal data.

[0010] In some embodiments, an electromagnetic radiation source emits a plurality of electromagnetic radiation pulses including a first A-line and a second A-line, the first A-line including a first subset of the plurality of electromagnetic radiation pulses emitted during a first time period and the second A-line including a second subset of the plurality of electromagnetic radiation pulses emitted during a second time period, and scanning the sample further includes: scanning the sample using the first A-line during the first time period and scanning the sample using the second A-line during the second time period.

[0011] In some embodiments, a first subset of the plurality of electromagnetic radiation pulses corresponds to a particular wave number sequence and a second subset of the plurality of electromagnetic radiation pulses corresponds to a particular wave number sequence.

[0012] In some embodiments, a first wave number in a particular wave number sequence is different from a second wave number in the particular wave number sequence.

[0013] In some embodiments, a first A-line is scanned at a first location in the sample, and a second A-line is scanned at a second location in the sample that is different from the first location.

[0014] In some embodiments, the electromagnetic radiation source emits a third A-line, which includes a third subset of multiple electromagnetic radiation pulses emitted during a third time period and corresponding to a specific wave number sequence, scanning the sample further includes: scanning the sample using the third A-line during the third time period; applying a phase shift further includes: applying a 0 degree phase shift to a reference portion of the electromagnetic radiation source during the third time period; the in-phase data includes first in-phase data, and the orthogonal data includes second orthogonal data, collecting the in-phase data further includes: collecting the third in-phase data based on the third interference between the third backscattered electromagnetic radiation during the third time period and the reference portion shifted by 0 degrees; and determining the composite interference signal further includes: determining the second in-phase data based on interpolation between the first in-phase data and the third in-phase data, and determining the composite interference signal based on the second in-phase data and the second orthogonal data.

[0015] In some embodiments, the electromagnetic radiation source emits a beam having a beam diameter, wherein the distance between the first location and the second location is one quarter of the beam diameter or less.

[0016] In some embodiments, the electromagnetic radiation source emits a plurality of electromagnetic radiation pulses, and each of the plurality of electromagnetic radiation pulses is divided into two time periods corresponding to a first time period and a second time period.

[0017] In some embodiments, the method further includes: providing a modified electromagnetic radiation source based on combining a plurality of electromagnetic radiation pulses with delayed copies of a plurality of electromagnetic radiation pulses; and scanning the sample further includes: scanning the sample using the modified electromagnetic radiation source, wherein the modified electromagnetic radiation source alternately emits a plurality of electromagnetic pulses, each occurring during a first time period, and a corresponding plurality of delayed pulses, each occurring during a second time period.

[0018] In some embodiments, providing a modified electromagnetic radiation source includes providing a modified electromagnetic radiation source based on combining a plurality of electromagnetic radiation pulses with delayed copies of the plurality of electromagnetic radiation pulses, wherein the delayed copies of the plurality of electromagnetic radiation pulses are delayed by a time that is shorter than a time of one of the plurality of electromagnetic radiation pulses.

[0019] In some embodiments, determining the composite interference signal further comprises determining the composite interference signal based on applying a depth-dependent calibration for correcting for an additional phase shift from the detection system.

[0020] In some embodiments, acquiring in-phase data further includes: acquiring an in-phase frame associated with a first phase shift; acquiring orthogonal data further includes: acquiring an orthogonal frame associated with a second phase shift; and the method further includes: generating a phase difference frame based on a phase difference between the in-phase frame and the orthogonal frame, determining a correction factor based on the phase difference frame, applying the correction factor to the in-phase frame and the orthogonal frame, and determining a composite interference frame based on the corrected in-phase frame and the corrected orthogonal frame.

[0021] In some embodiments, determining the correction factor further comprises: determining an average phase difference based on the phase difference frame; and determining the correction factor based on a difference between the average phase difference and the 90 degree phase shift.

[0022] In some embodiments, determining the average phase difference further comprises: determining a histogram of phase differences in the phase difference frame; and identifying the average phase difference based on the histogram of phase differences.

[0023] In some embodiments, the electromagnetic radiation source comprises an optically subsampled wavelength stepped source (OSWSS).

[0024] In some embodiments, the electromagnetic radiation source comprises a chirped fiber Bragg grating stretched pulse mode locked (CFBG-SPML) laser.

[0025] In some embodiments, the CFBG-SPML laser includes an imaging band of 1.3 μm.

[0026] In some embodiments, the phase modulator comprises a lithium niobate phase modulator.

[0027] In another embodiment of the present invention, an apparatus is provided, the apparatus comprising: an optical interferometer system, the optical interferometer system comprising a sample arm and a reference arm; at least one phase modulator, the at least one phase modulator being associated with at least one of the reference arm or the sample arm of the optical interferometer system; an electromagnetic radiation source, the electromagnetic radiation source being coupled to the optical interferometer system, the electromagnetic radiation source scanning a sample during a time period, the time period comprising a first time period and a second time period, a sample portion of the electromagnetic radiation source being directed to the sample in the sample arm of the optical interferometer system, a reference portion of the electromagnetic radiation source being directed to the reference arm of the optical interferometer system, and a phase shift comprising a first phase shift and a second phase shift being applied to the electromagnetic radiation source by the at least one phase modulator. at least one of a reference portion or a sample portion of a radiation source, applying a first phase shift during a first time period and applying a second phase shift during a second time period, the second phase shift being 90 degrees out of phase with the first phase shift; and a microprocessor coupled to the phase modulator and the electromagnetic radiation source, the microprocessor being used to: collect in-phase data based on a first interference between first backscattered electromagnetic radiation during the first time period and at least one of the reference portion or the sample portion subjected to the first phase shift; collect orthogonal data based on a second interference between second backscattered electromagnetic radiation during the second time period and at least one of the reference portion or the sample portion subjected to the second phase shift; and determine a composite interference signal based on the in-phase data and the orthogonal data.

[0028] In some embodiments, the first phase shift is 0 degrees during the first time period and the second phase shift is 90 degrees during the second time period.

[0029] In some embodiments, a phase shift is applied to a reference portion of an electromagnetic radiation source through at least one phase modulator, and when collecting in-phase data, the microprocessor is further used to: collect in-phase data based on a first interference between a first backscattered electromagnetic radiation during a first time period and the reference portion subjected to a first phase shift, and when collecting orthogonal data, the microprocessor is further used to: collect orthogonal data based on a second interference between a second backscattered electromagnetic radiation during a second time period and the reference portion subjected to a second phase shift, and when determining a composite interference signal, the microprocessor is further used to: determine the composite interference signal based on the in-phase data and the orthogonal data.

[0030] In some embodiments, an electromagnetic radiation source emits a plurality of electromagnetic radiation pulses including a first A-line and a second A-line, the first A-line including a first subset of the plurality of electromagnetic radiation pulses emitted during a first time period and the second A-line including a second subset of the plurality of electromagnetic radiation pulses emitted during a second time period, and the electromagnetic radiation source is further used to: scan the sample using the first A-line during the first time period and scan the sample using the second A-line during the second time period.

[0031] In some embodiments, a first subset of the plurality of electromagnetic radiation pulses corresponds to a particular wave number sequence and a second subset of the plurality of electromagnetic radiation pulses corresponds to a particular wave number sequence.

[0032] In some embodiments, a first wave number in a particular wave number sequence is different from a second wave number in the particular wave number sequence.

[0033] In some embodiments, a first A-line is scanned at a first location in the sample, and a second A-line is scanned at a second location in the sample that is different from the first location.

[0034] In some embodiments, the electromagnetic radiation source emits a third A-line, which includes a third subset of multiple electromagnetic radiation pulses emitted during a third time period and corresponding to a specific wave number sequence. The electromagnetic radiation source is further used to: scan the sample using the third A-line during the third time period; the phase modulator is further used to: apply a 0 degree phase shift to a reference portion of the electromagnetic radiation source during the third time period; the in-phase data includes first in-phase data, and the orthogonal data includes second orthogonal data. When collecting the in-phase data, the microprocessor is further used to: collect the third in-phase data based on the third interference between the third backscattered electromagnetic radiation during the third time period and the reference portion shifted by 0 degrees; and when determining the composite interference signal, the microprocessor is further used to: determine the second in-phase data based on interpolation between the first in-phase data and the third in-phase data, and determine the composite interference signal based on the second in-phase data and the second orthogonal data.

[0035] In some embodiments, the electromagnetic radiation source emits a beam having a beam diameter, and the distance between the first location and the second location is one quarter of the beam diameter or less.

[0036] In some embodiments, the electromagnetic radiation source is further configured to emit a plurality of electromagnetic radiation pulses, wherein each of the plurality of electromagnetic radiation pulses is divided into two time periods corresponding to a first time period and a second time period.

[0037] In some embodiments, the microprocessor is further used to: provide a modified electromagnetic radiation source based on combining multiple electromagnetic radiation pulses with delayed copies of the multiple electromagnetic radiation pulses; and the electromagnetic radiation source is further used to: scan the sample using the modified electromagnetic radiation source, wherein the modified electromagnetic radiation source alternately emits multiple electromagnetic pulses that each occur during a first time period and a corresponding multiple delayed pulses that each occur during a second time period.

[0038] In some embodiments, when providing a modified electromagnetic radiation source, the microprocessor is further used to: provide a modified electromagnetic radiation source based on combining multiple electromagnetic radiation pulses with delayed copies of the multiple electromagnetic radiation pulses, wherein the delayed copies of the multiple electromagnetic radiation pulses are delayed for a time shorter than the time of one of the multiple electromagnetic radiation pulses.

[0039] In some embodiments, determining the composite interference signal further comprises determining the composite interference signal based on applying a depth-dependent calibration for correcting for an additional phase shift from the detection system.

[0040] In some embodiments, when acquiring in-phase data, the microprocessor is further used to: acquire an in-phase frame associated with a first phase shift; when acquiring orthogonal data, the microprocessor is further used to: acquire an orthogonal frame associated with a second phase shift; and the microprocessor is further used to: generate a phase difference frame based on the phase difference between the in-phase frame and the orthogonal frame, determine a correction factor based on the phase difference frame, apply the correction factor to the in-phase frame and the orthogonal frame, and determine a composite interference frame based on the corrected in-phase frame and the corrected orthogonal frame.

[0041] In some embodiments, when determining the correction factor, the microprocessor is further configured to: determine an average phase difference based on the phase difference frame; and determine the correction factor based on a difference between the average phase difference and the 90 degree phase shift.

[0042] In some embodiments, when determining the average phase difference, the microprocessor is further configured to: determine a histogram of phase differences in the phase difference frame; and identify the average phase difference based on the histogram of phase differences.

[0043] In some embodiments, the electromagnetic radiation source comprises an optically subsampled wavelength stepped source (OSWSS).

[0044] In some embodiments, the electromagnetic radiation source comprises a chirped fiber Bragg grating stretched pulse mode locked (CFBG-SPML) laser.

[0045] In some embodiments, the CFBG-SPML laser includes an imaging band of 1.3 μm.

[0046] In some embodiments, at least one phase modulator comprises a lithium niobate phase modulator.

[0047] In yet another embodiment of the present invention, a method is provided, the method comprising: using an electromagnetic radiation source to scan a sample within a time period, the time period comprising a first time period and a second time period, the sample portion of the electromagnetic radiation source being directed to the sample in a sample arm of an optical interferometer system, and the reference portion of the electromagnetic radiation source being directed to a reference arm of the optical interferometer system; using a phase modulator to apply a phase shift comprising a first phase shift and a second phase shift to at least one of the reference portion or the sample portion of the electromagnetic radiation source, the first phase shift being applied during the first time period and the second phase shift being applied during the second time period, the second phase shift being 90 degrees out of phase with the first phase shift; The invention relates to a method for collecting in-phase frame data based on a first interference between first backscattered electromagnetic radiation during a first time period and at least one of a reference portion or a sample portion subjected to a first phase shift; collecting orthogonal frame data based on a second interference between second backscattered electromagnetic radiation during a second time period and at least one of a reference portion or a sample portion subjected to a second phase shift; and generating a phase difference frame based on a phase difference between the in-phase frame data and the orthogonal frame data; determining a correction factor based on the phase difference frame; applying the correction factor to the in-phase frame data and the orthogonal frame data; and determining a composite interference frame based on the corrected in-phase frame and the corrected orthogonal frame.

[0048] In some embodiments, determining the correction factor further comprises: determining an average phase difference based on the phase difference frame; and determining the correction factor based on a difference between the average phase difference and the 90 degree phase shift.

[0049] In some embodiments, determining the average phase difference further comprises: determining a histogram of phase differences in the phase difference frame; and identifying the average phase difference based on the histogram of phase differences.

[0050] In some embodiments, the electromagnetic radiation source comprises a chirped fiber Bragg grating stretched pulse mode locked (CFBG-SPML) laser.

[0051] In some embodiments, the CFBG-SPML laser includes an imaging band of 1.3 μm.

[0052] In some embodiments, the phase modulator comprises a lithium niobate phase modulator.

[0053] In yet another embodiment of the present invention, an apparatus is provided, the apparatus comprising: an optical interferometer system comprising a sample arm and a reference arm; at least one phase modulator associated with at least one of the reference arm or the sample arm of the optical interferometer system; an electromagnetic radiation source coupled to the optical interferometer system, the electromagnetic radiation source scanning a sample over a time period, the time period comprising a first time period and a second time period, a sample portion of the electromagnetic radiation source being directed to the sample in the sample arm of the optical interferometer system, a reference portion of the electromagnetic radiation source being directed to the reference arm of the optical interferometer system, and a phase shift comprising a first phase shift and a second phase shift being applied to at least one of the reference portion or the sample portion of the electromagnetic radiation source by the at least one phase modulator, the first phase shift being applied during the first time period and the second phase shift being applied during the first time period A second phase shift is applied during the second time period, and the second phase shift is 90 degrees out of phase with the first phase shift; and a microprocessor is coupled to the phase modulator and the electromagnetic radiation source, and the microprocessor is used to: collect in-phase frame data based on a first interference between the first backscattered electromagnetic radiation during the first time period and at least one of the reference part or the sample part subjected to the first phase shift; collect orthogonal frame data based on a second interference between the second backscattered electromagnetic radiation during the second time period and at least one of the reference part or the sample part subjected to the second phase shift; and generate a phase difference frame based on the phase difference between the in-phase frame data and the orthogonal frame data; determine a correction factor based on the phase difference frame; apply the correction factor to the in-phase frame data and the orthogonal frame data; and determine a composite interference frame based on the corrected in-phase frame data and the corrected orthogonal frame data.

[0054] In some embodiments, when determining the correction factor, the microprocessor is further configured to: determine an average phase difference based on the phase difference frame; and determine the correction factor based on a difference between the average phase difference and the 90 degree phase shift.

[0055] In some embodiments, when determining the average phase difference, the microprocessor is further configured to: determine a histogram of phase differences in the phase difference frame; and identify the average phase difference based on the histogram of phase differences.

[0056] In some embodiments, the electromagnetic radiation source comprises a chirped fiber Bragg grating stretched pulse mode locked (CFBG-SPML) laser.

[0057] In some embodiments, the CFBG-SPML laser includes an imaging band of 1.3 μm.

[0058] In some embodiments, at least one phase modulator comprises a lithium niobate phase modulator.

[0059] The above and other aspects and advantages of the present disclosure will become apparent through the following description. In this description, reference is made to the accompanying drawings which form a part hereof, and in which one or more example versions are shown by way of illustration. These versions do not necessarily represent the full scope of the present invention. BRIEF DESCRIPTION OF THE DRAWINGS

[0060] The following drawings are provided to help illustrate various features of example embodiments of the present disclosure and are not intended to limit the scope of the present disclosure or to exclude alternative implementations.

[0061] Figure 1 Shown is an optically subsampled wavelength-stepped source (OSWSS) (top), a schematic diagram of the power output of the OSWSS over time for a series of pulses ts_i to ts_i+12 (middle), an arrangement of the wavenumbers of the OSWSS for each of the series of pulses ts_i to ts_i+12 (bottom), and a close-up of two such wavenumbers k_2 and k_1 (bottom, inset).

[0062] Figure 2 An embodiment of an interferometric system using an OSWSS is shown.

[0063] Figure 3 The power and phase modulation levels are shown for an embodiment where the phase switches between 0 degrees for even time slots and 90 degrees for odd time slots (upper panel), or where the phase switches between 0 degrees and 90 degrees every six time slots (lower panel).

[0064] Figure 4A The wave number, power and phase modulation levels are shown for an embodiment in which the wave numbers k_1 to k_5 are repeated every five time slots and the phase alternates between 0 degrees and 90 degrees every five time slots.

[0065] Figure 4B The power and phase modulation levels are shown for an embodiment where the phase is switched between 0 and 90 degrees within a time slot.

[0066] Figure 5 An embodiment of an interferometric system is shown that includes a phase modulator in a reference arm and has a controller for coordinating the functions of one or more of the OSWSS, the phase modulator, the detector(s), and the computing system.

[0067] Figure 6 The positions of the scanned beam at two locations corresponding to A-lines Al and A3 are shown, along with an indication of the other A-line positions, phase modulation states, and possible interpolations that may be performed between the A-lines.

[0068] Figure 7An embodiment of a beam scan pattern superimposed on a phase modulation pattern is shown for a series of frames i-1 to i+8 (where a frame may include a series of A-line scans) (left panel), where each alternating frame has a different phase (e.g., 0 degrees or 90 degrees), and a schematic diagram of the x and y positions of the scans over time (right panel).

[0069] Figure 8 Shows Figure 7 The x and y positions of the alternative embodiment shown in , where the y position is moved in a stepwise manner after two frame scans (e.g., one with a phase of 0 degrees and one with a phase of 90 degrees) in order to obtain properly registered composite demodulated data.

[0070] Fig. 9 A wavelength-stepping light source (S) is shown, which produces a wavelength-swept comb that produces a beam Bl that is amplified by a gain medium G and emitted as an output beam B8.

[0071] Fig.10 shows the use of Fig. 9 The light source LS shown in exemplifies an embodiment of a Mach-Zehnder interferometer of an embodiment of the present invention, wherein panel A shows an embodiment based on free-space optics and panel B shows an embodiment based on fiber optics.

[0072] Fig.11 Embodiments of phase modulation patterns used in conjunction with continuous wave (CW) or quasi-CW lasers are shown.

[0073] Fig.12 Embodiments of phase modulation patterns used in conjunction with a pulsed laser output are shown, wherein panel A shows a sampling pattern with a pulsed laser output, panel B shows a sampling and phase modulation pattern in conjunction with a pulsed laser output and a delayed copy of the pulsed laser output, and panel C shows a sampling and phase modulation pattern in conjunction with a pulsed laser output and a further delayed copy of the laser output.

[0074] Fig.13 An embodiment of a wavelength-stepped light source (S) coupled to an optical assembly that combines the output of source S with a delayed replica of source S is shown.

[0075] Fig.14Schematic diagram of the SPML laser and circular ranging imaging system is shown: LD, laser diode driver; DDG, digital delay generator; PG, pattern generator; A, amplifier; EOM, electro-optic modulator; PC, polarization controller; CFBG, continuous fiber Bragg grating; SOA, semiconductor optical amplifier; FP, Fabry-Perot etalon spectral filter; C, circulator; PZT, piezoelectric actuator; PM, partial mirror; OSA, optical spectrum analyzer; VD, variable delay line; ISO, optical isolator; FG, signal generator; D, dispersion compensation; LO, local oscillator; CAL, calibration signal; FC, fiber collimator; G, galvanometer mirror; L, lens; S, sample; BD, balanced photodiode; DAQ, data acquisition board.

[0076] Fig.15 Measured SPML laser performance is shown, including panels showing the following: (a) Spectrum. (b) Zoomed-in spectrum. (c) Time trace of a single scan using a 260ps electrical pulse. (d) Zoomed-in time trace showing the pulsation. (e) Time trace showing five scans. (f) Measured coherence length.

[0077] Fig.16 Frame demodulation is shown, including panels showing the following: (a) Schematic diagram of frame demodulation. (b), (c) Artifact suppression of image signals at baseband showing the original dispersion compensated signal (b) and the I, Q demodulated signal (c). FSR = 150 GHz. -1, baseband and 1st order signals are indicated. (d), (e) IR card imaging showing the original dispersion compensated image (d) and the I, Q demodulated image (e). (f), (g) Tape imaging showing the original dispersion compensated image (f) and the I, Q demodulated image (g). A FSR of 100 GHz was used during imaging. The scale bar corresponds to 1 mm.

[0078] Fig.17 Illustrations of phase correction using IR cards (top row) and tape (bottom row), including panels: (a), (b) are images with residual artifacts after I, Q demodulation (yellow arrows). (c), (d) are the phase differences between phase modulation frames. (e) and (f) represent the data from (c) and (d). The histogram is used to obtain the global phase offset from the quadrature points. (g) and (h) are phase-corrected images. (i) Phase error (offset) of the ideal quadrature point for 251 consecutive recorded frames. (j) The corresponding measured suppression due to the phase error in (i) before (lower black line) and after correction (upper red line). The bold line shows the average suppression using 5 frames. The scale bar corresponds to 1 mm.

[0079] Fig.18 Examples of imaging using frame demodulation are shown, including panels: (a) Imaging of a human nail wrinkle showing the original dispersion compensated image and the I,Q demodulated images. The demodulated images are stitched to make the borderless wrapping of structures beyond the baseband range clearer. (b) Imaging displayed on an iPhone 7, showing the original and I,Q demodulated images and the volume map. The different layers can be seen below the top glass plate. The scale bar corresponds to 1 mm.

[0080] Fig.19 A-line demodulation is shown, including panels: (a) is a schematic diagram of A-line demodulation. (b), (c) are artifact suppression of image signals at baseband, showing the original dispersion compensated signal (b) and the I, Q demodulated signals (c). FSR = 100 GHz. The -1 order signal, baseband signal, and 1 order signal are indicated. (d) is the complex conjugate suppression of the image signal measured over a period of one hour.

[0081] Fig. 20 Composite interpolation is shown, including panels: (a) measured suppression before (circles) and after (squares) composite interpolation according to the beam step size Δx (normalized to the beam spot size δx). (b) is an imaging example of an IR card, illustrating composite interpolation. It shows the original image and the I,Q demodulated images. The demodulated images are stitched three times to make the borderless wrapping of sample structures beyond the baseband range clearer. FSR is 100GHz (L B =1.5 mm). The scale bar corresponds to 1 mm.

[0082] Fig.21 Examples of imaging using A-line demodulation are shown, including panels: (a) A cross-sectional image of a human nail fold showing the original dispersion compensated image (upper left) and the I,Q demodulated image (lower left). The demodulated image is stitched twice (right) to make the borderless wrapping of structures beyond the main measurement range of the frequency comb clear. (b) An I,Q demodulated image of a human tooth, showing the enamel (1), dentin (2), and gingiva (3). For illustrative purposes, we show the entire acquisition range here, with the baseband range represented by the dashed yellow line. (c) Volumetric depth projection of a human tooth using a 50 mm lens (left) and a 150 mm lens (right) at a rate of 15 volumes per second. The scale bar corresponds to 1 mm.

[0083] Fig. 22 An example of a system for active quadrature demodulation for circular ranging optical coherence tomography imaging is shown in accordance with some embodiments of the disclosed subject matter.

[0084] Fig.23Examples of hardware that can be used to implement computing devices and servers in accordance with some embodiments of the disclosed subject matter are shown.

[0085] Fig.24 An example of a process for active quadrature demodulation for circular ranging optical coherence tomography imaging is shown in accordance with some embodiments of the disclosed subject matter.

[0086] Fig.25 An example of another process for active quadrature demodulation for circular ranging optical coherence tomography imaging is shown in accordance with some embodiments of the disclosed subject matter. DETAILED DESCRIPTION

[0087] Before explaining any embodiment of the present invention in detail, it should be understood that the present invention is not limited to the construction details and component arrangements set forth in the following specification or shown in the accompanying drawings in this application. The present invention can be applied to other embodiments and practiced or executed in various ways. Moreover, it should be understood that the wording and terminology used herein are for the purpose of description and should not be considered as restrictive. For example, the use of "including", "comprising" or "having" and its variants is meant to cover the items listed later and their equivalents and additional items.

[0088] In various embodiments, the present invention provides methods and apparatus for generating a composite interferometric signal and using the signal to demodulate data obtained using subsampling / circular ranging optical coherence tomography. Various techniques are disclosed for obtaining orthogonal electric field components (π / 2 / 90 degree phase shift relative to each other) by actively (electronically) controlling the phase of the interferometric signal to obtain the composite interferometric signal and mitigate imaging artifacts at very high imaging speeds (e.g., at MHz laser repetition rates).

[0089] In certain embodiments, a preferred implementation for phase modulation uses a lithium niobate (LiNbO3) phase modulator device that converts a voltage signal into an optical phase shift, but other methods of phase modulating a light beam may also be used. Other methods of inducing a phase shift in a light beam may also be used, including but not limited to other electro-optic phase modulators (such as those based on silicon or semiconductor materials), thermal phase modulators that use temperature to affect the phase of a light beam, and acousto-optic phase modulators that convert acoustic waves or acoustic energy into optical phase shifts. An example architecture incorporating a phase modulator is described below.

[0090] Thus, disclosed herein are techniques for using active phase shifting (e.g., phase modulation) in either or both of the sample and reference arms of an interferometer system to generate a composite interferometer signal associated with each time period (referred to herein as a "time slot," see below) of an optically subsampled wavelength stepped source (OSWSS). The techniques may include: hardware embodiments describing the placement and drive of a phase shifting arrangement in an interferometer; an arrangement for synchronizing the phase shifted signal to the OSWSS and / or a detector device and / or a microscope; a signal processing method for generating a composite signal from a measured signal; and a signal processing method and arrangement for calibrating and compensating for demodulation errors from various sources.

[0091] The technique can be performed in an "inter-pulse" manner or an "intra-pulse" manner. The inter-pulse technique uses at least two different pulses (which may or may not have the same wave number; see below for a discussion of wave number) to generate a composite interference signal associated with a single wave number. The intra-pulse technique uses measurements within a single pulse to generate a composite signal associated with the wave number of that pulse. The inter-pulse method is easier to implement, but may require more additional pulses for each composite measurement and, therefore, may reduce imaging speed. The intra-pulse method is more complex to implement, but can achieve the fastest imaging by requiring only a single pulse to collect the data required to generate the composite signal (i.e., two signals that are 90 degrees phase shifted relative to each other).

[0092] Optically Second-Sampled Wavelength Stepped Source (OSWSS)

[0093] Disclosed herein are apparatus and methods for performing quadrature demodulation (also referred to as in-phase (I) / quadrature (Q) demodulation or I / Q demodulation) in the context of an interferometric measurement system using a particular source, namely an optically subsampled wavelength stepped source (OSWSS). The following is an overview of an OSWSS source that can be used in implementations of the I / Q demodulation embodiments disclosed herein.

[0094] Figure 1 The OSWSS schematically shown in (top) provides a specific optical output signal having one or more of the following properties and parameters:

[0095] - The light output provided by the OSWSS is discretely stepped in time between light wavelengths (also called light wavenumbers (i.e., spatial frequency of the waves) or optical frequencies);

[0096] -OSWSS output can be divided into a set of consecutive time slots, where ts_i refers to the i-th time slot ( Figure 1 ,central);

[0097] - Each time slot ts_i has a specific duration. The time slots usually have the same duration, but do not necessarily have to, and in various embodiments have different durations.

[0098] - A time slot is defined as a period of time in which the laser provides light with a specific optical wave number, and the time slot ends when the laser starts providing light with a different optical wave number. During the transition across the time slot, the laser wave number transitions discretely between wave numbers. This Figure 1 (Bottom) middle picture, Figure 1 (Bottom) shows an exemplary time / wavenumber relationship. Here, ts_i produces wavenumber k_2, while ts_i+1 produces wavenumber k_4, and so on;

[0099] -OSWSS output has a certain finite linewidth Δk, such as Figure 1 (bottom, inset). Here, the line width is smaller than the average wavenumber difference between adjacent time slots, indicating that the source provides a different wavenumber output at each time slot. This clearly shows the difference between this laser and a source with a continuously swept wavenumber output; and

[0100] -The light in the time slot is called a pulse. Figure 1 (center), it can have a varying power that rises and falls during each time slot, or it can have a constant power. In both cases, the light within a time slot is referred to herein as a pulse because it describes a finite duration of light at an associated wavenumber.

[0101] Examples of OSWSS sources include sources designed to generate wavenumbers equally spaced in wavenumber space, such as frequency comb sources.

[0102] Interferometry using OSWSS

[0103] Figure 2 A general setup for interferometry using an OSWSS is shown. Components include a source such as an OSWSS; an interferometer, including a sample arm and a reference arm; an interferometer output; a detector; and a signal processing computer / arrangement, in various embodiments, including a processor / microprocessor. Using an OSWSS with an interferometer can allow range measurements (such as those performed in optical subsampled optical coherence tomography), or more generally, allows interrogation of an object in the sample arm using interferometric methods. I / Q demodulation is used to measure the composite output signal from the interferometer, as described below.

[0104] Without I / Q demodulation, a (non-composite) interference signal can be measured. This signal is proportional to the sine of the phase difference between the sample and reference arm light. On the other hand, with I / Q demodulation, a composite interference signal can be measured, which has a real component proportional to the sine of the aforementioned phase difference and an imaginary component proportional to the cosine of the aforementioned phase difference. Using I / Q demodulation, the composite interference signal obtained from a given positive optical delay (+d) in an interferometric ranging system can be distinguished from the composite interference signal given by a negative optical delay (-d) of the same amplitude, a distinction that cannot be made when measuring a non-composite interference signal. In subsampled OCT, this ability to distinguish between positive and negative delays allows circular ranging (CR) to reduce the large physical delay space to a reduced circular delay space.

[0105] In such Figure 2 In the simplest implementation of the interferometer shown in , and without I / Q demodulation, both output signals 1 and 2 will be proportional to the sine of the phase difference between the sample arm light and the reference arm light. I / Q demodulation can be performed in such a system using methods such as redesigning the way the sample and reference arm light are combined (e.g., using polarization-based optical demodulation), or as disclosed herein, by modulating the phase of the light in one or both of the sample and reference arms at different points in time, so that at one point in time the output is proportional to the sine of the phase difference, while at a later point in time the output is proportional to the cosine of the phase difference. More generally, measurements collected over time can be combined together (e.g., using a computer processor) to generate composite fringes using additional signal processing.

[0106] In various embodiments, pulse-to-pulse phase modulation comprises modulating the phase of light in a reference arm or a sample arm between pulses (or equivalently, between time slots) such that, for example, data obtained from a set of two pulses can be used to produce a composite signal in which the first pulse produces a sine component and the second pulse produces a cosine component.

[0107] exist Figure 3In the example shown in (upper panel), a phase modulation of 0 degrees is induced for even-numbered time slots / pulses (e.g., ts_i, ts_i+2, etc.), while a phase modulation of 90 degrees is induced for odd-numbered time slots / pulses (e.g., ts_i+1, ts_i+3, etc.). Although in many of the embodiments disclosed herein, the phase is indicated as varying between 0 and 90 degrees, in various embodiments, the phase may vary between other angles, provided that a difference of 90 degrees (π / 2) is assumed between these angles. In this example, assuming that time slots / pulses ts_i and ts_i+3 have the same wave number k, pulse ts_i can be used to measure a sinusoidal interference signal associated with wave number k, while pulse ts_i+3 can be used to measure a cosine (due to the introduction of a 90 degree phase shift into the data associated with pulse ts_i+3) interference signal associated with wave number k. Figure 3 In the example shown in the lower panel of , the phase modulation changes every six pulses. Here, assuming that time slots / pulses ts_i and ts_i+6 have the same wave number k, time slots / pulses ts_i and ts_i+6 can be used to determine the sine and cosine interference signals, respectively, to generate a composite fringe signal associated with the wave number k.

[0108] Phase modulation between consecutive A lines

[0109] In some embodiments, phase modulation may be performed between consecutive A-lines. In one particular embodiment, the A-line includes a set of pulses having a particular wave number sequence, and the wave number sequence is repeated for each subsequent A-line. Figure 4A The output of an OSWSS with five wave numbers numbered k_1 to k_5 is shown, where each set of wave numbers k_1 to k_5 represents a single A-line. Thus, A-lines y and y+1 (using the same wave number sequence) measure the same property, but at different times. In this embodiment, adjacent / sequential A-lines are phase shifted to have a phase shift of Figure 4A The 90 degree relationship shown in . The signal from A line y is used to form the sine (real) fringe signal, while the signal from A line y+1 is used to form the cosine (imaginary) fringe signal. After combination, A lines y and y+1 form a single composite A line that can be used to perform I / Q demodulation.

[0110] In some embodiments, phase modulation may be performed within time slots, including embodiments where phase modulation occurs within pulses. Figure 4B In this case, a single time slot is divided into at least two smaller time slots, wherein a first phase signal is applied during a first time slot and a second phase signal is applied during a second time slot. Figure 4B In this way, both the in-phase and quadrature components can be measured for a single laser pulse / time slot. Here, at least two measurements are used per time slot.

[0111] Figure 5 An example architecture incorporating a phase modulator (e.g., a lithium niobate (LiNbO3) phase modulator device) is shown in FIG. Figure 5 An embodiment of an interferometer system is shown that includes a phase modulator in a reference arm and has a controller (electronic control device and signal generator, which may include a processor / microprocessor) for coordinating the functions of one or more of the OSWSS, the phase modulator, (multiple) detectors, and a computing system.

[0112] In some embodiments, a beam that emits a series of A-lines (each A-line comprising the same sequence of wave numbers) may be applied to the sample to acquire data as the beam is scanned across the sample. In an OCT system such as this one, where the beam is scanned across the sample, the two A-lines used to construct the composite fringes described herein may be from scans of different portions of the sample, and thus in some cases the data may need to be corrected to account for sample differences between paired data points.

[0113] This phenomenon is Figure 6 It is shown in Figure 6 The beam profiles (circles) for A-lines A1 and A3 are shown. Here, A-lines A1 and A3 provide in-phase (I or sine) fringes, while A-line 2 provides quadrature (Q or cosine) fringes. Ideally, the I and Q A-lines would be measured at the same sample position. However, if A-lines A1 and A2 were used to construct the composite fringe in this example, phase relationship errors could occur due to beam displacement between the A-lines; similar situations would occur if A-lines A2 and A3 were used. To reduce this error, a new (I) A-line can be constructed from the measured A-lines A1 and A3 by interpolation. This interpolated A-line approximates the measurement of the IA-line at position A2 (in Figure 6 Then, a composite fringe is generated from the measured QA line (A2) and the interpolated IA line formed by A1 and A3.

[0114] To achieve A-line interpolation, in various embodiments, the OSWSS and the data acquisition system may be synchronized with each other so that the A-lines are phase stable. Without phase stability, interpolation between A-lines A1 and A3 may result in errors due to unknown phase shifts between these A-lines. Phase synchronization allows direct interpolation without phase compensation of the A-lines. One embodiment of a phase synchronization scheme is Figure 5 , which depicts a controller / signal generator electronically coupled to the OSWSS, phase modulator, detector(s), and a computer (eg, which may be used for data collection and / or analysis).

[0115] The A-line interpolation may be determined in the fringe domain (ie, on the captured fringe data) or may be determined after performing an FFT on a single measured A-line fringe (ie, on the composite data generated by the FFT operation).

[0116] Phase modulation between consecutive frames

[0117] The phase modulation methods disclosed above do not necessarily need to be implemented in adjacent A-lines. For some imaging systems, it may be optimal to perform phase modulation between frames, where a frame may be a collection of A-lines (e.g., a collection of A-lines along a particular direction, such as the x-direction) that can be used to generate a continuous image. Thus, the beam may be repeatedly scanned in a first direction (e.g., the x-direction) across the sample while also translating in a second, typically orthogonal direction (e.g., the y-direction).

[0118] like Figure 7 As shown in the left panel of , a sinusoidal scan along the x direction is used, where every other frame has a 0 degree phase shift for generating in-phase (I) data, while every other A line has a 90 degree phase shift for generating quadrature (Q) data. If the scan along the y direction is slow enough at a constant speed, adjacent frames sample sufficiently similar positions that the data from these frames can be combined to generate a composite frame. For example, A lines at positions b and c can be used, where b and c are at the same x position but slightly offset in y position. Figure 7 The right panel in depicts the x-scan position and y-scan position versus time.

[0119] Alternatively, if Figure 8 As shown, the y scan can be modified to keep the y position constant during the two frame acquisitions so that the A-lines from the two frames have the same y position, and therefore the A-lines have the same x position and y position, which is beneficial for constructing the composite A-line / frame. Figure 8 It is shown in Figure 8 The relationship between the position in x and y and time is shown in . Here, frame i and frame i+1 can have a phase shift of 0 and 90 degrees respectively, and can be used to generate composite fringe / A line / frame data, where all data have the same y position.

[0120] The following non-limiting examples further illustrate embodiments of the invention disclosed herein.

[0121] Alternate A-line demodulation

[0122] Fig. 9 A wavelength-stepping source (S) is shown that produces a wavelength-swept comb with a frequency-comb free spectral range (FSR) and may or may not be pulsed in the time domain, where the pulse repetition rate (f P) depends on the FSR and the scanning speed. The light source S generates a beam B1 which is amplified using a gain medium (G) made of a semiconductor optical amplifier, an enhanced optical amplifier or another type of gain medium to obtain an output beam B8.

[0123] Fig.10 It is shown that free space optics can be used ( Fig.10 , Panel A) or Fiber Arrangement ( Fig.10 , Panel B) implements a Mach-Zehnder interferometer. Other interferometer types (e.g., Michelson) can also be applied. Fig.10 The light source LS in panel A or panel B indicates Fig. 9 Output beam B8 in. Beam B9 emitted from the LS is directed to the interferometer input where it is split into two paths of approximately equal length using a beam splitter (BS3). B10 is directed towards the sample S. The backscattered light from the object of interest is then directed towards the interferometer output (B11). In the reference arm, beam B12 is directed towards the phase modulator (PM). In this example, the PM applies a phase shift of zero or π / 2 (90 degrees) at a frequency of half the laser repetition (A-line) rate, meaning that every other group of pulses corresponding to the A-line has a phase shift of 0 or 90 degrees applied in the reference arm. Alternatively, the laser repetition rate can be an integer multiple of the PM frequency, f L =nf PM (n≥2) (e.g., the phase may change every 3, every 4, every 5, ..., or every n A lines). The beam after PM (i.e., beam B13) is directed toward the interferometer output to interfere with beam B11 after combining with BS4. The output beam B14 is then detected by the photodiode D. Alternatively, due to the phase shift π between the output beams B14 and B15, Fig.10 The fiber-based interferometer shown in panel B of FIG. 1 is easily used to implement balanced detection. Using a data acquisition board or a real-time oscilloscope (DAQ) with a sampling rate of f S The detected signal is digitized. Several wavelength scans (A1, A2, ..., An) can be acquired to form a two-dimensional or three-dimensional image.

[0124] During acquisition, the sample arm beam B10 is scanned in the lateral (or transverse) direction to form a two-dimensional image (B scan), e.g. Figure 6 As shown. In this case, the phase modulator PM induces a phase shift of zero and π / 2 (90 degrees) between alternating A lines that form the I and Q components of the composite interference signal, respectively. In some embodiments, the beam scanning speed is adjusted so that consecutive A lines are spatially spaced by a beam step size of 1 / 4 of the beam diameter or less. For a beam step size much smaller than the beam diameter, a composite interference signal (analytical signal) can be obtained as or Its complex conjugate term is suppressed after Fourier transform in the depth domain.

[0125] For beam step sizes approaching the beam diameter, the detected signals may become increasingly uncorrelated, resulting in a reduced reduction in the complex conjugate terms. As described above, complex value interpolation may be applied between An and An+2 (i.e., between adjacent IA lines and / or adjacent QA lines) to obtain corrected (interpolated) I' or Q' components at the spatial locations of their Q counterparts or I counterparts, respectively. In this case, the composite interference signal is then formed as or or or

[0126] Alternating sampling point demodulation for continuous wave (CW) or quasi-CW laser operation

[0127] In this embodiment, such as Fig. 9 The light source shown in is input to a Fig.10 The interferometer shown in Panel A or Panel B of FIG. The output beam B14 is detected by the photodiode D as shown above, or alternatively, due to the phase shift of π between B14 and B15, Fig.10 Balanced detection is easily accomplished using a fiber-based interferometer as shown in panel B. Using a data acquisition board or real-time oscilloscope (DAQ) with a sampling rate of f S The detected signal is digitized. In various embodiments, the sampling rate can be adjusted to twice the pulse repetition rate, f S =2f P =2v / FSR, where v is the scanning speed in Hz / s. The phase modulator PM is arranged to modulate the phase of the interference signal between zero and π / 2 (90 degrees) at a rate equal to half the pulse repetition rate and the digitizer (DAQ) sampling frequency.

[0128] The relative phase of sampling and phase modulation can be adjusted with respect to the pulse, e.g. Fig.11 In this embodiment, λ n (n = 1, 2, 3, 4, 5, ...) represents the comb lines of each "basic" narrow line width in the spectral domain. Each pulse collects two points, which are phase-shifted by π / 2 to form the I and Q components of the composite interference signal, or

[0129] Alternating sampling point demodulation of pulsed laser output

[0130] In the case of pulsed laser output, sampling between pulses may result in invalid I / Q components, such as Fig.12This may become more apparent for pulses with low duty cycles as shown in Panel A of Fig.12 In, λ n represents the comb lines of individual "fundamental" narrow linewidths in the spectral domain. In this embodiment, the sampling problem between pulses can be solved by generating a copy of the pulse train and pasting the copy into the low duty cycle region of the original pulse train, such as Fig.12 In other words, the same comb line (λ n ) and the pulses are copied and pasted with a certain delay (λ n ′), and is therefore sampled by alternating sampling points, each sampling point being phase-shifted by π / 2 relative to each other.

[0131] Fig.13 An embodiment of a wavelength stepping source (S) coupled to an optical assembly that combines the output of source S with a delayed replica of source S to produce a wavelength stepping source (S) such as Fig.12 The signal output is shown in panels B or C. Fig.13 A wavelength stepping light source S generates a wavelength swept frequency comb with a frequency comb free spectral range (FSR). The pulses of source S have a pulse width and a pulse repetition rate that depend on the FSR and the scanning speed. Source S generates beam B1, which is divided into two paths (P1, P2) by a beam splitter / coupler (BS1). Beam B2 is directed to a polarization controller (PC1) that ensures that the polarization between P1 and P2 is equal. After PC1, beam B3 is directed to a beam splitter (BS2) that combines P1 and P2. Beam B4 is directed to a delay line (DL) that causes a time delay between P1 and P2. In the absence of a delay line DL, paths P1 and P2 would have equal lengths, and therefore the replicated pulses would simply recombine to form a train of pulses equivalent to the output of source S.

[0132] In various embodiments, the time delay caused by DL may be equal to half the inverse of the pulse repetition rate (1 / (2f P )) corresponds to the replicated pulse train ( Fig.12 The dashed line in panel B of Figure 1 is delayed by half a pulse period. Fig.12 As shown in panel C of Figure 1 (where the pulse is delayed by one and a half pulse period), it is possible to paste λ in other regions of the signal gap (between pulses) n ' to apply the principle more generally. The applicable time delay can be more generally written as n / (2f P), where n is an odd integer. After the delay loop DL, beam B5 is directed to beam splitter BS2, which combines beam B3 with beam B5. This produces two interleaved pulse trains at the output of BS2. Both beams pass through a polarization controller (PC2) before being amplified by the gain medium G. The output beam B8 is then directed to a source such as Fig.10 The interferometers shown in panels A and B are used for imaging, where Fig.13 The arrangement is that the light source LS emits beam B9. In various embodiments, the sampling rate can be adjusted to twice the pulse repetition rate, f S =2f P = 2v / FSR, where v is the scanning speed in Hz / s. The phase modulator PM modulates the phase of the interference signal between zero and π / 2 at a frequency equal to the pulse repetition rate and half the sampling rate of the digital converter (DAQ). The I and Q components are extracted from the separately sampled pulses to form a composite interference signal or

[0133]

[0134] In an alternating sampling point demodulation embodiment for continuous wave (CW) or quasi-CW laser operation or an alternating sampling point demodulation embodiment for pulsed laser output, the phase modulation frequency and sampling rate can be similar to or equal to the analog bandwidth of the photodiode D and the digitizer DAQ. However, frequencies close to the bandwidth limit are subject to additional frequency-dependent phase shifts, which can result in erroneously measured I, Q components because the interferometric signal experiences a phase shift from the PM and a depth-dependent phase shift from the detection electronics. This in turn can result in inadequate suppression of complex conjugate artifacts. In certain embodiments, this additional phase shift can be corrected by including a depth-dependent calibration. The corrected quadrature components (I and Q) can be written as where Q' is the measured phase shift component of the complex fringe and α is the complex correction factor obtained by applying the condition 0 = I + Q'α at the position of the residual complex conjugate component.

[0135] Phase Error Correction

[0136] Certain embodiments of circular ranging optical coherence tomography (CR-OCT) systems help reduce the number of measurements required for imaging across an extended depth range in most applications. However, in some cases, the imaging system architecture used in these embodiments, while sufficient to demonstrate the core principles of the CR approach, may be unstable and may be limited to operation at 1.55 μm. Therefore, embodiments of improved CR-OCT architectures are presented that improve operational stability and / or convert the technology to other wavelength bands including the more common 1.3 μm range imaging band. In various embodiments, the long distances of dispersive optical fiber used when a stretched pulse mode locked (SPML) laser source is employed can be replaced by a single continuously chirped fiber Bragg grating (CFBG) design, a modification that not only transforms the operating wavelength range to a window from 1260 to 1360 nm, but also significantly stabilizes the laser operation. Further, embodiments of active quadrature demodulation schemes using lithium niobate phase modulators have been implemented to replace polarization-based optical quadrature demodulation circuits that may be sensitive to environmental factors. The performance of these modified systems has been quantified and imaging examples are provided with an operating speed of 7.6 MHz, a scan bandwidth of 100 nm, a coherence length of 4 cm, and a circular ranging depth adjustable over a wide range, which in some embodiments can range between 100 μm and 4 mm. In various embodiments, the combination of one or more of the stability and simplicity of the CFBG-SPML laser, operation at 1.3 μm, and / or an active quadrature demodulation scheme defines a more compact, stable, and straightforward to operate CR-OCT system, which may help broaden the exploration of CR-OCT technology in medical and non-medical applications.

[0137] Circular ranging optical coherence tomography (CR-OCT) can address two technical barriers faced by traditional Fourier domain OCT methods. First, when trying to interrogate an extended depth range at a high depth scan rate using FD-OCT methods, a large number of measurements (i.e., digitized samples) need to be made in a short duration. This can place high, often unattainable demands on the electronic bandwidth of the detection, digitization, transmission, and processing modules in the OCT system. This is particularly challenging in applications that require continuous (as opposed to burst mode) imaging. CR-OCT overcomes this obstacle by exploiting the inherent sparsity in depth-resolved profiles in most applications. Specifically, the depth space is circularly folded so that only a small circular depth range needs to be collected. This circular depth range can be set to approximate the depth range of the signal reflected from the sample, typically in the range of 0.5-2mm, but not limited to this range. However, the sample can be over a wider physical depth range that may span many centimeters. In short, CR-OCT achieves efficient sampling of the sample signal by discarding the absolute position information of the sample.

[0138] The second technical barrier that can be overcome by adopting a CR-OCT approach is that of imaging source speed. Conventional Fourier domain OCT lasers can be used comfortably in the 100-400kHz range and can be operated beyond 1MHz by optical buffering to circumvent the mechanical limitations of critical swept wavelength filters. CR-OCT is based on the use of frequency comb sources. The ability to use discretely stepped wavelengths rather than continuously swept wavelengths opens new opportunities for extremely fast, >10MHz source designs. Previous CR-OCT imaging results were obtained using stretched pulse mode-locking designs based on intracavity dispersion. Although sources such as these have demonstrated the core CR principle, they are complex and unstable; on the other hand, superior frequency comb laser sources based on chirped fiber Bragg gratings (CFBGs) with a theta cavity structure have been reported, but imaging with frequency comb CFBG-SPML sources has not been demonstrated so far. In previous work, CR-OCT was used to capture tissue signals with measured signal compression factors ranging from 10-42 at A-scan rates up to 22MHz.

[0139] In certain embodiments of CR-OCT systems, the source and interferometer may require frequent adjustment and calibration to maintain high performance. Factors that contribute to light source instability include the use of long (>10km) dispersion fibers. The use of long dispersion fibers can make the laser cavity resonant frequency sensitive to temperature, and since the entire system is phase-locked, changes in the laser drive frequency may need to be propagated to the beam scanning and digitizing clock sources. A more stable laser design will greatly simplify operation in both research and preclinical / clinical environments. In order to achieve the circular folding of the depth space that defines the technology, a composite fringe signal including in-phase and orthogonal signals is detected. In certain embodiments, a specific polarization-based orthogonal demodulation circuit can be used. This circuit, combined with a correction algorithm, can provide high-performance demodulation, but the correction parameters may have to be frequently remeasured through a fairly complex calibration process. Finally, a third drawback of certain designs that are not related to stability is that they cannot be converted to other imaging wavelengths, such as 1.3μm or 1.0μm. This is due to the use of matched positive and negative dispersion fibers, which are widely used above 1.5μm. To enable the technology to be studied in a clinical setting and to enable a more robust adoption of CR technology, a simplified, more stable laser and interferometer architecture operating at conventional imaging wavelengths is required.

[0140] CR-OCT techniques / systems operate on composite interference fringes to achieve circular ranging. Unfortunately, the use of frequency combs produces RF errors that can cause artifacts when imaging at multiples of the primary measurement range of the frequency comb free spectral range (FSR). In order to access the entire depth range, it is generally necessary to use a resolved (i.e., composite) interference signal. To avoid overlap of artifacts, signal folding is performed in a cyclic manner through a combination of the use of a frequency comb light source and complex demodulation (i.e., detecting the in-phase (I) and quadrature (Q) fringe signals of the sign of the measured delays resolved), as disclosed herein. In conventional Fourier domain OCT, complex demodulation methods have been explored to expand the imaging range by a factor of two by using positive and negative delay spaces. Passive methods based on optical orthogonal circuits and active methods based on dynamic phase modulation have been demonstrated. For conventional OCT, complex demodulation is optional because the entire depth signal can be placed in both the positive and negative delay spaces. The same strategy can be applied to optical subsampling, but the depth range is limited to the fundamental frequency comb order because aliasing exceeds the primary measurement range. Therefore, circular ranging can be achieved by a combination of optical subsampling and complex demodulation. That is, in the case of CR-OCT, the composite signal is not optional but essential. In some CR-OCT demonstrations, passive methods have been adopted. Although this provides the desired performance, active methods have some advantages, reducing the channel count of the digitizer and avoiding the need for environmental stability. We have demonstrated circular ranging by frequency shifting using an acousto-optic modulator and a Fourier-domain mode-locked frequency comb laser. This method can provide stable single-channel intra-A-line complex demodulation, but may be limited to laser repetition rates up to ∼3 MHz. Therefore, this paper describes the extension of the active demodulation scheme to CR-OCT systems. Specifically, by describing two inter-A-line demodulation techniques, an embodiment of CR-specific changes for an efficient, high-speed and stable method based on a LiNbO3 electro-optic modulator (EOM) is highlighted. In addition, in order to move the high speed of the CFBG-based SPML architecture to the more relevant OCT wavelength, a 1.3μm SPML laser is demonstrated for the first time, which provides a compact and stable SPML-based circular ranging system for OCT imaging.

[0141] set up

[0142] Fig.14An optical configuration including an SPML laser and a Mach-Zehnder interferometer is shown. An electro-optic modulator EOM (MX1300-LN-10, Photline) is driven resonantly at a harmonic of the cavity round-trip time of 3.87 GHz. An electrical signal consisting of 1024 bits (32 words) is generated using a pattern generator PG (PAT 5000, SYMPULS) and amplified by an amplifier A (DR-PL-10-MO, iXblue) to produce a 520ps pulse, where a single bit corresponds to a 260ps pulse. An additional signal generator (SG384, Stanford Research Systems) is used to generate an external clock signal for the pattern generator. A continuous fiber Bragg grating (CFBG, Proximion) is placed between two circulators to access both normal and anomalous dispersion. The 9.5m grating is designed to produce a linear group delay relative to the optical frequency over a continuous wavelength range of 1240nm to 1340nm (Δλ=100nm). The dispersion at 1290nm is ±930ps / nm (98,000ps / nm / km), corresponding to a sweep time of 93ns. By using CFBG from both directions, we generate matched normal and anomalous dispersion from the same device. The CFBG-based SPML design is critical for implementation in the 1.3μm range, where the dispersion of standard SMFs is zero. The laser output is tapped after the SOA using a 20% output tap coupler (at Fig.14To generate the frequency comb output, a fixed Fabry-Perot etalon with a single-pass finesse of 12 and a continuously adjustable free spectral range (FSR) was used in the cavity. The etalon was constructed using two glass plates (Korea Electro-Optics Co., Ltd.). Both plates have flat surfaces with a reflectivity of 85% on one side and an inclined surface on the other side to mitigate back reflections. The measured single-pass finesse is lower than the expected finesse of 18, probably due to surface irregularities and / or deviations from the set reflectivity. Note that this frequency comb is nested on top of the ~3.8 MHz frequency comb of the actively mode-locked laser cavity, where only the nested frequency comb is relevant for imaging. Within the CFBG passband, approximately 30% of the light is transmitted, forming three optical cavities (A, B, AB). To suppress light from circulating in cavities A and B, we use SOA modulation (T160, Highland Technology) with a frequency given by the cavity AB round-trip time and a conduction state determined by the CFBG dispersion (93 ns). The modulation is applied using a digital delay generator DDG (DG645, Stanford Research Systems) triggered by a pattern generator. The external clock generator, pattern generator and digital delay generator are phase synchronized using a 10 MHz reference signal, with the clock generator acting as a master clock. The implemented configuration has a 38% duty cycle, resulting in a 3.8 MHz repetition rate at the laser output. The repetition rate is doubled to 7.6 MHz (76% duty cycle) using a delay line made of 27 m SMF28 followed by post-amplification. The repetition rate can be further optimized to a maximum of 10 MHz by reducing the cavity length or the number of bits used by the pattern generator. The laser repetition rate is a relative parameter that depends on the duty cycle and the scan bandwidth. The scan speed is 194 THz / μs and is an absolute measure of the laser performance.

[0143] The interferometer is made of a reference arm that provides a local oscillator and houses a LiNbO3 electro-optic phase modulator (EO space). The EOM is designed for the 1.3 μm wavelength region and is made of polarization waveguides (no integrated polarizers) with a bandwidth of 10 GHz, an insertion loss of 3 dB, and a π voltage of 5.3 V. For modulations above 30 kHz, a broadband amplifier (MTC5515, Multilink Technologies) is used to amplify the electrical signal. Although the phase modulator is expected to have a π voltage wavelength dependence, there is no significant effect (i.e., suppression) on the orthogonal detection over the scanning bandwidth. In the sample arm, a galvanometer (504 Hz, Sorebo or 4 kHz, EOPC) enables two-dimensional scanning (see below). Imaging is performed using a lens with a focal length of 50 mm, which provides a spot size of 41 μm. The dispersion matching in the sample arm accounts for the waveguide dispersion in the EOM. A 1.6 GHz balanced detector and a 4 GS / s, 12-bit data acquisition board (AlazarTech, ATS9373) are used to acquire the signal.

[0144] result

[0145] Fig.15 Panel (a) shows the laser output after post-amplification in the spectral domain, where the frequency comb structure is clearly visible. Fig.15 Panel (b) shows a zoomed-in view of the spectrum, depicting a frequency comb with a 200 GHz FSR. The etalon FSR can be continuously tuned and demonstrates good performance up to 450 GHz. Fig.15 Panel (c) shows the time trace of a single scan using a 260ps (1 bit) electrical pulse and a 100GHz FSR. A zoomed-in view of the time trace is shown in Fig.15 Panel (d) shows the trace. The trace was recorded using a 35 GHz photodetector (1474-A, New Focus) and a 20 GHz sampling oscilloscope (HP 54120B). The basic structure of the frequency comb is clearly visible in the form of pulsations. Since the CFBG applies a large linear chirp, the pulses directly map the corresponding optical frequencies of the frequency comb in the spectral domain. Fig.15 Panels (c) and (d) also depict a constant pulse interval of 0.52 ns throughout the scan. This is consistent with the f p =v / FSR, where v is the scan speed (i.e., 194 THz / μs). This demonstrates that linear scanning does not require linearization or acquisition clocks. In addition, the measured optical pulse widths are in the range of 300ps to 470ps, which is wider than the 260ps electrical pulse width of the pattern generator. This indicates that the spectrum of the optical pulses is broadened due to gain saturation with a pulse width comparable to the carrier lifetime. Fig.15 A sequence of five scans is shown in panel (e), showing a repetition rate of 7.6 MHz and a duty cycle of 76% at a scan speed of 194 THz / μs. The trace was recorded using a 520 ps electrical pulse on the EOM and a 2 GHz real-time oscilloscope (Tektronix, MSO5204). In this case, no pulsation is observed because the pulse width is similar to the inverse of the pulse repetition rate and the digitizer bandwidth is reduced. Finally, in Fig.15 Panel (f) shows the coherence length measurement after laser post-amplification. It shows a 6 dB signal attenuation at an optical path difference of 4 cm.

[0146] Active, high-speed complex demodulation

[0147] In the following, we describe two active A-line demodulation techniques using LiNbO3 phase modulators. Active phase modulation based on waveguide-based lithium niobate devices provides a straightforward and highly configurable way to modulate the phase and operate comfortably in the GHz range.

[0148] Frame Demodulation

[0149] like Fig.16 As shown in panel (a), inter-frame demodulation of A lines acquires orthogonal components from two frames by phase modulating alternating B scans. Previously, this approach has been challenging due to slow imaging speeds of less than 100 Hz frame rates and the need for high phase stability. The high repetition rate of the SPML laser allows frame rates in the kilohertz range, where environmental fluctuations and vibrations from the microscope are reduced and more manageable. The orthogonal components are extracted from the recorded odd and even B scans to form S(x,z)=S I (x,z)+S Q (x,z) complex signal, where S I (z) = FFT{A I (ω)} and Among them A I (ω), A Q (ω) is the A-line within the I and QB scans, and ω is the angular optical frequency. A resonant scanner was used, with a resonant frequency of 3908.45 Hz (PLD-1S, EOPC). The resonant scanner frequency was slightly detuned from its mechanical resonance to 3908.17 Hz to match an integer of the laser repetition rate for synchronization, providing 1932 A-lines per scan cycle. The phase modulation frequency was adjusted to half the frame rate, f PM =1.95kHz. The sampling rate is adjusted to f s = 3.87 GS / s, which conveniently matches the clock rate of the pattern generator, resulting in 389 points per A line and a Nyquist depth of 1.5 mm, given by z NQ =cf s / (4v) is given. Fig.16 Panels (b) and (c) show the point spread function (PSF) over the baseband range. Using a 150 GHz FSR, the primary measurement range is ±0.5 mm L B =c / (2FSR). Theoretically, considering f s =2vL B / c, which only requires f s = 1.29 GS / s sampling rate. However, in order to additionally demonstrate the suppression of first-order artifacts, the optical Nyquist frequency was intentionally oversampled. Fig.16 Panel (b) shows the original signal after numerical dispersion compensation, while Fig.16 Panel (c) shows the depth signal after I,Q demodulation. For baseband and ±1 frequency comb steps, approximately 40 dB of suppression can be seen. Note that the measured suppression is limited by the system noise floor. Fig.16 Panels (d) and (e)) and tape ( Fig.16Panels (f) and (g)) are imaged to demonstrate complex demodulation. A 100 GHz FSR (L B =1.5mm). Only the base tape area is plotted in the image. The tape sample is highly transparent. Fig.16 The structure seen above the sample surface in panel (g) corresponds to the first frequency comb order showing the sample structure below the sample, and this first frequency comb order is folded back into baseband. Fig.16 The IR card structure seen at the bottom right corner of the image in panel (e) is a sample structure beyond the upper baseband range and corresponds to the -1 frequency comb order. It is worth mentioning that since the scanning function of the resonant scanner is symmetrical (sine wave), in some embodiments, the forward and backward scans can be phase modulated to obtain orthogonal frames, thereby reducing the acquisition time by a factor of two. In this case, the expected scanner phase is critical and needs to be carefully adjusted to ensure frame correlation.

[0150] Phase Correction

[0151] Although the high frame rate of the resonant scanner and the phase stability of the SPML laser greatly reduce the phase noise between frames, small phase fluctuations are still observed, which can lead to visible residual complex conjugation artifacts. Even small amounts of axial motion (e.g., ~3% of the wavelength, such as ~40 nm when using the 1.3 μm imaging band) can cause artifacts. Fig.17 Examples of residual artifacts for the IR card and tape samples are shown in panels (a) and (b) (arrows marked with "cc"). The phase fluctuations are periodic and originate from mechanical microscope instabilities. Care must be taken to construct a robust sample arm by isolating the microscope from vibration sources with frequencies higher than the EOM modulation (half the frame rate). The I,Q component imperfections are due to axial motion and can be corrected.

[0152] Phase difference between composite I frame and Q frame The IR card and tape samples are shown Fig.17 In panels (c) and (d), S* Q' (x,z) is an incorrect Q value frame including phase error. The phase map illustrates a constant phase difference, which confirms the global phase error in the B scan. By plotting a histogram of the phase difference ( Fig.17 Panels (e) and (f)), for example from Fig.17 This can be better understood in the phase difference images shown in panels (e) and (f). The histograms clearly identify the average phase difference, which is higher than the ideal quadrature phase difference of π / 2 for the IR card and lower than the ideal quadrature phase difference of π / 2 in the case of the tape. The shift of the average phase difference relative to the quadrature point represents the global phase error that prevents quadrature detection. During imaging, this phase error can be obtained from the single-sided histogram (positive or negative) of each frame and used as a global phase correction factor

[0153] Thus, to summarize an embodiment of the phase correction scheme: the phase difference between the composite I frame and the Q frame is given by Given, where S I、Q' (z) = FFT{A I、Q' (ω)}, where A I (ω) is the I value A line and A Q' (ω) is the inaccurate Q value A line in the frame. This phase map is used to obtain the global phase error in the B scan (frame). First, a phase histogram is plotted from the phase map. Second, a one-sided histogram (positive or negative side) is used to obtain the average or maximum value of the phase difference. Third, the deviation of the average or maximum phase difference from the ideal orthogonal phase difference (±π / 2) is extracted. The deviation from the orthogonal point represents the global phase error Global phase error as a global phase correction factor is applied to the composite value frame. The corrected artifact-free frame is calculated as: S(x,z)=S I (x,z)+S Q (x,z).

[0154] Since the composite frame is directly available after Fourier transformation, the additional computation time is minimal. Fig.17 This is shown in panels (g) and (h). The highly overlapping structure of the complex conjugate artifacts and the first-order signals of the tape sample represent a particularly challenging scenario. This results in a large standard deviation of the phase in the histogram. Importantly, the phase error is identified and corrected. Fig.17 Panel (i) shows the phase error of 250 frames in a time span of 128 ms before (black line) and after correction (red line) and the corresponding inhibition ( Fig.17 Panel (j). The overlaid bold line shows the average suppression using 5 frames. The vertical dashed blue lines highlight the three points where the phase error exceeds 0.2 radians (3% of the central wavelength). The time-varying phase error produces imperfect I and Q components, which reduces the suppression to 20 dB. Global phase correction reduces the suppression to approximately 40 dB, corresponding to the system noise floor.

[0155] It is important to note that in the case of large phase errors (of the order of π / 2), the detected frames become degraded and cannot be phase corrected. Phase changes observed from microscope instabilities do not exceed 0.4 radians and never approach degeneration. However, sample motion causes large phase errors and leads to frame degradation. Sample motion may also have lateral components that cannot be corrected, and therefore, the correction method described here is more suitable for static or slowly moving samples.

[0156] Imaging

[0157] exist Fig.18 An example of imaging a human nail wrinkle is demonstrated in panel (a). It shows the original dispersion compensated image as well as the artifact-free I and Q demodulated images. No phase correction is required. A 100 GHz FSR is used and only the baseband range, i.e., L B = 1.5 mm. The I,Q demodulated image is stitched on top of itself to make the borderless wrapping of the sample structure beyond the baseband more clear. In addition, Fig.18 Panel (b) shows imaging of an iPhone display. Rendered visualizations of the original and I,Q demodulated images as well as the volumetric image are shown, clearly highlighting the many different layers beneath the surface. Due to a lack of literature, it is difficult to reliably identify the layers of the iPhone 7 display, but they are known to include the polarizing layer, liquid crystal and touch screen, a potential second polarizing layer, a directional diffusion layer, a light guide with backlight, and an end reflector. This also demonstrates the strong applicability of SPML-based frame-by-frame circular ranging demodulation in industrial applications where volumetric video rate and long-distance imaging may be beneficial, including wide field-of-view material, display, or paint inspection.

[0158] It is worth noting that conventional OCT using a continuously scanned laser with an A-line rate of 10 MHz and an imaging range of 4 cm requires a sampling rate on the order of 100 GS / s. We compare this to an FSR of 100 GHz (L B =1.5mm) and a scanning speed of 194THz / μs (i.e., 10MHz A-line rate with Δλ=100nm, full duty cycle), which only requires a sampling rate of 2GS / s because it is independent of the coherence length (i.e., imaging range). This reduces the sampling speed by a factor of 50, and the data loading speed slows down the signal and image processing. Depending on the application, FSR may be increased to 200GHz (L B =750 μm), which may further increase the compression factor to 100. In addition, the 4 kHz scan rate of the resonant scanner can be increased to at least 6 kHz at a duty cycle of 100%, which will further reduce the phase error.

[0159] A-line demodulation

[0160] exist Fig.19 The concept of inter-A line demodulation is shown in panel (a). In this method, a phase modulator in the reference arm induces a phase shift between alternating A lines during a horizontal scan. The odd-numbered A lines and the even-numbered A lines represent the I quadrature component A, respectively. I and Q quadrature components A Q , which are used to form the analytical depth signal, such as A-line demodulation requires high phase stability between A-lines and high A-line rate to avoid phase noise from sample motion. The high phase stability of the SPML laser due to dispersion-based scanning and high repetition rate makes it an excellent choice for A-line demodulation. For A-line spacings much smaller than the beam spot size, the A-lines are correlated and provide valid I, Q components. For large step sizes, the acquired A-lines correspond to different spatial positions and become increasingly uncorrelated, resulting in reduced suppression. In this case, the suppression can be restored by composite A-line interpolation.

[0161] A Galvo scanner (Sorebo) was used, with a fast axis frequency of 504.3 Hz. The frequency was set to an integer multiple of the master clock (external clock of the pattern generator). The phase modulation frequency was adjusted to half the A-line rate, f PM = 3.7MHz. It must be carefully chosen to match an integer multiple of the laser repetition rate for synchronization. The sampling rate is adjusted to f s =3.87GS / s, which conveniently matches the clock rate of the pattern generator, producing 389 points per A line and a Nyquist depth of 1.5mm.

[0162] Fig.19 Panels (b) and (c) show the FSR of 100 GHz (L B =1.5mm) before I, Q demodulation ( Fig.19 Panel (b)) and after ( Fig.19 Panel (c) has the A-line of the PSF in the main measurement range. The 100GHz FSR only requires f s = 1.94 GS / s sampling rate. To additionally demonstrate first order artifact suppression, we intentionally oversampled the optical Nyquist frequency. Note that in this case no transverse beam scanning is applied but complex interpolation. The suppression of the complex conjugate terms is clear and about 40 dB in baseband as well as in ±1st order. The suppression measurement is limited to the system noise floor. Coherent averaging (100 A lines) shows suppression down to -50 dB noise floor. The demodulation is highly stable and permanent, without the need for re-adjustment within days. For demonstration, the suppression is measured continuously for up to one hour without interference ( Fig.19 Panel (d)).

[0163] Composite Average

[0164] For lateral beam steps larger than the beam spot size, the phase modulated A lines become increasingly uncorrelated and produce reduced complex conjugate term suppression, since the IA lines and QA lines correspond to different spatial positions. The lateral beam step can be reduced to one-quarter the beam spot size by complex interpolation (or complex averaging). The Q quadrature component is obtained at its I corresponding spatial position (and vice versa) by interpolating adjacent QA line spatial positions in the depth domain after Fourier transformation.

[0165] Fig. 20 Panel (a) shows the measured suppression of the complex conjugate term as a function of the beam step size Δx / δx normalized to the beam diameter, where δx corresponds to twice the Gaussian beam waist parameter, δx = 4λ0F / (πD), where F is the lens focal length and D is the collimated beam diameter. In this work, the beam step size is adjusted by varying the scan amplitude while maintaining the scan frequency. The green area represents the effective beam step size where adjacent A lines have sufficient correlation (Δx ≤ 1 / 4δx). For step sizes less than one-quarter of the beam spot size, composite interpolation improves the suppression by about 10 dB. The results show that spatial oversampling and composite interpolation can produce suppression better than 35 dB. Note that the 40 dB suppression limit for small step sizes is due to the system noise floor and does not represent a hard limit of the technique. For beam step sizes less than Δx / δx = 0.1, the suppression after interpolation stabilization is about 40 dB, indicating a potential suppression better than 40 dB for systems that offer better noise performance. In general, the application of this technique is limited by the beam step size, which compromises the lateral scanning speed (B-scan rate) or the scanning amplitude (field of view). Fig. 20 Panel (b) shows an example of complex interpolation by imaging the IR card using a beam step size of Δx / δx = 0.23. The original image clearly shows the complex conjugate term (in Fig. 20 The I,Q demodulation provides suppression, but the artifact is still visible. Composite interpolation further improves suppression, reducing the artifact to the -30dB noise floor typically observed during imaging. Finally, the demodulated and corrected images are stitched three times to make the borderless wrapping of sample structure beyond the baseband range more clear.

[0166] Imaging

[0167] exist Fig.21 An imaging example of a human nail wrinkle is illustrated in panel (a). The original dispersion compensated image and the artifact-free I,Q demodulated images are shown. The beam step size is Δx / δx = 0.15. No complex interpolation is applied. A FSR of 100 GHz is used and the image shows only the baseband range, i.e., L B= 1.5 mm. The I,Q demodulated image is stitched on top of itself twice to make the borderless wrapping of the sample structure beyond the baseband clearer. In addition, Fig.21 Panels (b) and (c) show human teeth imaging using a 100 GHz FSR. Fig.21 Visible in panel (b) are the different layers of enamel (1), dentin (2) and gingiva (3). No composite interpolation is required. For illustrative purposes, only the entire acquisition depth range is plotted here, as opposed to only the baseband range seen in all previous images. The baseband is represented by the dashed yellow line. Sample structures that exceed the baseband to positive 1 order (upper yellow line) are folded back to the bottom of the baseband, while sample structures that reach negative 1 order (lower yellow line) are wrapped back to the top of the baseband. Therefore, structures above the enamel surface are the bottom of the tooth folded onto the top of the baseband. The high imaging depth of the tooth suggests the use of a larger baseband (smaller FSR) in order to avoid structure overlap. Finally, SPML-based circular ranging is used to demonstrate the use of a 50mm lens ( Fig.21 Panel (c), left) and 200mm lens ( Figure 1 Panel (c), right) A camera-like device imaging a tooth at 15 volumes per second. As for the frame demodulation technique, it represents a 50-fold compression factor in digitizer bandwidth and data load for signal and image processing compared to a continuously scanned laser with similar performance. Fig.21 The layer imaging shown in panel (b) illustrates the versatility for medical and industrial applications offered by simultaneous high-speed and long-range circular ranging.

[0168] Thus, for the first time, a 1.3 μm CFBG-based SPML laser is demonstrated, making the design usable in a more suitable OCT wavelength region. The A-line rate of the laser is 7.6 MHz and the duty cycle is 76%. The scanning speed is 194 THz / μs, allowing a maximum of 10 MHz at full duty cycle. The scanning range is 100 nm and the coherence length is 4 cm, while the FSR (i.e. the main measurement range) is continuously adjustable. Circular ranging OCT is achieved by complex demodulation using a LiNbO3 phase modulator with A-line phase modulation. Frame demodulation has no speed limit and is in principle limited only by the B-scan rate. Demodulation is demonstrated at 2 kHz using two phase modulated frames from a 4 kHz resonant scanner. Small phase changes due to axial optical path changes in the microscope of the sample arm can be corrected by applying a global phase correction. B-scan rates in the range of 6-10 kHz are required to further minimize this phase noise.

[0169] A-line demodulation is demonstrated as a second inter-A-line demodulation technique that constructs a composite resolved signal from two A-lines and phase modulates it at 3.7 MHz (half the A-line rate). The application of this technique is limited by the use of a beam step size that requires spatial oversampling, which compromises the lateral scan speed (B-scan rate) or the scan amplitude (field of view). For sufficient oversampling, adjacent A-lines are correlated and provide up to 40 dB of suppression, which is only limited by the system floor noise. Larger step sizes sacrifice suppression. In this case, composite interpolation improves suppression by approximately 10 dB for step sizes less than one-quarter of the beam spot size. The active demodulation method presented in this article is very stable, requires minimal or no post-processing, is wavelength independent, and can be performed using a single acquisition channel. Together with the CFBG-based SPML, an embodiment of a compact and stable circular ranging imaging system can be provided.

[0170] Steering Fig. 22 , shows an example 2200 of a system for active quadrature demodulation for circular ranging optical coherence tomography imaging according to some embodiments of the disclosed subject matter. Fig. 22 As shown in , computing device 2210 may receive in-phase data and / or quadrature data from optical interferometry system 2200. In some embodiments, computing device 2210 may execute at least a portion of a system for active quadrature demodulation 2204 to determine a composite interference signal based on the in-phase data and / or quadrature data received from optical interferometry system 2200. Additionally or alternatively, in some embodiments, computing device 2210 may communicate information about the in-phase data and / or quadrature data received from optical interferometry system 2200 via communication network 2206 to server 2220, which may execute at least a portion of a system for active quadrature demodulation 2204 to determine a composite interference signal based on the in-phase data and / or quadrature data. In some such embodiments, server 2220 may return information indicating an output of system for active quadrature demodulation 2204 (such as a composite interference signal) to computing device 2210 (and / or any other suitable computing device). This information may be transmitted and / or presented to a user (eg, a researcher, operator, clinician, etc.) and / or may be stored (eg, as part of a research database or medical record associated with the subject).

[0171] In some embodiments, computing device 2210 and / or server 2220 may be any suitable computing device or combination of devices, such as a desktop computer, a laptop computer, a smart phone, a tablet computer, a wearable computer, a server computer, a virtual machine executed by a physical computing device, etc. As described herein, system 2204 for active quadrature demodulation may present information about in-phase data, quadrature data, and / or a composite interferometric signal to a user (e.g., a researcher and / or a physician).

[0172] In some embodiments, electromagnetic radiation source 2202 may be any source suitable for optical interferometry, such as CR-OCT. In some embodiments, electromagnetic radiation source 2202 may be local to computing device 2210. For example, electromagnetic radiation source 2202 may be combined with computing device 2210 (e.g., computing device 2210 may be configured as part of a device for capturing and / or storing optical interferometry information). As another example, electromagnetic radiation source 2202 may be connected to computing device 2210 via a cable, a direct wireless link, etc. Additionally or alternatively, in some embodiments, electromagnetic radiation source 2202 may be located locally and / or remotely from computing device 2210, and information may be communicated to computing device 2210 (and / or server 2220) via a communication network (e.g., communication network 2206).

[0173] In some embodiments, the communication network 2206 may be any suitable communication network or combination of communication networks. For example, the communication network 2206 may include a Wi-Fi network (a Wi-Fi network may include one or more wireless routers, one or more switches, etc.), a peer-to-peer network (e.g., a Bluetooth network), a cellular network (e.g., a 3G network, a 4G network, etc. that complies with any suitable standard (such as CDMA, GSM, LTE, Advanced LTE, WiMAX, etc.)), a wired network, etc. In some embodiments, the communication network 2206 may be a local area network, a wide area network, a public network (e.g., the Internet), a private or semi-private network (e.g., a company or university intranet), any other suitable type of network, or any suitable combination of networks. Fig. 22 The communication links shown may each be any suitable communication link or combination of communication links, such as a wired link, a fiber optic link, a Wi-Fi link, a Bluetooth link, a cellular link, or the like.

[0174] Fig.23 An example 2300 of hardware that can be used to implement a computing device 2210 and a server 2220 according to some embodiments of the disclosed subject matter is shown. Fig.23As shown, in some embodiments, computing device 2210 may include processor 2302, display 2304, one or more inputs 2306, one or more communication systems 2308, and / or memory 2310. In some embodiments, processor 2302 may be any suitable hardware processor or combination of processors, such as a central processing unit, a graphics processing unit, etc. In some embodiments, display 2304 may include any suitable display device, such as a computer monitor, a touch screen, a television, etc. In some embodiments, input 2306 may include any suitable input device and / or sensor that can be used to receive user input, such as a keyboard, a mouse, a touch screen, a microphone, etc.

[0175] In some embodiments, the communication system 2308 may include any suitable hardware, firmware, and / or software for communicating information over the communication network 2206 and / or any other suitable communication network. For example, the communication system 2308 may include one or more transceivers, one or more communication chips and / or chipsets, etc. In a more specific example, the communication system 2308 may include hardware, firmware, and / or software that can be used to establish a Wi-Fi connection, a Bluetooth connection, a cellular connection, an Ethernet connection, etc.

[0176] In some embodiments, the memory 2310 may include any suitable one or more storage devices that can be used to store instructions, values, etc., which can be used, for example, by the processor 2302 to present content using the display 2304, to communicate with the server 2220 via (multiple) communication systems 2308, etc. The memory 2310 may include any suitable volatile memory, non-volatile memory, storage, or any suitable combination thereof. For example, the memory 2310 may include RAM, ROM, EEPROM, one or more flash drives, one or more hard disks, one or more solid-state drives, one or more optical drives, etc. In some embodiments, the memory 2310 may have encoded thereon a computer program for controlling the operation of the computing device 2210. In such embodiments, the processor 2302 may execute at least a portion of the computer program to present content (e.g., images, user interfaces, graphics, tables, etc.), receive content from the server 2220, transmit information to the server 2220, etc.

[0177] In some embodiments, server 2220 may include a processor 2312, a display 2314, one or more inputs 2316, one or more communication systems 2318, and / or memory 2320. In some embodiments, processor 2312 may be any suitable hardware processor or combination of processors, such as a central processing unit, a graphics processing unit, etc. In some embodiments, display 2314 may include any suitable display device, such as a computer monitor, a touch screen, a television, etc. In some embodiments, input 2316 may include any suitable input device and / or sensor that can be used to receive user input, such as a keyboard, a mouse, a touch screen, a microphone, etc.

[0178] In some embodiments, the communication system 2318 may include any suitable hardware, firmware, and / or software for communicating information over the communication network 2206 and / or any other suitable communication network. For example, the communication system 2318 may include one or more transceivers, one or more communication chips and / or chipsets, etc. In more specific examples, the communication system 2318 may include hardware, firmware, and / or software that can be used to establish a Wi-Fi connection, a Bluetooth connection, a cellular connection, an Ethernet connection, etc.

[0179] In some embodiments, the memory 2320 may include any suitable one or more storage devices that can be used to store instructions, values, etc., which can be used, for example, by the processor 2312 to present content using the display 2314, to communicate with one or more computing devices 2210, etc. The memory 2320 may include any suitable volatile memory, non-volatile memory, storage, or any suitable combination thereof. For example, the memory 2320 may include RAM, ROM, EEPROM, one or more flash drives, one or more hard disks, one or more solid-state drives, one or more optical drives, etc. In some embodiments, the memory 2320 may encode thereon a server program for controlling the operation of the server 2220. In such embodiments, the processor 2312 may execute at least a portion of the server program to transmit information and / or content (e.g., results of organizational identification and / or classification, user interfaces, etc.) to one or more computing devices 2210, receive information and / or content from one or more computing devices 2210, receive instructions from one or more devices (e.g., personal computers, laptops, tablet computers, smart phones, etc.), etc.

[0180] In some embodiments, any suitable computer-readable medium may be used to store instructions for performing the functions and / or processes described herein. For example, in some embodiments, the computer-readable medium may be transient or non-transient. For example, non-transient computer-readable media may include the following media, such as: magnetic media (such as hard disks, floppy disks, etc.), optical media (such as compact disks, digital video disks, Blu-ray disks, etc.), semiconductor media (such as RAM, flash memory, electrically programmable read-only memory (EPROM), electrically erasable programmable read-only memory (EEPROM), etc.), any suitable medium that is not transient or does not have any persistent appearance during transmission, and / or any suitable tangible medium. As another example, transient computer-readable media may include signals on a network, signals in wires, signals in conductors, signals in optical fibers, signals in circuits, or any suitable medium that is transient and does not have any persistent appearance during transmission, and / or any suitable intangible medium.

[0181] In some embodiments, the light signal is detected by a photodiode. It should be appreciated that any alternative electronic conversion device, including but not limited to photodetectors, photodiodes, line scan and two-dimensional cameras, and photodiode arrays, can be used to perform this detection function.

[0182] It should be noted that although the embodiments describe inducing a 90 degree phase shift between measurements to produce a composite signal, it is known that these composite signals can be generated from measurements that are phase shifted by values ​​other than 90 degrees using, for example, correction routines in post-processing. Thus, embodiments can also be configured to generate a non-zero but not necessarily 90 degree phase shift to create a composite signal.

[0183] It should be noted that the term mechanism as used herein may cover hardware, software, firmware, or any suitable combination thereof.

[0184] Fig.24 An example 2400 of a process for active quadrature demodulation for circular ranging optical coherence tomography imaging is shown in accordance with some embodiments of the disclosed subject matter. Fig.24As shown in , at 2402, the process 2400 may scan a sample using an electromagnetic radiation source over a time period. The time period may include a first time period and a second time period, the sample portion of the electromagnetic radiation source may be directed to the sample in the sample arm of the optical interference system, and the reference portion of the electromagnetic radiation source may be directed to the reference arm of the optical interference system. At 2404, the process 2400 may apply a phase shift including a first phase shift and a second phase shift to at least one of the reference portion or the sample portion of the electromagnetic radiation source. The phase shift may be applied using a phase modulator, the first phase shift may be applied during the first time period and the second phase shift may be applied during the second time period, and the second phase shift may differ from the first phase shift by 90 degrees. At 2406, the process 2400 may collect in-phase data based on a first interference between the first backscattered electromagnetic radiation during the first time period and at least one of the reference portion or the sample portion subjected to the first phase shift. At 2408, the process 2400 may collect orthogonal data based on a second interference between the second backscattered electromagnetic radiation during the second time period and at least one of the reference portion or the sample portion subjected to the second phase shift. Finally, at 2410, process 2400 can determine a composite interference signal based on the in-phase data and / or the quadrature data.

[0185] Fig.25 An example 2500 of another process for active quadrature demodulation for circular ranging optical coherence tomography imaging is shown in accordance with some embodiments of the disclosed subject matter. Fig.25As shown in , at 2502, the process 2500 may scan a sample using an electromagnetic radiation source over a time period. The time period may include a first time period and a second time period, the sample portion of the electromagnetic radiation source may be directed to the sample in the sample arm of the optical interference system, and the reference portion of the electromagnetic radiation source may be directed to the reference arm of the optical interference system. At 2504, the process 2500 may apply a phase shift including a first phase shift and a second phase shift to at least one of the reference portion or the sample portion of the electromagnetic radiation source. The phase shift may be applied using a phase modulator, the first phase shift may be applied during the first time period and the second phase shift may be applied during the second time period, and the second phase shift may differ from the first phase shift by 90 degrees. At 2506, the process 2500 may collect in-phase frame data based on a first interference between the first backscattered electromagnetic radiation during the first time period and at least one of the reference portion or the sample portion subjected to the first phase shift. At 2508, the process 2500 may collect orthogonal frame data based on a second interference between the second backscattered electromagnetic radiation during the second time period and at least one of the reference portion or the sample portion subjected to the second phase shift. At 2510, process 2500 may generate a phase difference frame based on the phase difference between the in-phase frame data and the quadrature frame data. At 2512, process 2500 may determine a correction factor based on the phase difference frame. At 2514, process 2500 may apply the correction factor to the in-phase frame data and the quadrature frame data. Finally, at 2516, process 2500 may determine a composite interferometric frame based on the corrected in-phase frame data and the corrected quadrature frame data.

[0186] It should be understood that Fig.24 and Fig.25 The above steps of the process may be performed or carried out in any order or sequence, not limited to the order and sequence shown and described in the figures. Moreover, they may be performed or carried out substantially simultaneously or in parallel where appropriate. Fig.24 or Fig.25 Some of the above steps are performed to reduce waiting time and processing time.

[0187] Therefore, although the invention has been described above in conjunction with particular embodiments and examples, the invention is not necessarily so limited, and many other embodiments, examples, uses, and modifications and deviations from the described embodiments, examples and uses are intended to be encompassed by the appended claims.

Claims

1. A method for optical coherence tomography imaging, comprising: Use an electromagnetic radiation source to scan a sample over a period of time. The time period includes a first time period and a second time period, A sample portion of the electromagnetic radiation source is directed to the sample in a sample arm of an optical interferometer system, and A reference portion of the electromagnetic radiation source is directed to a reference arm of the optical interferometer system; applying a phase shift comprising a first phase shift and a second phase shift to at least one of the reference portion or the sample portion of the electromagnetic radiation source using a phase modulator, applying the first phase shift during the first time period and applying the second phase shift during the second time period, The second phase shift is 90 degrees different from the first phase shift; acquiring in-phase data based on a first interference between first backscattered electromagnetic radiation during the first time period and at least one of the reference portion or the sample portion subjected to the first phase shift, wherein collecting in-phase data further comprises collecting an in-phase frame associated with the first phase shift; acquiring orthogonal data based on a second interference between second backscattered electromagnetic radiation during the second time period and at least one of the reference portion or the sample portion subjected to the second phase shift, wherein collecting quadrature data further comprises collecting a quadrature frame associated with the second phase shift; generating a phase difference frame based on the phase difference between the in-phase frame and the orthogonal frame, determining a correction factor based on the phase difference frame, applying the correction factor to the in-phase frame and the quadrature frame, and A composite interferometric frame is determined based on the corrected in-phase frame and the corrected quadrature frame.

2. The method according to claim 1, characterized in that The first phase shift is 0 degrees during the first time period and the second phase shift is 90 degrees during the second time period.

3. The method according to claim 2, in, Applying the phase shift further comprises: applying said phase shift to said reference portion of said electromagnetic radiation source, Characterized in that collecting in-phase data further comprises: acquiring the in-phase data based on the first interference between the first backscattered electromagnetic radiation during the first time period and the reference portion subjected to the first phase shift, Wherein, collecting orthogonal data further comprises: acquiring the orthogonal data based on the second interference between the second backscattered electromagnetic radiation during the second time period and the reference portion subjected to the second phase shift, and Wherein, determining the composite interference frame further comprises: The composite interference frame is determined based on the in-phase data and the quadrature data.

4. The method according to claim 1, characterized in that The electromagnetic radiation source emits a plurality of electromagnetic radiation pulses, wherein the plurality of electromagnetic radiation pulses comprises a first A-line and a second A-line, the first A-line comprising a first subset of the plurality of electromagnetic radiation pulses emitted during the first time period, The second A-line comprises a second subset of the plurality of electromagnetic radiation pulses emitted during the second time period, and Wherein, scanning the sample further comprises: The sample is scanned using the first A-line during the first time period, and the sample is scanned using the second A-line during the second time period.

5. The method according to claim 4, characterized in that The first subset of the plurality of electromagnetic radiation pulses corresponds to a particular wave number sequence, and Wherein, the second subset of the plurality of electromagnetic radiation pulses corresponds to the specific wave number sequence.

6. The method according to claim 5, characterized in that The first wave number in the specific wave number sequence is different from the second wave number in the specific wave number sequence.

7. The method according to claim 6, characterized in that The first A-line is scanned at a first location in the sample, and the second A-line is scanned at a second location in the sample that is different from the first location.

8. The method according to claim 7, characterized in that the electromagnetic radiation source emitting a third A-line comprising a third subset of the plurality of electromagnetic radiation pulses emitted during a third time period and corresponding to the particular wavenumber sequence, Among them, the scanned samples further include: scanning the sample using the third A-line during the third time period; Wherein applying the phase shift further comprises: applying a 0 degree phase shift to the reference portion of the electromagnetic radiation source during the third time period; Wherein, the in-phase data includes first in-phase data, Wherein, the orthogonal data includes second orthogonal data, Wherein, collecting in-phase data further includes: acquiring third in-phase data based on third interference between third backscattered electromagnetic radiation during the third time period and the reference portion offset by 0 degrees; and Wherein, determining the composite interference frame further comprises: determining second in-phase data based on interpolation between the first in-phase data and the third in-phase data, and The composite interference frame is determined based on the second in-phase data and the second quadrature data.

9. The method according to claim 8, characterized in that The electromagnetic radiation source emits a beam having a beam diameter, Wherein a distance between the first position and the second position is one quarter of the beam diameter or less.

10. The method according to claim 1, characterized in that The electromagnetic radiation source emits a plurality of electromagnetic radiation pulses, and Each of the plurality of electromagnetic radiation pulses is divided into two time periods corresponding to the first time period and the second time period.

11. The method according to claim 10, characterized in that The method further comprises: providing a modified electromagnetic radiation source based on combining the plurality of electromagnetic radiation pulses with delayed copies of the plurality of electromagnetic radiation pulses; and Wherein, scanning the sample further comprises: scanning the sample using the modified electromagnetic radiation source, wherein the modified electromagnetic radiation source alternately emits the plurality of electromagnetic radiation pulses each occurring during the first time period and a corresponding plurality of delayed pulses each occurring during the second time period.

12. The method according to claim 11, characterized in that Providing the modified electromagnetic radiation source comprises: providing the modified electromagnetic radiation source based on combining the plurality of electromagnetic radiation pulses with the delayed copies of the plurality of electromagnetic radiation pulses, wherein the delayed copy of the plurality of electromagnetic radiation pulses is delayed by a time that is shorter than a time of one of the plurality of electromagnetic radiation pulses.

13. The method according to claim 12, characterized in that Determining the composite interference frame further comprises: The composite interferometric frame is determined based on applying a depth-dependent calibration for correcting for additional phase shifts from the detection system.

14. The method according to claim 1, wherein: Determining the correction factor further includes: determining an average phase difference based on the phase difference frame, and The correction factor is determined based on a difference between the average phase difference and a 90 degree phase shift.

15. The method according to claim 14, characterized in that Determining the average phase difference further comprises: determining a histogram of phase differences in the phase difference frame, and The average phase difference is identified based on a histogram of the phase differences.

16. The method according to any one of claims 1 to 15, characterized in that The electromagnetic radiation source comprises an optical sub-sampling wavelength stepping source OSWSS.

17. The method according to any one of claims 1 to 15, characterized in that The electromagnetic radiation source includes a chirped fiber Bragg grating stretched pulse mode-locked CFBG-SPML laser.

18. The method according to claim 17, characterized in that The CFBG-SPML laser includes a 1.3 μm imaging band.

19. The method according to any one of claims 1 to 15, characterized in that The phase modulator comprises a lithium niobate phase modulator.

20. An apparatus for optical coherence tomography imaging, comprising: an optical interferometry system, the optical interferometry system comprising a sample arm and a reference arm; at least one phase modulator associated with at least one of the reference arm or the sample arm of the optical interferometer system; an electromagnetic radiation source coupled to the optical interference system, The electromagnetic radiation source scans the sample over a period of time, The time period includes a first time period and a second time period, A sample portion of the electromagnetic radiation source is directed to the sample in the sample arm of the optical interferometer system, A reference portion of the electromagnetic radiation source is directed to the reference arm of the optical interferometer system, and applying a phase shift comprising a first phase shift and a second phase shift to at least one of the reference portion or the sample portion of the electromagnetic radiation source by the at least one phase modulator, applying the first phase shift during the first time period and applying the second phase shift during the second time period, The second phase shift is 90 degrees different from the first phase shift; as well as a microprocessor coupled to the phase modulator and the electromagnetic radiation source, the microprocessor configured to: acquiring in-phase data based on a first interference between first backscattered electromagnetic radiation during the first time period and at least one of the reference portion or the sample portion subjected to the first phase shift, wherein collecting in-phase data further comprises collecting an in-phase frame associated with the first phase shift, acquiring orthogonal data based on a second interference between second backscattered electromagnetic radiation during the second time period and at least one of the reference portion or the sample portion subjected to the second phase shift, Wherein collecting orthogonal data further comprises an orthogonal frame associated with the second phase shift, generating a phase difference frame based on a phase difference between the in-phase frame and the orthogonal frame, determining a correction factor based on the phase difference frame, applying the correction factor to the in-phase frame and the quadrature frame, and A composite interferometric frame is determined based on the corrected in-phase frame and the corrected quadrature frame.

21. The device according to claim 20, characterized in that The first phase shift is 0 degrees during the first time period and the second phase shift is 90 degrees during the second time period.

22. The device according to claim 21, in, said phase shift being applied to said reference portion of said electromagnetic radiation source by said at least one phase modulator, Wherein, when collecting in-phase data, the microprocessor is further used for: acquiring the in-phase data based on a first interference between the first backscattered electromagnetic radiation during the first time period and the reference portion subjected to the first phase shift, Wherein, when collecting orthogonal data, the microprocessor is further used for: acquiring the orthogonal data based on a second interference between the second backscattered electromagnetic radiation during the second time period and the reference portion subjected to the second phase shift, and Wherein, when determining the composite interference frame, the microprocessor is further used for: The composite interference frame is determined based on the in-phase data and the quadrature data.

23. The device according to claim 20, characterized in that The electromagnetic radiation source emits a plurality of electromagnetic radiation pulses, wherein the plurality of electromagnetic radiation pulses comprises a first A-line comprising a first subset of the plurality of electromagnetic radiation pulses emitted during the first time period and a second A-line comprising a second subset of the plurality of electromagnetic radiation pulses emitted during the second time period, and Wherein, the electromagnetic radiation source is further used for: The sample is scanned using the first A-line during the first time period, and the sample is scanned using the second A-line during the second time period.

24. The device according to claim 23, characterized in that The first subset of the plurality of electromagnetic radiation pulses corresponds to a particular wave number sequence, and Wherein, the second subset of the plurality of electromagnetic radiation pulses corresponds to the specific wave number sequence.

25. The device according to claim 24, characterized in that The first wave number in the specific wave number sequence is different from the second wave number in the specific wave number sequence.

26. The device according to claim 25, characterized in that The first A-line is scanned at a first location in the sample, and the second A-line is scanned at a second location in the sample that is different from the first location.

27. The device according to claim 26, characterized in that the electromagnetic radiation source emitting a third A-line comprising a third subset of the plurality of electromagnetic radiation pulses emitted during a third time period and corresponding to the particular wavenumber sequence, Wherein, the electromagnetic radiation source is further used for: scanning the sample using the third A-line during the third time period; Wherein, the phase modulator is further used for: applying a 0 degree phase shift to the reference portion of the electromagnetic radiation source during the third time period; Wherein, the in-phase data includes first in-phase data, Wherein, the orthogonal data includes second orthogonal data, Wherein, when collecting in-phase data, the microprocessor is further used for: acquiring third in-phase data based on third interference between third backscattered electromagnetic radiation during the third time period and the reference portion offset by 0 degrees; and Wherein, when determining the composite interference frame, the microprocessor is further used for: determining second in-phase data based on interpolation between the first in-phase data and the third in-phase data, and The composite interference frame is determined based on the second in-phase data and the second quadrature data.

28. The device according to claim 27, characterized in that The electromagnetic radiation source emits a beam having a beam diameter, Wherein a distance between the first position and the second position is one quarter of the beam diameter or less.

29. The device according to claim 20, characterized in that The electromagnetic radiation source is further used for: emits multiple pulses of electromagnetic radiation, Each of the plurality of electromagnetic radiation pulses is divided into two time periods corresponding to the first time period and the second time period.

30. The device according to claim 29, characterized in that The microprocessor is further configured to: providing a modified electromagnetic radiation source based on combining the plurality of electromagnetic radiation pulses with delayed copies of the plurality of electromagnetic radiation pulses; and Wherein, the electromagnetic radiation source is further used for: scanning the sample using the modified electromagnetic radiation source, Wherein the modified electromagnetic radiation source alternately emits the plurality of electromagnetic radiation pulses each occurring during a first time and a corresponding plurality of delayed pulses each occurring during a second time.

31. The device according to claim 30, characterized in that In providing the modified electromagnetic radiation source, the microprocessor is further configured to: providing the modified electromagnetic radiation source based on combining the plurality of electromagnetic radiation pulses with the delayed copies of the plurality of electromagnetic radiation pulses, wherein the delayed copy of the plurality of electromagnetic radiation pulses is delayed by a time that is shorter than a time of one of the plurality of electromagnetic radiation pulses.

32. The device according to claim 31, characterized in that When determining the composite interference frame, the microprocessor is further configured to: The composite interferometric frame is determined based on applying a depth-dependent calibration for correcting for additional phase shifts from the detection system.

33. The device according to claim 20, characterized in that When determining the correction factor, the microprocessor is further configured to: determining an average phase difference based on the phase difference frame, and The correction factor is determined based on a difference between the average phase difference and a 90 degree phase shift.

34. The device according to claim 33, characterized in that When determining the average phase difference, the microprocessor is further configured to: determining a histogram of phase differences in the phase difference frame, and The average phase difference is identified based on a histogram of the phase differences.

35. The device according to any one of claims 20 to 34, characterized in that The electromagnetic radiation source comprises an optical sub-sampling wavelength stepping source OSWSS.

36. The device according to any one of claims 20 to 34, characterized in that The electromagnetic radiation source includes a chirped fiber Bragg grating stretched pulse mode-locked CFBG-SPML laser.

37. The device according to claim 36, characterized in that The CFBG-SPML laser includes a 1.3 μm imaging band.

38. The device according to any one of claims 20 to 34, characterized in that The at least one phase modulator comprises a lithium niobate phase modulator.

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