Power conversion device and electric power steering device

Abstract

The power conversion device of the present invention is configured to obtain a first phase current detection value of a first phase corresponding to a phase voltage instruction compared with a first carrier signal from a current detector at timing A, and calculate a first phase current calculation value corresponding to the first phase based on the acquisition result; and to obtain a second phase current detection value of a second phase corresponding to a phase voltage instruction compared with a second carrier signal from a current detector at timing B, and calculate a second phase current calculation value corresponding to the second phase based on the acquisition result.

Description

Technical Field

[0001] The invention relates to a power conversion device and an electric power steering device. Background Art

[0002] A conventional inverter device (see, for example, Patent Document 1) includes an inverter circuit including a high potential side switching element and a low potential side switching element, a controller for controlling the inverter circuit, and a smoothing capacitor for smoothing an output voltage.

[0003] In the inverter device described in Patent Document 1, an evaluation formula for evaluating the ripple size of the input current input to the inverter circuit is used when switching each switching element of the inverter to be turned on and off. In the inverter device, using the evaluation formula, any one of the first carrier waveform and the second carrier waveform is selected as a carrier signal to be compared with a phase voltage command for controlling the phase voltage applied to the three-phase motor.

[0004] Prior art literature

[0005] Patent Literature

[0006] Patent Document 1: Japanese Patent No. 5447136

[0007] Non-patent literature

[0008] Non-patent document 1: The Institute of Electrical Engineers, Research Committee on Sensorless Vector Control, "Sensorless Vector Control of AC Drive Systems", Ohm Co., Ltd., September 25, 2011, p. 70 (see Tables 1 and 3) Summary of the invention

[0009] Technical problem to be solved by the invention

[0010] Here, as a first example, in the above-mentioned inverter device, a current detector is used that is provided with a switching element connected in series to the low potential side of the inverter circuit and detects the phase current flowing through the three-phase motor. In this case, it is necessary to consider the current acquisition timing. The current acquisition timing mentioned here refers to the timing at which the controller acquires the detection value of the current detector.

[0011] That is, in the inverter device described above, the timing of turning on the low potential side switch element needs to be consistent with the current acquisition timing. However, in the inverter device described above, the timing of turning on the low potential side switch element varies depending on the selected carrier waveform, so the current acquisition timing varies.

[0012] As a second example, in the above-mentioned inverter device, a current detector is set to detect the bus current input from the DC power supply to the inverter circuit and a phase current is obtained based on the bus current detected by the current detector. Even in such a case, the current acquisition timing needs to be considered.

[0013] As described above, in the inverter device, although the current acquisition timing should be considered, the current acquisition timing is not specifically mentioned in Patent Document 1. In addition, the accuracy of the value of the phase current obtained by the controller may deteriorate depending on the current acquisition timing.

[0014] The present invention has been made in view of the above circumstances, and an object of the present invention is to provide a power conversion device capable of suppressing degradation in accuracy of a phase current value obtained by a controller, and an electric power steering device having the power conversion device.

[0015] Technical means for solving technical problems

[0016] The power conversion device of the present invention includes: an inverter, which has a high-potential side switch element and a low-potential side switch element set corresponding to each phase of the three phases, and converts the DC voltage output by the DC power supply into an AC voltage and outputs the AC voltage by switching each switch element to on and off; a current detector, which is connected in series with each low-potential side switch element of the inverter, and respectively detects the values ​​of the phase currents of the three phases as phase current detection values; and a controller, which calculates the phase voltage instructions of the three phases and controls the inverter according to the calculated phase voltage instructions of the three phases, and the controller performs the following actions: compares the phase voltage instruction of at least one phase of the three-phase phase voltage instructions with a first carrier signal, and compares the phase voltage instructions of the remaining phases of the three-phase phase voltage instructions Compare with a second carrier signal whose phase differs from the first carrier signal by 1 / 2 of the carrier period of the first carrier signal, thereby switching the switching elements of the inverter to be controlled to be on and off, obtain the first phase current detection value of the phase current detection value of the first phase corresponding to the phase voltage instruction compared with the first carrier signal from the current detector at timing A, and calculate the first phase current calculation value corresponding to the first phase based on the first phase current detection value obtained at the timing, and obtain the second phase current detection value of the phase current detection value of the second phase corresponding to the phase voltage instruction compared with the second carrier signal from the current detector at timing B, and calculate the second phase current calculation value corresponding to the second phase based on the second phase current detection value obtained at the timing.

[0017] The power conversion device of the present invention includes: an inverter, which has a high-potential side switching element and a low-potential side switching element set corresponding to each phase of the three phases, and converts the DC voltage output by the DC power supply into an AC voltage and outputs the AC voltage by switching each switching element to on and off; a current detector, which detects the value of the bus current input from the DC power supply to the inverter as the bus current detection value; and a controller, which calculates the phase voltage instructions of the three phases and controls the inverter according to the calculated phase voltage instructions of the three phases. When the controller sets the phase voltage instructions of the three phases to the maximum phase voltage instruction, the middle phase voltage instruction and the minimum phase voltage instruction in the order from large to small, the first voltage shifting method of equally shifting the phase voltage instructions of the three phases so that the maximum phase voltage instruction is consistent with the maximum value of the first carrier signal is applied, or the phase voltage instructions of the three phases are equally shifted so that the minimum phase voltage instruction is consistent with the maximum value of the first carrier signal is applied. A second voltage shifting method consistent with the minimum value of the first carrier signal, compares the phase voltage command of at least one phase of the shifted three-phase phase voltage commands with the first carrier signal, and compares the phase voltage commands of the remaining phases of the shifted three-phase phase voltage commands with a second carrier signal whose phase differs from the first carrier signal by 1 / 2 of the carrier period of the first carrier signal, thereby switching the switching elements of the inverter to be turned on and off, obtains the bus current detection value from the current detector at each timing of timing A and timing B, and based on the bus current detection value obtained at each timing, calculates the first phase current calculation value corresponding to the phase corresponding to the phase voltage command compared with the first carrier signal, that is, the first phase, and the second phase current calculation value corresponding to the phase corresponding to the phase voltage command compared with the second carrier signal, that is, the second phase, wherein timing A and timing B are different timings and are timings at which the bus current flows from the DC power supply to the inverter.

[0018] An electric power steering device according to the present invention includes the above-mentioned power conversion device.

[0019] Effects of the Invention

[0020] According to the present invention, it is possible to obtain a power conversion device capable of suppressing degradation in accuracy of a phase current value obtained by a controller, and an electric power steering device having the power conversion device. BRIEF DESCRIPTION OF THE DRAWINGS

[0021] Figure 1 It is a diagram showing the overall configuration of a power conversion device according to Embodiment 1 of the present invention.

[0022] Figure 2 1 is a timing chart showing a first example of the time change of the phase current detection values ​​of the U-phase, the V-phase, and the W-phase in the first embodiment of the present invention.

[0023] Figure 31 is a timing chart showing a second example of the time change of the phase current detection values ​​of the U-phase, V-phase, and W-phase in the first embodiment of the present invention.

[0024] Figure 4 1 is a timing chart showing a third example of the time change of the phase current detection values ​​of the U-phase, V-phase, and W-phase in the first embodiment of the present invention.

[0025] Figure 5 It is a diagram showing the overall configuration of a power conversion device according to Embodiment 2 of the present invention.

[0026] Figure 6 This is a table showing voltage vectors output according to the pattern of the on / off signal in the second embodiment of the present invention.

[0027] Figure 7 Yes means Figure 6 Graph of the voltage vector.

[0028] Figure 8 This is a timing chart showing a first example of a temporal change in a bus current detection value in Embodiment 2 of the present invention.

[0029] Fig. 9 This is a timing chart showing a second example of the temporal change of the bus current detection value in the second embodiment of the present invention.

[0030] Fig.10 This is a diagram showing current vectors in Embodiment 2 of the present invention.

[0031] Fig.11 This is a table showing a voltage shift method selected according to the current phase of the current vector in the second embodiment of the present invention.

[0032] Fig.12 It is a diagram showing the overall configuration of a power conversion device according to Embodiment 3 of the present invention.

[0033] Fig.13 This is a diagram showing the structure of a position estimator according to Embodiment 3 of the present invention.

[0034] Fig.14 It is a diagram showing the overall structure of an electric power steering device according to a fourth embodiment of the present invention. DETAILED DESCRIPTION

[0035] Hereinafter, the power conversion device and the electric power steering device of the present invention will be described based on preferred embodiments using the accompanying drawings. In the description of the accompanying drawings, the same reference numerals are given to the same or corresponding parts, and repeated descriptions are omitted.

[0036] Implementation Method 1

[0037] Figure 1: is a diagram showing the overall structure of a power conversion device according to Embodiment 1 of the present invention. Figure 1 Also shown are a DC power source 3 connected to the input side of the power conversion device and an AC rotating electric machine 1 connected to the output side thereof.

[0038] like Figure 1 As shown, the power conversion device in the first embodiment includes a smoothing capacitor 4 , an inverter 6 , a current detector 7 , and a controller 8 .

[0039] The AC rotating electric machine 1 is a three-phase AC electric motor having a three-phase winding composed of a U-phase winding U, a V-phase winding V, and a W-phase winding W. Specific examples of the AC rotating electric machine 1 include a permanent magnet synchronous motor, an induction motor, a synchronous reluctance motor, etc. However, as the AC rotating electric machine 1, any type of motor can be used as long as it is a motor having a three-phase winding. Here, a case where a non-salient pole type permanent magnet synchronous motor is used as the AC rotating electric machine 1 is exemplified.

[0040] The position detector 2 is provided in the AC rotating electric machine 1 . The position detector 2 detects a rotation position θ of the AC rotating electric machine 1 , and outputs the detected rotation position θ to the controller 8 .

[0041] The DC power source 3 has a high potential side terminal and a low potential side terminal, and outputs a DC voltage Vdc as a voltage between the two terminals to the inverter 6. The DC power source 3 includes all devices that output a DC voltage, such as a battery, a DC-DC converter, a diode rectifier, and a PWM (Pulse Width Modulation) rectifier.

[0042] The smoothing capacitor 4 is provided in parallel between the DC power source 3 and the inverter 6. The smoothing capacitor 4 is connected in parallel with the DC power source 3 to suppress the fluctuation of the DC voltage Vdc input to the inverter 6 and realize a stable DC voltage. The current Ic flows through the smoothing capacitor 4.

[0043] Inductor 5 indicates the inductance value included in the DC power supply 3 and in the wiring from the DC power supply 3 to the smoothing capacitor 4. Generally, in a power conversion device, a common mode choke coil is connected near the DC power supply 3 as a noise filter to suppress noise flowing from the inverter 6 to the DC power supply 3. The inductance value of such a noise filter is also included in the inductor 5. Current Ib flows through the inductor 5.

[0044] The inverter 6 has high potential side switching elements Sup~Swp and low potential side switching elements Sun~Swn provided corresponding to each phase of the three phases of U phase, V phase and W phase. Specifically, the inverter 6 is a three-phase inverter composed of high potential side switching elements Sup~Swp and low potential side switching elements Sun~Swn. The inverter 6 switches each switching element Sup~Swn to on and off based on the on / off signals Qup~Qwn from the controller 8. Through such switching, the inverter 6 converts the DC voltage Vdc output by the DC power supply 3 into an AC voltage and outputs the AC voltage. Therefore, the AC voltage is applied to the windings U~W of the AC rotating motor 1, and the phase currents Iu~Iw are energized to the windings U~W respectively.

[0045] Here, the on / off signals Qup~Qwn are switch signals for switching the switch elements Sup~Swn to on and off, respectively. In the following, it is assumed that when the value of the on / off signals Qup~Qwn is "1", a signal for turning on the switch element corresponding to the on / off signal is output, and when the value of the signal is "0", a signal for turning off the switch element corresponding to the on / off signal is output.

[0046] As the switching elements Sup to Swn, a semiconductor switch such as an IGBT (Insulated Gate Bipolar Transistor), a bipolar transistor, or a MOS (Metal-Oxide-Semiconductor) power transistor and a diode connected in antiparallel are used.

[0047] The current detector 7 is connected in series with the switching elements Sun to Swn on the low potential side of the inverter 6. The current detector 7 detects the values ​​of the phase currents Iu to Iw of the three phases as phase current detection values ​​Ius to Iws, respectively.

[0048] Specifically, the phase current detection value Ius is detected by detecting the voltage across the resistor Ru inside the current detector 7. Similarly, the phase current detection value Ivs is detected by detecting the voltage across the resistor Rv inside the current detector 7. Similarly, the phase current detection value Iws is detected by detecting the voltage across the resistor Rw inside the current detector 7.

[0049] The current commands Id_target and Iq_target set as control commands for controlling the AC rotating electric machine 1 are input to the controller 8. Here, a case where the current command Id_target=0 is set is exemplified.

[0050] The controller 8 calculates the three-phase phase voltage instructions Vu~Vw, and controls the inverter 6 according to the calculated three-phase phase voltage instructions Vu~Vw. Specifically, the controller 8 outputs the on-off signals Qup~Qwn based on the input current instructions Id_target and Iq_target, the rotation position θ input from the position detector 2, and the phase current calculation values ​​Ius_cal~Iws_cal. In addition, the details of the phase current calculation values ​​Ius_cal~Iws_cal will be described later.

[0051] In addition, the case where the command value of the current supplied to the AC rotating electric machine 1 is set as the control command of the AC rotating electric machine 1 is exemplified, but the present invention is not limited thereto. For example, when V / F control is performed on the AC rotating electric machine 1, the control command becomes the speed command value of the AC rotating electric machine 1. When the rotation position of the AC rotating electric machine 1 is controlled, the control command becomes the position command value of the AC rotating electric machine 1.

[0052] Next, each element constituting the controller 8 will be described. The controller 8 includes a coordinate converter 9 , a subtractor 10 , a subtractor 11 , a current controller 12 , a current controller 13 , a coordinate converter 14 , an on / off signal generator 15 , and a current acquisition unit 16 .

[0053] The coordinate converter 9 calculates the d-axis current Id and the q-axis current Iq on two rotation axes, i.e., the dq axes, based on the phase current calculation values ​​Ius_cal to Iws_cal input from the current acquisition unit 16 and the rotation position θ input from the position detector 2. The coordinate converter 9 outputs the calculated current Id to the subtractor 10 and outputs the calculated current Iq to the subtractor 11.

[0054] The subtractor 10 subtracts the current Id input from the coordinate converter 9 from the current command Id_target, and outputs the result to the current controller 12 .

[0055] The subtractor 11 subtracts the current Iq input from the coordinate converter 9 from the current command Iq_target, and outputs the result to the current controller 13 .

[0056] The current controller 12 performs proportional and integral control on the output value of the subtractor 10 so that the value becomes zero, thereby calculating the d-axis voltage Vd on the rotational two axes, and outputs the calculated voltage Vd to the coordinate converter 14 .

[0057] The current controller 13 performs proportional and integral control on the output value of the subtractor 11 so that the value becomes zero, thereby calculating the q-axis voltage Vq on the two rotating axes, and outputs the calculated voltage Vq to the coordinate converter 14 .

[0058] The coordinate converter 14 calculates three-phase phase voltage commands Vu to Vw based on the voltages Vd and Vq input from the current controller 12 and the current controller 13 and the rotational position θ input from the position detector 2 , and outputs the results to the on / off signal generator 15 .

[0059] Next, refer to Figure 2 The respective operations of the on / off signal generator 15 and the current acquisition unit 16 will be described. Figure 2 1 is a timing chart showing a first example of time changes in the phase current detection values ​​Ius, Ivs, and Iws of the U-phase, V-phase, and W-phase, respectively, in the first embodiment of the present invention.

[0060] First, the operation of the on / off signal generator 15 will be described. Figure 2 As shown, the on / off signal generator 15 compares the phase voltage command Vu input from the coordinate converter 14 with the first carrier signal. The period of the first carrier signal is period Tc. The maximum value of the first carrier signal is located at the peak value of the amplitude of the first carrier signal. The minimum value of the first carrier signal is located at the valley value of the amplitude of the first carrier signal.

[0061] As a result of the comparison, when the phase voltage command Vu is greater than the first carrier signal, the on / off signal generator 15 outputs the on / off signal Qup having a value of "1" and outputs the on / off signal Qun having a value of "0".

[0062] On the other hand, when the phase voltage command Vu is smaller than the first carrier signal, the on / off signal generator 15 outputs an on / off signal Qup with a value of "0" and outputs an on / off signal Qun with a value of "1". Figure 2 In FIG. 1 , the waveform of the on / off signal Qun is shown, but since the relationship Qup+Qun=1 exists, the waveform of the on / off signal Qup is omitted.

[0063] The on / off signal generator 15 compares the phase voltage command Vw input from the coordinate converter 14 with the second carrier signal. The second carrier signal has the same shape as the first carrier signal, and the phase is different from the first carrier signal by 1 / 2 period of period Tc. In addition, the second carrier signal has the same period Tc as the first carrier signal.

[0064] As a result of the comparison, when the phase voltage command Vw is greater than the second carrier signal, the on / off signal generator 15 outputs the on / off signal Qwp having a value of "1" and outputs the on / off signal Qwn having a value of "0".

[0065] On the other hand, when the phase voltage command Vw is smaller than the second carrier signal, the on / off signal generator 15 outputs an on / off signal Qwp with a value of "0" and outputs an on / off signal Qwn with a value of "1". Figure 2 In FIG. 1 , the waveform of the on / off signal Qwn is shown, but since the relationship Qwp+Qwn=1 exists, the waveform of the on / off signal Qwp is omitted.

[0066] The on / off signal generator 15 compares the phase voltage command Vv input from the coordinate converter 14 with any one of the first carrier signal and the second carrier signal. As a result of the comparison, when the phase voltage command Vv is greater than the carrier signal, the on / off signal generator 15 outputs an on / off signal Qvp with a value of "1" and outputs an on / off signal Qvn with a value of "0".

[0067] Here, if Figure 2 As shown, the phase voltage command Vv is consistent with the maximum value of the first carrier signal. Therefore, even if the phase voltage command Vv is compared with any one of the first carrier signal and the second carrier signal, the on / off signal generator 15 outputs an on / off signal Qvp with a value of "1" and outputs an on / off signal Qvn with a value of "0".

[0068] As described above, there is no difference even if the on / off signal generator 15 compares the phase voltage command Vv with either the first carrier signal or the second carrier signal. However, as a specific example, the on / off signal generator 15 compares the phase voltage command Vv with the first carrier signal. Figure 2 In FIG. 1 , the waveform of the on / off signal Qvn is shown, but since the relationship Qvp+Qvn=1 exists, the waveform of the on / off signal Qvp is omitted.

[0069] As described above, the on / off signal generator 15 compares the phase voltage command of at least one phase among the three-phase phase voltage commands Vu to Vw with the first carrier signal, and compares the phase voltage commands of the remaining phases among the three-phase phase voltage commands Vu to Vw with the second carrier signal. The on / off signal generator 15 switches the switching elements Sup to Swn of the inverter 6 between on and off through such comparison.

[0070] Next, the operation of the current acquisition unit 16 will be described. The current acquisition unit 16 acquires the first phase current detection value from the current detector 7 at each of timing A and timing B as the first phase current detection value corresponding to the phase voltage command to be compared with the first carrier signal.

[0071] Furthermore, the current acquisition unit 16 acquires the second phase current detection value from the current detector 7 at each of the timing A and the timing B as the phase current detection value of the second phase corresponding to the phase voltage command to be compared with the second carrier signal.

[0072] Here, the timing A is the timing when the phase current of the first phase flows through the resistor corresponding to the first phase constituting the current detector 7, and more specifically, for example, the timing when the low-potential side switch element corresponding to the first phase becomes conductive when the phase voltage command of the first phase is zero. In addition, the timing B is the timing when the phase current of the second phase flows through the resistor corresponding to the second phase constituting the current detector 7, and more specifically, for example, the timing when the phase voltage command of the second phase is zero, and the low-potential side switch element corresponding to the second phase becomes conductive. The timing B is offset from the timing A by, for example, 1 / 2 of the period Tc.

[0073] In addition, in embodiment 1, if Figure 2 As shown, the case where the phase voltage command compared with the first carrier signal is the phase voltage command Vu of the U phase, and the phase voltage command compared with the second carrier signal is the phase voltage command Vw of the W phase is illustrated. In this case, the first phase becomes the U phase, and the first phase current detection value becomes the phase current detection value Ius of the U phase. In addition, the second phase becomes the W phase, and the second phase current detection value becomes the phase current detection value Iws of the W phase.

[0074] In addition, in Embodiment 1, if Figure 2 As shown, a case where the first carrier signal becomes a maximum value at timing A and becomes a minimum value at timing B is illustrated.

[0075] The following describes a method for calculating the phase current calculation values ​​Ius_cal to Iws_cal by the current acquisition unit 16. In the following, for convenience, the phase current detection value Ius obtained by the current acquisition unit 16 from the current detector 7 at timing A is recorded as the phase current detection value Ius(A), and the phase current detection value Ius obtained by the current acquisition unit 16 from the current detector 7 at timing B is recorded as the phase current detection value Ius(B). The phase current detection value Ivs and the phase current detection value Iws are also set to the same symbol as the phase current detection value Ius.

[0076] First, a method of calculating the first phase current calculated value corresponding to the first phase, that is, the U-phase phase current calculated value Ius_cal will be described. Figure 2 As shown in FIG. 1 , the phase current detection value Ius(B) is equivalent to the offset current value Iu_offset. The reason is as follows.

[0077] That is, at timing B, since the value of the on / off signal Qun is "0", the switch element Sun corresponding to the U phase becomes off. Therefore, the phase current Iu does not flow through the resistor Ru of the current detector 7, and the phase current detection value Ius(B) should be 0. However, in reality, the phase current detection value Ius(B) does not become 0, and the phase current detection value Ius(B) is a value other than 0. This value can be determined as an offset caused by the temperature drift of the current detector 7, that is, the offset current value Iu_offset. That is, the phase current detection value Ius(B) is equal to the offset current value Iu_offset.

[0078] At timing A, since the value of the on / off signal Qun is "1", the switch element Sun corresponding to the U phase becomes on. Therefore, the phase current Iu flows through the resistor Ru of the current detector 7, and the phase current detection value Ius (A) becomes a value other than 0. The phase current detection value Ius (A) is equal to the sum of the value of the phase current Iu and the offset current value Iu_offset.

[0079] Thus, the current acquisition unit 16 calculates the first phase current calculated value based on the first phase current detection value acquired at each of the timings A and B. That is, the current acquisition unit 16 calculates the phase current calculated value Ius_cal based on the phase current detection value Ius(A) and the phase current detection value Ius(B) acquired at each of the timings A and B.

[0080] Specifically, the current acquisition unit 16 calculates the first phase current calculation value by subtracting the offset current value equivalent to the first phase current detection value acquired at timing B from the first phase current detection value acquired at timing A. That is, the current acquisition unit 16 calculates the phase current calculation value Ius_cal by subtracting the offset current value Ius_offset equivalent to the phase current detection value Ius(B) acquired at timing B from the phase current detection value Ius(A) acquired at timing A. Thus, by calculating the phase current calculation value Ius_cal using the two phase current detection values ​​Ius(A) and Ius(B) acquired at two current acquisition timings, the influence of the offset current value Iu_offset can be eliminated.

[0081] In addition, here, the case where the phase current detection value Ius(B) is directly used to calculate the phase current calculation value Ius_cal is exemplified, but the present invention is not limited to this. That is, considering the case where the time constant of the temperature drift described above is greater than the period Tc, in order to reduce the error caused by noise, the phase current calculation value Ius_cal may be calculated using the phase current detection value Ius(B) that has been processed by moving average processing, filtering processing, etc.

[0082] In the above, an example is given in which the first phase is the U phase. However, even if the first phase is another phase, the above method is also applicable, and a phase current calculation value corresponding to the first phase can be calculated.

[0083] Next, a method of calculating the second phase current calculated value corresponding to the second phase, that is, the W phase current calculated value Iws_cal will be described. Figure 2 As shown in FIG. 1 , the phase current detection value Iws (A) is equivalent to the offset current value Iw_offset. The reason is as follows.

[0084] That is, at timing A, since the value of the on / off signal Qwn is "0", the switch element Swn corresponding to the W phase becomes off. Therefore, the phase current Iw does not flow through the resistor Rw of the current detector 7, and the phase current detection value Iws (A) should be 0. However, in reality, the phase current detection value Iws (A) does not become 0, and the phase current detection value Iws (A) is a value other than 0. This value can be determined as an offset caused by the temperature drift of the current detector 7, that is, the offset current value Iw_offset. That is, the phase current detection value Iws (A) is equal to the offset current value Iw_offset.

[0085] Next, at timing B, since the value of the on / off signal Qwn is "1", the switch element Swn corresponding to the W phase becomes on. Therefore, the phase current Iw flows through the resistor Rw of the current detector 7, and the phase current detection value Iws(B) becomes a value other than 0. The phase current detection value Iws(B) is equal to the sum of the value of the phase current Iw and the offset current value Iw_offset.

[0086] Thus, the current acquisition unit 16 calculates the second phase current calculated value based on the second phase current detection value obtained at each timing A and timing B. That is, the current acquisition unit 16 calculates the phase current calculated value Iws_cal based on the phase current detection value Iws(A) and the phase current detection value Iws(B) obtained at each timing A and timing B.

[0087] Specifically, the current acquisition unit 16 calculates the second phase current calculation value by subtracting the offset current value equivalent to the second phase current detection value acquired at timing A from the second phase current detection value acquired at timing B. That is, the current acquisition unit 16 calculates the phase current calculation value Iws_cal by subtracting the offset current value Iw_offset equivalent to the phase current detection value Iws(A) acquired at timing A from the phase current detection value Iws(B) acquired at timing B. Thus, the phase current calculation value Iws_cal with the influence of the offset current value Iw_offset eliminated can be obtained.

[0088] In the above, an example is given in which the second phase is the W phase. However, even if the second phase is another phase, the above method is also applicable, and the phase current calculation value corresponding to the second phase can be calculated.

[0089] The controller 8 calculates the phase voltage commands Vu to Vw using the first phase current calculated value and the second phase current calculated value calculated by the above-described method, that is, the phase current calculated values ​​Ius_cal and Iws_cal.

[0090] Specifically, the controller 8 calculates the phase voltage commands Vu to Vw using the phase current calculated values ​​Ius_cal and Iws_cal and the phase current calculated value Ivs_cal calculated from the phase current calculated values ​​Ius_cal and Iws_cal by the following method.

[0091] The method of calculating the phase current calculation value Ivs_cal of the V phase is as follows. That is, the current acquisition unit 16 calculates the phase current calculation value Ivs_cal using the relationship that the sum of the phase currents Iu to Iw is 0, that is, Iu+Iv+Iw=0. Specifically, the current acquisition unit 16 uses the phase current calculation values ​​Ius_cal and Iws_cal to calculate the phase current calculation value Ivs_cal according to the relationship Ivs_cal=-Ius_cal-Iws_cal.

[0092] In addition, the controller 8 may be configured to perform calculations of the three-phase phase voltage commands Vu to Vw in a cycle that is two or more natural number times the cycle Tc. Figure 3 and Figure 4 The structure is described below.

[0093] Figure 3 1 is a timing chart showing a second example of the time change of the phase current detection values ​​Ius, Ivs, and Iws of the U-phase, V-phase, and W-phase, respectively, in the first embodiment of the present invention. Figure 4 1 is a timing chart showing a third example of the time changes of the phase current detection values ​​Ius, Ivs, and Iws of the U-phase, V-phase, and W-phase, respectively, in the first embodiment of the present invention.

[0094] like Figure 3 As shown, when comparing cycle 1 and cycle 2, in cycle 2, the phase voltage command Vw of the V phase and the phase voltage command Vw of the W phase are exchanged with respect to cycle 1. That is, in Figure 3 In the embodiment, the calculation of the phase voltage commands Vu to Vw is performed at a period Tc.

[0095] like Figure 3As shown, at timing B, the value of the on / off signal Qwn is switched from "1" to "0", and the value of the on / off signal Qvn is switched from "0" to "1". Therefore, in the inverter 6, the switching elements Svp and Svn corresponding to the V phase and the switching elements Swp and Swn corresponding to the W phase are switched.

[0096] In this case, the noise component originally caused by the above-mentioned switching is superimposed on the current flowing through the AC rotating electric machine 1, so the accuracy of the phase current detection values ​​Ius to Iws detected by the current detector 7 deteriorates.

[0097] Therefore, if Figure 4 As shown, the cycle of calculating and updating the voltage command, that is, the voltage command update cycle, is set equal to the sum of cycle 1 and cycle 2. Therefore, similarly to timing A, at timing B, a current detection value with high accuracy can be obtained, and a high accuracy offset current value can be obtained.

[0098] Above, in Figure 4 In FIG. 1 , the case where the phase voltage commands Vu to Vw are calculated and updated at a period twice the period Tc is exemplified, but the phase voltage commands Vu to Vw may be calculated and updated at a period that is a natural number multiple of three or more times the period Tc.

[0099] In addition, when calculating the first-phase current calculation value, the current acquisition unit 16 may use the average value of the first-phase current detection value acquired at the latest timing A and the first-phase current detection value acquired at the last timing A before the latest timing A. In this case, the current acquisition unit 16 is configured to calculate the first-phase current calculation value by subtracting the offset current value from the average value. Similarly, when calculating the second-phase current calculation value, the current acquisition unit 16 may use the average value of the second-phase current detection value acquired at the latest timing B and the second-phase current detection value acquired at the last timing B before the latest timing B. In this case, the current acquisition unit 16 is configured to calculate the second-phase current calculation value by subtracting the offset current from the average value.

[0100] The above configuration makes it possible to obtain a phase current calculation value that is more resistant to noise, and as a result, it is possible to obtain an effect of reducing torque pulsation, vibration, and noise of the AC rotating electric machine 1 that are caused and generated by noise.

[0101] As described above, according to the present first embodiment, the controller 8 of the power conversion device is configured to acquire the first phase current detection value of the first phase corresponding to the phase voltage command compared with the first carrier signal from the current detector 7 at timing A, and calculate the first phase current calculation value corresponding to the first phase based on the acquisition result. In addition, the controller 8 is configured to acquire the second phase current detection value of the second phase corresponding to the phase voltage command compared with the second carrier signal from the current detector 7 at timing B, and calculate the second phase current calculation value corresponding to the second phase based on the acquisition result.

[0102] Thus, even when any one of the first carrier signal and the second carrier signal is used for comparison with the phase voltage command, the two phase current detection values ​​respectively acquired at the two current acquisition timings can be used to obtain the phase current calculation value after removing the offset current value. That is, a power conversion device can be obtained that can suppress the accuracy degradation of the value of the phase current acquired by the controller 8.

[0103] In addition, when the error required by the current acquisition unit 16 is smaller than the temperature change or annual change of the offset current value of the power conversion device, the calculation of the offset current value is not necessary. In this case, the controller 8 is configured as follows. That is, the controller 8 is configured to obtain the first phase current detection value from the current detector 7 at timing A, and calculate the first phase current calculation value based on the first phase current detection value obtained at timing A. Specifically, the controller 8 sets the first phase current detection value obtained at timing A as the first phase current calculation value. In addition, the controller 8 is configured to obtain the second phase current detection value from the current detector 7 at timing B, and calculate the second phase current calculation value based on the second phase current detection value obtained at timing B. Specifically, the controller 8 obtains the second phase current detection value from the current detector 7 at timing B, and sets the second phase current detection value obtained at timing B as the second phase current calculation value.

[0104] Implementation Method 2

[0105] In Embodiment 2 of the present invention, a power conversion device having a different structure from that of Embodiment 1 is described. In Embodiment 2, description of the same points as Embodiment 1 is omitted, and description is mainly focused on the points different from Embodiment 1.

[0106] Figure 5 2 is a diagram showing the overall structure of a power conversion device according to Embodiment 2 of the present invention. Figure 5 As shown, the power conversion device in the second embodiment includes a smoothing capacitor 4 , an inverter 6 , a current detector 7 , and a controller 8 .

[0107] The current detector 7 has a different structure from the previous embodiment 1, and detects the value of the bus current Iin input from the DC power supply 3 to the inverter 6 as the bus current detection value Iins. Specifically, the bus current detection value Iins is detected by detecting the voltage across the resistor Rdc inside the current detector 7. The current detector 7 outputs the detected bus current detection value Iins to the current acquisition unit 16.

[0108] Figure 6 This is a table showing voltage vectors output according to the pattern of the on / off signals Qup to Qwn in the second embodiment of the present invention. Figure 7 It is shown Figure 6 Diagram of voltage vectors V0~V7.

[0109] The on / off signal generator 15 outputs Figure 6 The voltage vector corresponding to the mode of the on-off signals Qup~Qwn shown in FIG. Figure 6 As shown in FIG. 1 , the voltage vector is determined according to the mode of the on-off signals Qup to Qwn. Figure 6 The voltage vectors V0 to V7 are shown in the figure, then Figure 7 .like Figure 7 As shown, the voltage vectors V1 to V6 are effective voltage vectors having magnitudes every 60-degree phase difference, and the voltage vectors V0 and V7 are zero voltage vectors having no magnitudes.

[0110] like Figure 6 As shown, the current acquisition unit 16 obtains a phase current calculated value equivalent to the bus current detection value Iins acquired from the current detector 7 based on the pattern of the on / off signals Qup to Qwn.

[0111] For example, Figure 6 As shown, when the pattern of the on / off signals Qup to Qwn is Qup=1, Qun=0, Qvp=0, Qvn=1, Qwp=0, Qwn=1, the voltage vector V1 is output. In this case, the current acquisition unit 16 calculates the phase current calculation value Ius_cal equivalent to the bus current detection value Iins. That is, the current acquisition unit 16 determines that the bus current detection value Iins acquired at the timing of outputting the voltage vector V1 is equivalent to the phase current calculation value Ius_cal. When the pattern of the on / off signals Qup to Qwn is Figure 6 The same can also be considered when another mode is shown.

[0112] Next, refer to Figure 8 The respective operations of the on / off signal generator 15 and the current acquisition unit 16 will be described. Figure 8 1 is a timing chart showing a first example of a temporal change in the bus current detection value Iins in the second embodiment of the present invention.

[0113] Hereinafter, the phase voltage commands Vu to Vw are set in descending order of value as the maximum phase voltage command, the middle phase voltage command, and the minimum phase voltage command. Figure 8 In the example shown, the phase voltage command Vv is the maximum phase voltage command, the phase voltage command Vu is the intermediate phase voltage command, and the phase voltage command Vw is the minimum phase voltage command.

[0114] The on / off signal generator 15 applies a first voltage shifting method, which shifts the three-phase phase voltage commands Vu to Vw equally so that the maximum phase voltage command is consistent with the maximum value of the first carrier signal. Figure 8 As shown, the phase voltage commands Vu to Vw are equally shifted so that the phase voltage command Vv, which is the maximum phase voltage command, coincides with the maximum value of the first carrier signal.

[0115] like Figure 8 As shown, the on / off signal generator 15 compares the phase voltage command Vu with the first carrier signal. As a result of the comparison, when the phase voltage command Vu is greater than the first carrier signal, the on / off signal generator 15 outputs an on / off signal Qup with a value of "1" and outputs an on / off signal Qun with a value of "0".

[0116] On the other hand, when the phase voltage command Vu is smaller than the first carrier signal, the on / off signal generator 15 outputs an on / off signal Qup with a value of "0" and outputs an on / off signal Qun with a value of "1". Figure 8 In FIG. 1 , the waveform of the on / off signal Qup is shown, but since the relationship Qup+Qun=1 exists, the waveform of the on / off signal Qun is omitted.

[0117] The on / off signal generator 15 compares the phase voltage command Vw with the second carrier signal. As a result of the comparison, when the phase voltage command Vw is greater than the second carrier signal, the on / off signal generator 15 outputs an on / off signal Qwp of "1" and outputs an on / off signal Qwn of "0".

[0118] On the other hand, when the phase voltage command Vw is smaller than the second carrier signal, the on / off signal generator 15 outputs an on / off signal Qwp with a value of "0" and outputs an on / off signal Qwn with a value of "1". Figure 8 In FIG. 1 , the waveform of the on / off signal Qwp is shown, but since the relationship Qwp+Qwn=1 exists, the waveform of the on / off signal Qwn is omitted.

[0119] The on / off signal generator 15 compares the phase voltage command Vv with any one of the first carrier signal and the second carrier signal. As a result of the comparison, when the phase voltage command Vv is greater than the carrier signal, the on / off signal generator 15 outputs an on / off signal Qvp with a value of "1" and outputs an on / off signal Qvn with a value of "0".

[0120] Here, if Figure 8 As shown, the phase voltage command Vv is consistent with the maximum value of the first carrier signal. Therefore, even if the phase voltage command Vv is compared with any one of the first carrier signal and the second carrier signal, the on / off signal generator 15 outputs an on / off signal Qvp with a value of "1" and outputs an on / off signal Qvn with a value of "0".

[0121] As described above, there is no difference even if the on / off signal generator 15 compares the phase voltage command Vv with either the first carrier signal or the second carrier signal. However, as a specific example, the on / off signal generator 15 compares the phase voltage command Vv with the first carrier signal. Figure 8 In FIG. 1 , the waveform of the on / off signal Qvp is shown, but since the relationship Qvp+Qvn=1 exists, the waveform of the on / off signal Qvn is omitted.

[0122] As described above, the on / off signal generator 15 compares the phase voltage command of at least one phase among the three-phase phase voltage commands Vu to Vw shifted by the first voltage shifting method with the first carrier signal, and compares the phase voltage commands of the remaining phases among the three-phase phase voltage commands Vu to Vw shifted with the second carrier signal. The on / off signal generator 15 switches the switching elements Sup to Swn of the inverter 6 to on and off through such comparison.

[0123] Next, the operation of the current acquisition unit 16 will be described. The current acquisition unit 16 acquires the bus current detection value Iins from the current detector 7 at each of the timing A and the timing B.

[0124] Here, the timing A and the timing B are different timings and are timings at which the bus current Iin flows from the DC power supply 3 to the inverter 6. The timing B is shifted from the timing A by 1 / 2 of the period Tc, for example.

[0125] In addition, in Embodiment 2, if Figure 8As shown, the case where the phase voltage command compared with the first carrier signal is the phase voltage command Vu of the U phase, and the phase voltage command compared with the second carrier signal is the phase voltage command of the W phase is illustrated. In this case, the first phase, which is a phase other than the phase corresponding to the maximum phase voltage command and the phase corresponding to the phase voltage command compared with the first carrier signal, becomes the U phase. In addition, the second phase, which is a phase other than the phase corresponding to the maximum phase voltage command and the phase corresponding to the phase voltage command compared with the second carrier signal, becomes the W phase.

[0126] In addition, in Embodiment 2, if Figure 8 As shown, a case where the first carrier signal becomes a maximum value at timing A and becomes a minimum value at timing B is illustrated.

[0127] Hereinafter, a method in which the current acquisition unit 16 calculates the phase current calculated values ​​Ius_cal to Iws_cal when the first voltage shift method is applied will be described.

[0128] Here, in Figure 8 In the example, if we focus on the period from time t3 to time t5, the voltage vector V4 is output during this period. Figure 6 It can be seen that the bus current detection value Iins is equivalent to the phase current calculation value -Ius_cal. During this period, since the voltage vector V4 is continuously output, the bus current detection value Iins continues to increase.

[0129] The timing A at which the current acquisition unit 16 acquires the bus current detection value Iins from the current detector 7 is set to the middle time t4 in the period from time t3 to time t5. The timing A is in the middle of the period during which the voltage vector V4 is continuously output. With such a setting, the current acquisition unit 16 can acquire the average value of the bus current detection value Iins that varies according to the output of the voltage vector V4.

[0130] Then, in Figure 8 In the example, if we focus on the period from time t6 to time t8, the voltage vector V2 is output during this period. Figure 6 It can be seen that the bus current detection value Iins is equivalent to the phase current calculation value -Iws_cal. During this period, since the voltage vector V2 is continuously output, the bus current detection value Iins continues to increase.

[0131] The timing B at which the current acquisition unit 16 acquires the bus current detection value Iins from the current detector 7 is set to the middle time t7 in the period from time t6 to time t8. This timing B is in the middle of the period during which the voltage vector V2 is continuously output. With such a setting, the current acquisition unit 16 can acquire the average value of the bus current detection value Iins that varies according to the output of the voltage vector V2.

[0132] As described above, the current acquisition unit 16 acquires the bus current detection value Iins from the current detector 7 at each of the timing A and the timing B, and calculates the first phase current calculation value and the second phase current calculation value based on the bus current detection value Iins acquired at each timing.

[0133] Specifically, when the first voltage shift method is applied, the current acquisition unit 16 calculates the first phase current calculated value based on the bus current detection value Iins acquired at timing A, and calculates the second phase current calculated value based on the bus current detection value Iins acquired at timing B.

[0134] More specifically, the current acquisition unit 16 sets the bus current detection value Iins acquired at timing A to the phase current calculation value -Ius_cal corresponding to U. In addition, the current acquisition unit 16 sets the bus current detection value Iins acquired at timing B to the phase current calculation value -Iws_cal corresponding to W. That is, the current acquisition unit 16 uses the bus current detection value Iins acquired at timing A and timing B, respectively, to calculate the phase current calculation values ​​corresponding to the U phase and the W phase other than the V phase corresponding to the maximum phase voltage command.

[0135] The current acquisition unit 16 uses the phase current calculated values ​​-Ius_cal and -Iws_cal calculated by the above method to calculate the phase current calculated value Ivs_cal in accordance with the relationship of Ivs_cal=-Ius_cal-Iws_cal, similarly to the first embodiment.

[0136] Then, in Figure 8 In the following, a case where an offset current value Iin_offset considered to be included in the bus current detection value Iins is considered will be described.

[0137] according to Figure 6 In the pattern of the on / off signals Qup to Qwn shown in FIG. 1 , when a zero voltage vector, that is, a voltage vector V0 or a voltage vector V7 is output, the bus current detection value Iins should be 0. However, in reality, the bus current detection value Iins does not become 0, and the bus current detection value Iins is a value other than 0. This value can be determined as an offset caused by temperature drift of the current detector 7, that is, an offset current value Iin_offset.

[0138] Therefore, if Figure 8 As shown, a timing between the timing A and the timing B is set, and this timing is a timing C at which the bus current Iin from the DC power supply 3 to the inverter 6 is cut off. Figure 8 The timing C shown is within the period in which the voltage vector V7, which is a zero voltage vector, is output. The timing C is, for example, between the timing A and the timing B.

[0139] For the sake of convenience, the bus current detection values ​​Iins acquired by the current acquisition unit 16 at timings A, B, C, and D are hereinafter referred to as bus current detection values ​​Iins(A), Iins(B), Iins(C), and Iins(D), respectively.

[0140] The current acquisition unit 16 further acquires the bus current detection value Iins from the current detector 7 at time tC within the period from time t5 to time t6, that is, timing C. The bus current detection value Iins(C) is equal to the offset current value Iin_offset.

[0141] The current acquisition unit 16 calculates the value obtained by subtracting the bus current detection value Iins(C) from the bus current detection value Iins(A) as the phase current calculation value -Ius_cal of the U phase. In addition, the current acquisition unit 16 calculates the value obtained by subtracting the bus current detection value Iins(C) from the bus current detection value Iins(B) as the phase current calculation value -Iws_cal of the W phase. In this way, the influence of the offset current value Iin_offset can be eliminated, and as a result, a phase current calculation value with high accuracy can be calculated.

[0142] Thus, the current acquisition unit 16 calculates the first phase current calculation value and the second phase current calculation value based on the bus current detection value Iins(A) obtained at timing A, the bus current detection value Iins(B) obtained at timing B, and the offset current value Iin_offset equivalent to the bus current detection value Iins(C) obtained at timing C.

[0143] In addition, if Figure 8 As shown, the current acquisition unit 16 may be configured to further acquire the bus current detection value Iins at timing D. Timing D is the timing at which the bus current Iin from the DC power supply 3 to the inverter 6 is cut off, and timing A exists between the timing D and timing C. For example, the middle of the period from timing D to timing C is timing A.

[0144] In the above case, the current acquisition unit 16 calculates an offset current value Iin_offset, which is equivalent to the average value of the bus current detection value Iins(C) acquired at timing C and the bus current detection value Iins(D) acquired at timing D. The current acquisition unit 16 calculates the first phase current calculation value and the second phase current calculation value based on the bus current detection value Iins(A) acquired at timing A, the bus current detection value Iins(B) acquired at timing B, and the offset current value Iin_offset. As a result, a more accurate phase current calculation value can be calculated.

[0145] In addition, in Embodiment 2, the application Figure 8 The first voltage shift method shown in the figure is not limited thereto, and the following method may also be applied. That is, the on / off signal generator 15 applies a second voltage shift method, which shifts the three-phase phase voltage commands Vu to Vw equally so that the minimum phase voltage command coincides with the minimum value of the first carrier signal. Specifically, Fig. 9 As shown, the phase voltage commands Vu to Vw are equally shifted so that the phase voltage command Vw, which is the minimum phase voltage command, coincides with the minimum value of the first carrier signal.

[0146] Fig. 9 1 is a timing chart showing a second example of the temporal change of the bus current detection value Iins in the second embodiment of the present invention.

[0147] like Fig. 9 As shown, the on / off signal generator 15 compares the phase voltage command Vu with the first carrier signal. As a result of the comparison, when the phase voltage command Vu is greater than the first carrier signal, the on / off signal generator 15 outputs an on / off signal Qup with a value of "1" and outputs an on / off signal Qun with a value of "0".

[0148] On the other hand, when the phase voltage command Vu is smaller than the first carrier signal, the on / off signal generator 15 outputs an on / off signal Qup with a value of "0" and outputs an on / off signal Qun with a value of "1". Fig. 9 In FIG. 1 , the waveform of the on / off signal Qup is shown, but since the relationship Qup+Qun=1 exists, the waveform of the on / off signal Qun is omitted.

[0149] The on / off signal generator 15 compares the phase voltage command Vv with the second carrier signal. As a result of the comparison, when the phase voltage command Vv is greater than the second carrier signal, the on / off signal generator 15 outputs an on / off signal Qvp of "1" and outputs an on / off signal Qvn of "0".

[0150] On the other hand, when the phase voltage command Vv is smaller than the second carrier signal, the on / off signal generator 15 outputs an on / off signal Qvp of "0" and outputs an on / off signal Qvn of "1". Fig. 9 In FIG. 1 , the waveform of the on / off signal Qvp is shown, but since the relationship Qvp+Qvn=1 exists, the waveform of the on / off signal Qvn is omitted.

[0151] The on / off signal generator 15 compares the phase voltage command Vw with any one of the first carrier signal and the second carrier signal. As a result of the comparison, when the phase voltage command Vw is smaller than the carrier signal, the on / off signal generator 15 outputs an on / off signal Qwp with a value of "0" and outputs an on / off signal Qwn with a value of "1".

[0152] Here, if Fig. 9 As shown, the phase voltage command Vw is consistent with the minimum value of the first carrier signal. Therefore, even if the phase voltage command Vw is compared with any one of the first carrier signal and the second carrier signal, the on / off signal generator 15 outputs an on / off signal Qwp with a value of "0" and outputs an on / off signal Qwn with a value of "1".

[0153] Therefore, even if the on / off signal generator 15 compares the phase voltage command Vw with either the first carrier signal or the second carrier signal, there is no difference. However, as a specific example, the on / off signal generator 15 compares the phase voltage command Vw with the first carrier signal. Fig. 9 In FIG. 1 , the waveform of the on / off signal Qwp is shown, but since the relationship Qwp+Qwn=1 exists, the waveform of the on / off signal Qwn is omitted.

[0154] Hereinafter, a method in which the current acquisition unit 16 calculates the phase current calculated values ​​Ius_cal to Iws_cal when the second voltage shift method is applied will be described.

[0155] Here, in Fig. 9 In the example, if we focus on the period from time t13 to time t15, the voltage vector V3 is output during this period. Figure 6 It can be seen that the bus current detection value Iins is equivalent to the phase current calculation value Ivs_cal. During this period, since the voltage vector V3 is continuously output, the bus current detection value Iins continues to increase.

[0156] The timing A at which the current acquisition unit 16 acquires the bus current detection value Iins from the current detector 7 is set to the middle time t14 in the period from time t13 to time t15. The timing A is in the middle of the period during which the voltage vector V3 is continuously output. With such a setting, the current acquisition unit 16 can acquire the average value of the bus current detection value Iins that varies according to the output of the voltage vector V3.

[0157] Then, in Fig. 9 In the example, if we focus on the period from time t16 to time t18, the voltage vector V1 is output during this period. Figure 6 It can be seen that the bus current detection value Iins is equivalent to the phase current calculation value Ius_cal. During this period, since the voltage vector V1 is continuously output, the bus current detection value Iins continues to increase.

[0158] The timing B at which the current acquisition unit 16 acquires the bus current detection value Iins from the current detector 7 is set to the middle time t17 between the time t16 and the time t18. The timing B is in the middle of the period during which the voltage vector V1 is continuously output. With such a setting, the current acquisition unit 16 can acquire the average value of the bus current detection value Iins that varies according to the output of the voltage vector V1.

[0159] As described above, the current acquisition unit 16 acquires the bus current detection value Iins from the current detector 7 at each of the timing A and the timing B, and calculates the first phase current calculation value and the second phase current calculation value based on the bus current detection value Iins acquired at each timing.

[0160] Specifically, when the second voltage shift method is applied, the current acquisition unit 16 calculates the second phase current calculated value based on the bus current detection value Iins acquired at timing A, and calculates the first phase current calculated value based on the bus current detection value Iins acquired at timing B.

[0161] More specifically, the current acquisition unit 16 sets the bus current detection value Iins acquired at timing A to the phase current calculation value Ivs_cal corresponding to V. In addition, the current acquisition unit 16 sets the bus current detection value Iins acquired at timing B to the phase current calculation value Ius_cal corresponding to U. That is, the current acquisition unit 16 uses the bus current detection value Iins acquired at timing A and timing B, respectively, to calculate the phase current calculation values ​​corresponding to the U phase and the V phase other than the W phase corresponding to the minimum phase voltage command.

[0162] The current acquisition unit 16 uses the phase current calculated values ​​Ius_cal and Ivs_cal calculated by the above method to calculate the phase current calculated value Iws_cal in accordance with the relationship of Iws_cal=-Ius_cal-Ivs_cal, similarly to the first embodiment.

[0163] Then, in Fig. 9 In the following, a case where an offset current value Iin_offset considered to be included in the bus current detection value Iins is considered will be described.

[0164] like Fig. 9 As shown, a timing between the timing A and the timing B is set, and this timing is a timing C at which the bus current Iin from the DC power supply 3 to the inverter 6 is cut off. Fig. 9 The timing C shown is within the period of outputting the voltage vector V0 which is a zero voltage vector. The timing C is, for example, between the timing A and the timing B.

[0165] The current acquisition unit 16 further acquires the bus current detection value Iins from the current detector 7 at time tC in the period from time t15 to time t16, that is, timing C. The bus current detection value Iins(C) is equal to the offset current value Iin_offset.

[0166] The current acquisition unit 16 calculates the value obtained by subtracting the bus current detection value Iins(C) from the bus current detection value Iins(A) as the phase current calculation value Ivs_cal of the V phase. In addition, the current acquisition unit 16 calculates the value obtained by subtracting the bus current detection value Iins(C) from the bus current detection value Iins(B) as the phase current calculation value Ius_cal of the U phase. Thus, the influence of the offset current value Iin_offset can be eliminated, and as a result, a phase current calculation value with high accuracy can be calculated.

[0167] In addition, compared with the previous Figure 8 Similarly, the current acquisition unit 16 may be configured to further acquire the bus current detection value Iins at timing D. In the above case, the current acquisition unit 16 calculates an offset current value Iin_offset, which is equivalent to the average value of the bus current detection value Iins(C) acquired at timing C and the bus current detection value Iins(D) acquired at timing D. The current acquisition unit 16 calculates the first phase current calculation value and the second phase current calculation value based on the bus current detection value Iins(A) acquired at timing A, the bus current detection value Iins(B) acquired at timing B, and the offset current value Iin_offset.

[0168] Preferably, the timing C is set between the timing A and the timing B. The reason is as follows.

[0169] If the amplitude of the phase voltage command Vu~Vw increases, the interval between time t15 and time t16 becomes close, so it is difficult to obtain the bus current detection value Iins(C) equal to the offset current value Iin_offset. Therefore, by setting the timing C to the middle of the timing A and the timing B, the bus current detection value Iins(C) can be obtained even in the higher amplitude region of the phase voltage command Vu~Vw.

[0170] In addition, the timing C can be set to a position that is later than the middle of the period from timing A to timing B by 1 / 2 of the period required for the current acquisition unit 16 to acquire the bus current detection value Iins. Specifically, the timing C can be set to be delayed by half of the sum of the ringing convergence period, the sampling / holding period, and the analog / digital conversion period relative to the middle of the period from timing A to timing B. Thus, even in the amplitude region where the phase voltage commands Vu to Vw are higher, the error included in the bus current detection value Iins (C) can be eliminated.

[0171] Here, when the phase voltage commands Vu to Vw have a higher amplitude and the interval between time t15 and time t16 becomes shorter than the period required to obtain the bus current detection value Iins(C), it is impossible to obtain the bus current detection value Iins(C). In this case, the accuracy of the offset current value Iin_offset can be maintained by not calculating the offset current value Iin_offset.

[0172] Thus, the on / off signal generator 15 is configured to apply Figure 8 The first voltage shift mode shown or Fig. 9 The second voltage shifting method shown in FIG. Fig.10 and Fig.11 An example will be described below in which the on / off signal generator 15 selects the first voltage shift method or the second voltage shift method.

[0173] Fig.10 This is a diagram showing current vectors in Embodiment 2 of the present invention. Fig.11 This is a table showing the voltage shifting method selected according to the current phase θi of the current vector in the second embodiment of the present invention.

[0174] First, the on / off signal generator 15 calculates the current phase θi according to the following equation (2-1) using the phase current calculated values ​​Ius_cal to Iws_cal calculated by the current acquisition unit 16 .

[0175] θi=atan{(0.866×Ivs_cal-0.866×Iws_cal) / (Ius_cal-0.5×Ivs_cal-0.5×Iws_cal)}[deg]

[0176] ···(2-1)

[0177] The current phase θi calculated according to formula (2-1) is as follows Fig.10 As shown, the phase of the current vector rotating on the U axis, V axis and W axis becomes.

[0178] like Fig.11 As shown, the on / off signal generator 15 selects one of the first voltage shift mode and the second voltage shift mode according to the range of the calculated current phase θi. For example, if the calculated current phase θi is within the range of 30[deg] or more and less than 90[deg], then Fig.11 It can be known that the on / off signal generator 15 selects the second voltage shifting mode.

[0179] Therefore, the on / off signal generator 15 selects the voltage shifting method according to the current phase θi. Therefore, among the voltage vectors V1 to V6 except for the zero voltage vectors, namely the voltage vectors V0 and V7, the output is Fig.10 The current vector shown is the second closest and the third closest to the voltage vector. As a result, even when the power factor is low, the current Ic flowing through the smoothing capacitor 4 can be reduced.

[0180] As described above, according to the second embodiment, the controller 8 of the power conversion device is configured to apply the first voltage shift method or the second voltage shift method to control the switching elements Sup to Swn of the inverter to be turned on and off. In addition, the controller 8 obtains the bus current detection value from the current detector 7 at each timing of timing A and timing B, and calculates the first phase current detection value and the second phase current detection value based on the acquisition result.

[0181] Thus, even when any one of the first carrier signal and the second carrier signal is used for comparison with the phase voltage command, the two bus current detection values ​​obtained at the two current acquisition timings can be used to obtain the phase current calculation value. That is, a power conversion device that can suppress the accuracy degradation of the value of the phase current obtained by the controller 8 can be obtained.

[0182] In addition, in the prior art, as described in Table 1·3 on p.70 of Non-Patent Document 1, in a motor drive system that calculates three-phase current based on bus current, it is impossible to obtain the average value of the ripple component of the carrier signal contained in the three-phase phase current. In contrast, in the present embodiment 2, by setting the current acquisition timing in the middle of the period of continuous output of the effective voltage vector, the average value of the ripple component of the carrier signal contained in the three-phase phase current can be obtained. As a result, the accuracy of the value of the phase current obtained by the controller 8 of the power conversion device is improved. In addition, the current Ic flowing through the smoothing capacitor 4 can be reduced, thereby contributing to the miniaturization of the power conversion device.

[0183] Implementation 3

[0184] In the third embodiment of the present invention, a power conversion device configured to estimate the rotation position θ of the AC rotating electric machine 1 connected to the output side of the inverter 6 is described with respect to each structure of the previous first and second embodiments. In addition, in this third embodiment, the description of the same points as the previous first and second embodiments is omitted, and the description is centered on the points different from the previous first and second embodiments.

[0185] Fig.12 1 is a diagram showing the overall structure of a power conversion device according to Embodiment 3 of the present invention. Fig.12 As shown, the power conversion device in the third embodiment includes a smoothing capacitor 4 , an inverter 6 , a current detector 7 , and a controller 8 .

[0186] Unlike the previous second embodiment, the AC rotating electric machine 1 is not provided with the position detector 2. The current detector 7 has the same configuration as that of the previous second embodiment, and detects the value of the bus current Iin input to the inverter 6 as the bus current detection value Iins.

[0187] The controller 8 has a different structure from that of the previous embodiment 2. The controller 8 includes a coordinate converter 9, a subtractor 10, a subtractor 11, a current controller 12, a current controller 13, a coordinate converter 14, an on / off signal generator 15, a current acquisition unit 16, a voltage command generator 17 for position estimation, an adder 18, and a position estimator 19.

[0188] The coordinate converter 9 calculates the currents Id and Iq on the two rotation axes based on the phase current calculated values ​​Ius_cal to Iws_cal input from the current acquisition unit 16 and the estimated rotation position θe input from the position estimator 19 .

[0189] The coordinate converter 14 calculates driving voltage commands Vub to Vwb for driving the AC rotating electric machine 1 based on the voltages Vd and Vq on the two rotating axes and the estimated rotational position θe input from the position estimator 19 .

[0190] The position estimation voltage command generator 17 outputs position estimation voltage commands Vuh to Vwh having a frequency higher than the frequency of the drive voltage commands Vub to Vwb output from the coordinate converter 14. The adder 18 calculates the phase voltage commands Vu to Vw by respectively adding the drive voltage commands Vub to Vwb input from the coordinate converter 14 and the position estimation voltage commands Vuh to Vwh input from the position estimation voltage command generator 17. The phase voltage commands Vu to Vw output from the adder 18 satisfy the following relationship.

[0191] Vu=Vub+Vuh

[0192] Vv=Vvb+Vvh

[0193] Vw=Vwb+Vwh

[0194] Thus, the on / off signal generator 15 controls the inverter 6 based on the three-phase phase voltage commands Vu to Vw on which the position estimation voltage commands Vuh to Vwh are superimposed.

[0195] The position estimator 19 estimates the rotation position θ of the AC rotating electric machine 1 connected to the output side of the inverter 6 based on the first phase current calculation value and the second phase current calculation value calculated by the current acquisition unit 16. The position estimator 19 outputs the estimated rotation position θ as the estimated rotation position θe. Fig.13 The position estimator 19 will be described in detail. Fig.13 1 is a diagram showing the structure of the position estimation device 19 according to the third embodiment of the present invention. Fig.13 As shown, the position estimator 19 includes a three-phase to two-phase converter 191 , a Fourier transformer 192 , a multiplier 193 , a subtractor 194 , and a position operator 195 .

[0196] The three-phase to two-phase converter 191 converts the phase current calculated values ​​Ius_cal to Iws_cal input from the current acquisition unit 16 into currents iαs and iβs on fixed orthogonal coordinates, that is, on the α-β axis.

[0197] Fourier transformer 192 extracts high frequency components from currents iαs and iβs input from three-phase to two-phase converter 191, respectively, and obtains amplitudes Iαs and Iβs of the extracted high frequency components. The high frequency components are generated by position estimation voltage commands Vuh to Vwh.

[0198] The multiplier 193 calculates the amplitude square value Iαs by taking the square root of the amplitudes Iαs and Iβs input from the Fourier transformer 192. 2 and Iβs 2 , and outputs the operation result to subtractor 194.

[0199] Subtractor 194 obtains the squared amplitude value Iβs by 2 Subtract the squared amplitude Iαs from 2 The differential value ΔIαβ is calculated and the calculation result is output to the position calculator 195.

[0200] The position calculator 195 calculates the estimated rotational position θe based on the difference value ΔIαβ input from the subtractor 194 , and outputs the calculation result.

[0201] Next, an example of a method for calculating the estimated rotation position θe by the position calculator 195 will be described. For example, when the AC rotating electric machine 1 is a permanent magnet synchronous motor, the voltage equation in a fixed orthogonal coordinate system can be expressed by the following equation (3-1).

[0202] [Mathematical formula 1]

[0203]

[0204] in,

[0205] [Vαs Vβs] T : Voltage in fixed orthogonal coordinates

[0206] [iαs iβs] T :Current under fixed orthogonal coordinates (two-phase current)

[0207] R: stator resistance P: differential operator

[0208] Ld: d-axis inductance Lq: q-axis inductance

[0209] ω: Angular velocity of rotation (electrical angle)

[0210] θ: Rotation position (U axis ( Figure 7 ) and the phase difference of the magnetic pole)

[0211]

[0212] Assume that the AC rotating electric machine 1 is stopped or running at a low speed, and set to ω = 0. In this case, if the Laplace operator s is used instead of the differential operator P, the currents iαs and iβs in fixed orthogonal coordinates can be expressed as the following equation (3-2).

[0213] [Mathematical formula 2]

[0214]

[0215] When the position estimation voltage commands Vuh to Vwh are respectively superimposed on the drive voltage commands Vub - Vwb, the relationships R << Lα·ωh and R << Lβ·ωh hold. However, the angular frequency ωh of the position estimation voltage commands Vuh to Vwh is set to be sufficiently higher than the angular frequency of the drive voltage commands Vub to Vwb. Further, let s = jωh. Here, j is the imaginary unit.

[0216] In the above case, if the influence of the stator resistance R of the AC rotating machine 1 is ignored, Equation (3 - 2) can be expressed as the following Equation (3 - 3).

[0217] [Mathematical Formula 3]

[0218]

[0219] Furthermore, the position estimation voltage commands Vuh to Vwh can be expressed in the fixed orthogonal coordinates as the following Equation (3 - 4).

[0220] [Mathematical Formula 4]

[0221]

[0222] Where,

[0223] [Vαsh Vβsh] T : Position estimation voltage in the fixed orthogonal coordinates

[0224] Vαβh: Amplitude of the position estimation voltage in the fixed orthogonal coordinates

[0225] ωh: Angular frequency of the position estimation voltage

[0226] If Equation (3 - 4) is substituted into [Vαs Vβs] in Equation (3 - 3) T , and s = jωh is set, the following Equation (3 - 5) can be obtained.

[0227] [Mathematical Formula 5]

[0228]

[0229] Where,

[0230] It can be seen from Equation (3 - 5) that the amplitudes of the currents iαs and iβs in the fixed orthogonal coordinates each contain the rotational position θ.

[0231] As described above, the amplitudes Ias and Iβs of the high-frequency components respectively included in the currents iαs and iβs are obtained by Fourier transformer 192. When the operation shown in the following equation (3-6) is performed based on the amplitudes Iαs and Iβs obtained by Fourier transformer 192, a term including the rotation position θ is obtained. The operation shown in equation (3-6) is realized by multiplier 193 and subtractor 194.

[0232] [Mathematical formula 6]

[0233]

[0234] The position calculator 195 divides the ΔIαβ shown in equation (3-6) by the following equation:

[0235] [Formula 7]

[0236]

[0237] Then, cos2θ is obtained. Next, the position calculator 195 calculates θ by calculating the arc cosine of cos2θ. The θ thus calculated corresponds to the estimated rotation position θe.

[0238] Alternatively, the position calculator 195 may be configured as follows, for example, instead of calculating the arccosine of cos2θ: that is, the position calculator 195 may store a table in advance that associates the value of cos2θ with the value of θ, and refer to the table to obtain θ corresponding to cos2θ as the estimated rotation position θe.

[0239] The position estimation voltage command generator 17 may be configured to superimpose a position estimation voltage command having an angular frequency ωh on the voltage Vd or Vq on the two rotating axes. In this case, the position estimator 19 is configured to calculate the estimated rotational position θe by extracting the component of the angular frequency ωh included in the phase current calculation values ​​Ius_cal to Iws_cal.

[0240] Furthermore, the position estimator 19 may be configured to obtain the estimated rotational position θe by applying a known position estimation method using an induced voltage of the AC rotating electric machine 1 when the angular velocity ω of the AC rotating electric machine 1 is high.

[0241] As described above, according to the third embodiment, the controller 8 of the power conversion device is configured to estimate the rotation position θ of the AC rotating electric machine 1 connected to the output side of the inverter 6, compared with the respective configurations of the previous first and second embodiments. That is, the controller 8 is configured to control the inverter 6 based on the three-phase phase voltage commands Vu to Vw on which the position estimation voltage commands Vuh to Vwh are superimposed, and to estimate the rotation position θ of the AC rotating electric machine 1 based on the calculated first phase current calculation value and the calculated second phase current calculation value. Thus, the same effects as those of the previous first and second embodiments are obtained, and since the position detector 2 can be omitted, it contributes to further miniaturization and cost reduction of the power conversion device.

[0242] Implementation 4

[0243] In Embodiment 4 of the present invention, an electric power steering device including the power conversion device of Embodiment 1 is described. In Embodiment 4, description of the same points as Embodiments 1 to 3 is omitted, and description will be focused on the points different from Embodiments 1 to 3.

[0244] Fig.14 It is a diagram showing the overall structure of an electric power steering device according to a fourth embodiment of the present invention.

[0245] The driver of a vehicle equipped with an electric power steering device steers the front wheels 102 by rotating the steering wheel 101 left and right. The torque detector 103 detects the steering torque Ts of the power steering system and outputs the detected steering torque Ts to the control command generator 105 described later. The AC rotating motor 1 generates an assist torque for assisting the driver in steering through the gear 104. The AC rotating motor 1 in the fourth embodiment has the same structure as the AC rotating motor 1 in the previous first embodiment, except that the rotor is mechanically connected to the gear 104.

[0246] The control command generator 105 calculates a control command for controlling the AC rotating electric machine 1 to a desired state based on the steering torque Ts input from the torque detector 103, and outputs the calculated control command. The control command generator 105 calculates a current command Iq_target as a control command according to the following equation (4-1), for example.

[0247] Iq_target=ka×Ts···(4-1)

[0248] Here, ka is a constant, but may be set to vary according to the steering torque Ts or the vehicle's running speed. Here, the current command Iq_target is determined according to equation (4-1), but the current command Iq_target may be determined based on a known compensation control corresponding to the steering state.

[0249] Next, effects obtained by the electric power steering device according to the fourth embodiment will be described.

[0250] The electric power steering device is required to be miniaturized. By miniaturizing the electric power steering device, the mountability on the vehicle is improved, the degree of freedom of arrangement is increased, and it can also contribute to the miniaturization of the vehicle itself.

[0251] The reduction in size of passive components such as capacitors and coils constituting the inverter of the electric power steering device directly affects the miniaturization of the inverter. Therefore, the prior art described in Patent Document 1, which generates a PWM signal using two carrier signals whose phases are shifted by a period of Tc / 2, is effective.

[0252] Here, in the electric power steering device, miniaturization of the current detector is also important for achieving system miniaturization. Therefore, we considered adopting the Figure 1 In the case where the current detector 7 shown is used as the current detector, the current detector is miniaturized in the power conversion device to which the above-mentioned prior art is applied. However, in this current detector, as described above, the accuracy of the value of the phase current obtained by the controller of the inverter of the power conversion device may be deteriorated.

[0253] Therefore, in the power conversion device of the first embodiment, the timing A and the timing B described in the first embodiment are set as the timing at which the controller 8 controlling the inverter 6 acquires the current. Figure 1 In the case of the current detector 7 shown in the figure, it is also possible to suppress the degradation of the accuracy of the value of the phase current obtained by the controller 8. Therefore, according to the power conversion device in the first embodiment, the miniaturization of the capacitor and the miniaturization of the current detector can be achieved at the same time. As a result, a significant effect of contributing to the miniaturization of the electric power steering device can be obtained.

[0254] In the fourth embodiment, the electric power steering device is configured by including the power conversion device in the first embodiment, but the present invention is not limited thereto. That is, the electric power steering device may be configured by including the power conversion device in any of the second and third embodiments.

[0255] As described above, according to the fourth embodiment, the electric power steering device may be configured to include any of the power conversion devices of the first to third embodiments. This can achieve the same effects as the first to third embodiments and further contribute to the miniaturization of the electric power steering device.

[0256] In addition, each function of the controller 8 of the power conversion device according to each of the above-mentioned Embodiments 1 to 3 is realized by a processing circuit. The processing circuit realizing each function may be dedicated hardware or a processor that executes a program stored in a memory.

[0257] When the processing circuit is dedicated hardware, the processing circuit is, for example, a single circuit, a composite circuit, a programmable processor, a parallel programmable processor, an ASIC (Application Specific Integrated Circuit), an FPGA (Field-Programmable Gate Array), or a combination thereof. The functions of each part of the controller 8 may be implemented by a separate processing circuit, or the functions of each part may be integrated and implemented by the processing circuit.

[0258] On the other hand, when the processing circuit is a processor, the functions of each part of the controller 8 are implemented by software, firmware, or a combination of software and firmware. Software and firmware are expressed as programs and stored in a memory. The processor reads and executes the program stored in the memory, thereby implementing the functions of each part.

[0259] The above-mentioned program can also be considered as a program that causes the computing machine to execute the functions of the above-mentioned parts. Here, the memory is equivalent to, for example, RAM (Random Access Memory), ROM (Read Only Memory), flash memory, EPROM (Erasable Programmable Read Only Memory), EEPROM (Electrically Erasable Programmable Read Only Memory) and other non-volatile or volatile semiconductor memories. In addition, magnetic disks, floppy disks, optical disks, compact disks, mini disks, DVDs, etc. are also equivalent to memories.

[0260] In addition, for the functions of the above-mentioned parts, part of them can be realized by dedicated hardware, and part of them can be realized by software or firmware.

[0261] Therefore, the processing circuit can use hardware, software, firmware or a combination thereof to implement the functions of the above-mentioned parts.

[0262] In addition, as embodiments of the present invention, embodiments 1 to 4 are described, but the present invention is not limited to the structures of embodiments 1 to 4. Without departing from the gist of the present application, the structures of embodiments 1 to 4 can be appropriately combined, and each structure can be partially deformed or partially omitted.

[0263] Description of symbols

[0264] 1 AC rotating motor, 2 position detector, 3 DC power supply, 4 smoothing capacitor, 5 inductor, 6 inverter, 7 current detector, 8 controller, 9 coordinate converter, 10 subtractor, 11 subtractor, 12 current controller, 13 current controller, 14 coordinate converter, 15 on / off signal generator, 16 current acquisition unit, 17 voltage command generator for position estimation, 18 adder, 19 position estimator, 191 three-phase-two-phase converter, 192 Fourier transformer, 193 multiplier, 194 subtractor, 195 position operator, 101 steering wheel, 102 front wheel, 103 torque detector, 104 gear, 105 control command generator.

Claims

1. A power conversion device, characterized in that: include: An inverter having a high potential side switch element and a low potential side switch element provided corresponding to each of the three phases, and converting a DC voltage output by a DC power supply into an AC voltage and outputting the AC voltage by switching each switch element on and off; a current detector connected in series with each low potential side switching element of the inverter, and detecting values ​​of phase currents of the three phases as phase current detection values; as well as a controller that calculates phase voltage instructions for the three phases and controls the inverter according to the calculated phase voltage instructions for the three phases, The controller performs the following actions: The phase voltage command of at least one phase among the three-phase phase voltage commands is compared with a first carrier signal, and the phase voltage commands of the remaining phases among the three-phase phase voltage commands are compared with a second carrier signal whose phase differs from the first carrier signal by 1 / 2 of the carrier period of the first carrier signal, thereby switching and controlling each switching element of the inverter to be on and off, acquiring a first phase current detection value as the phase current detection value of the first phase from the current detector at timing A, and calculating a first phase current calculation value corresponding to the first phase based on the first phase current detection value obtained at the timing, wherein the first phase is a phase corresponding to the phase voltage command compared with the first carrier signal, acquiring a second phase current detection value as the phase current detection value of the second phase from the current detector at timing B, and calculating a second phase current calculation value corresponding to the second phase based on the second phase current detection value obtained at the timing, wherein the second phase is the phase corresponding to the phase voltage command compared with the second carrier signal, The controller performs the following actions: further acquiring the first phase current detection value from the current detector at the timing B, and calculating the first phase current calculation value based on the first phase current detection value acquired at each of the timing A and the timing B, further acquiring the second-phase current detection value from the current detector at the timing A, and calculating the second-phase current calculation value based on the second-phase current detection value acquired at each of the timing A and the timing B, The first phase current calculated value is calculated by subtracting an offset current value equivalent to the first phase current detection value obtained at the timing B from the first phase current detection value obtained at the timing A, The second phase current calculated value is calculated by subtracting an offset current value equivalent to the second phase current detection value obtained at the timing A from the second phase current detection value obtained at the timing B, The controller calculates the phase voltage command using the first phase current calculated value and the second phase current calculated value.

2. The power conversion device according to claim 1, characterized in that: The timing A is the timing at which the low potential side switch element corresponding to the first phase becomes conductive when the phase voltage command of the first phase is zero. The timing B is the timing at which the low potential side switching element corresponding to the second phase becomes conductive when the phase voltage command of the second phase is zero.

3. The power conversion device according to claim 1 or 2, characterized in that: The timing B is offset from the timing A by 1 / 2 of the carrier cycle.

4. The power conversion device according to claim 3, characterized in that: At the timing A, the first carrier signal becomes a maximum value. At the timing B, the first carrier signal becomes a minimum value.

5. The power conversion device according to claim 1 or 2, characterized in that: The controller calculates the three-phase phase voltage commands at a period that is a natural number multiple of 2 or more of the carrier period.

6. The power conversion device according to claim 1 or 2, characterized in that: The controller uses an average value of the first phase current detection value acquired at the latest timing A and the first phase current detection value acquired at the previous timing A before the latest timing A when computing the first phase current computation value.

7. The power conversion device according to claim 1 or 2, characterized in that: The controller uses an average value of the second-phase current detection value acquired at the latest timing B and the second-phase current detection value acquired at the previous timing B before the latest timing B when calculating the second-phase current calculated value.

8. The power conversion device according to claim 1, characterized in that: The controller performs the following actions: controlling the inverter according to the phase voltage instructions of the three phases on which the voltage instructions for position estimation are superimposed, The rotation position of the AC rotating electric machine connected to the output side of the inverter is estimated based on the calculated first-phase current calculated value and the second-phase current calculated value.

9. An electric power steering device, characterized in that: The invention comprises a power conversion device as claimed in any one of claims 1 to 8.

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