High performance two-stage power converter with enhanced light load management

Abstract

The present disclosure relates to high performance two-stage power converters with enhanced light load management. The present disclosure provides a two-stage power converter that can include a buck pre-regulator and a resonant bus converter. Such converters can be operated to achieve unconditional soft-switching operation (zero voltage switching a / k / a ZVS) over a wide input and output range while delivering superior power conversion efficiency at lower power levels and no load conditions.

Description

BACKGROUND

[0001] Electronic devices often require some form of power supply. Power supply designs are constantly trading off between required power, efficiency, size, and other parameters. In particular, AC / DC adapters for providing primary power to portable consumer electronic devices such as laptops, tablets, smartphones, etc. present a variety of conflicting requirements. First, as these devices become more powerful, both the amount of power delivered and the overall efficiency of the power adapter become important. In addition, in many use cases such adapters can be plugged into the primary power even if the device they are powering has been disconnected. In this case, it can be desirable to greatly limit the amount of power wasted by the adapter. Furthermore, because such adapters are intended to be portable, i.e., users can take them with them when traveling with the device, size is also an important constraint. Finally, as with all consumer products, complexity can impact both reliability and cost. SUMMARY

[0002] Based on the foregoing, it is desirable to provide a power converter that addresses the foregoing problems. In some embodiments, such a power converter can be used to provide a high performance solution for low power, off-line AC / DC power converters, including those used in USB-C type AC / DC adapters or other low power consumer electronic power converters, as well as other applications.

[0003] A two-stage power converter can include a step-down pre-regulator stage configured to receive an input voltage and generate an intermediate voltage lower than the input voltage, and a resonant bus converter stage configured to receive the intermediate voltage generated by the step-down pre-regulator stage and generate an output voltage that is a fixed multiple of the intermediate voltage. The converter can also include a control circuit coupled to the step-down pre-regulator stage and the resonant bus converter stage, where the control circuit is configured to perform at least one of: adaptive soft-switching boundary mode operation of the step-down pre-regulator stage; and controlled burst mode operation of the resonant bus converter.

[0004] The resonant bus converter can be a half-bridge converter including first and second switching devices coupled in series across an intermediate voltage, first and second resonant capacitors coupled in series across the intermediate voltage, and a transformer having a primary winding coupled between the junction of the first and second switching devices and the junction of the first and second resonant capacitors, the transformer further having at least one secondary winding coupled through one or more rectifier devices to an output of the two-stage converter. The first and second switching devices can be alternately operated 180 degrees out of phase with approximately 50% duty cycle with a dead time to prevent cross conduction, resulting in a sinusoidal current in the transformer primary winding. The resonant bus converter can be further configured to achieve zero voltage switching of the first and second switching devices. The first and second resonant capacitors can be configured to resonate with the transformer and layout parasitic inductance or with additional discrete inductors in series with the transformer windings.

[0005] Alternatively, the resonant bus converter can be a full-bridge including first and second switching devices coupled in series across an intermediate voltage, third and fourth switching devices coupled in series across the intermediate voltage, and a resonant circuit including at least a resonant capacitor and a transformer primary winding coupled between the junction of the first and second switching devices and the junction of the third and fourth switching devices, wherein the transformer further has at least one secondary winding coupled through one or more rectifier devices to an output of the two-stage converter. The first and third switching devices and the second and fourth switching devices can be alternately operated 180 degrees out of phase with approximately 50% duty cycle with a dead time to prevent cross conduction, resulting in a sinusoidal current in the transformer primary winding. The resonant bus converter can be configured to achieve zero voltage switching of the first and second switching devices. The resonant circuit can include the transformer and layout parasitic inductance or discrete inductors in series with the transformer windings.

[0006] The buck pre-regulator stage can include a first switching device and a second switching device coupled in series across an input voltage. The first switching device can be a power control switch and the second switching device can be a reverse current control switch. A buck inductor can be coupled between a junction of the first and second switching devices and an input of the resonant bus converter. A freewheeling diode can be coupled in parallel with the reverse current control switch. A first current sensor can be coupled to the control circuit and configured to sense a buck pre-regulator current during a turn-on time of the first switching device. A second current sensor can be coupled to the control circuit and configured to sense a freewheeling current during a turn-off time of the first switching device. The first current sensor can be a current sense resistor connected in series with at least the buck inductor during the turn-on time of the first switching device. The second current sensor can be a current sense resistor connected in series with the second switching device.

[0007] The control circuit of the two-stage power converter can be configured to perform adaptive soft-switching boundary mode operation of the buck pre-regulator stage by: turning on the first switching device such that current flows through the buck inductor; comparing an output of the first current sensor to a peak current command derived by the control circuit from an output voltage of the two-stage converter; turning off the first switching device when the current flowing through the buck inductor reaches the peak current; turning on the second switching device after turning off the first switching device; detecting a reverse current flowing through the second switching device using the second current sensor; and turning on the first switching device in a zero voltage switching transition in response to the reverse current.

[0008] The control circuit of the two-stage power converter can be configured to perform controlled burst mode operation of the two-stage converter by: detecting a load condition at or below a point at which switching losses and bias power losses of the buck pre-regulator stage and conduction losses and core losses of the buck pre-regulator stage are in balance; and temporarily and intermittently disabling the switches of the buck pre-regulator stage and the resonant bus converter stage in response to the load condition. The control circuit can be configured to detect the load condition at or below the point at which switching losses and bias power losses of the buck pre-regulator stage and conduction losses and core losses of the buck pre-regulator stage are in balance by detecting a load current below a predetermined threshold. Temporarily and intermittently disabling the switches of the buck pre-regulator stage and the resonant bus converter stage can include disabling the switches of the buck pre-regulator stage and the resonant bus converter stage simultaneously or can include disabling the switches of the buck pre-regulator stage first and then disabling the resonant bus converter stage.

[0009] A method of operating a two-stage converter having a first buck pre-regulator stage and a second resonant bus converter stage can include adaptive soft-switching boundary mode operation of the buck pre-regulator stage by: turning on a power control switching device of the buck pre-regulator stage such that current flows through a buck inductor of the buck pre-regulator stage; comparing a sensed current flowing through the buck inductor to a peak current command derived from an output voltage of the two-stage converter; turning off the first switching device when the current flowing through the buck inductor reaches the peak current; turning on a reverse current control switching device of the buck pre-regulator stage after turning off the first switching device; detecting a reverse current flowing through the reverse current control switching device and turning on the power control switching device in a zero voltage switching transition in response to the reverse current.

[0010] The method of operating a two-stage converter having a first buck pre-regulator stage and a second resonant bus converter stage can alternatively or in addition thereto include controlled burst mode operation of the resonant bus converter by: detecting a load condition at or below a point at which switching losses and bias power losses of the buck pre-regulator stage and conduction losses and core losses of the buck pre-regulator stage are in balance and, in response thereto, temporarily and intermittently disabling the switches of the buck pre-regulator stage and the resonant bus converter stage. Detecting a load condition at or below a point at which switching losses and bias power losses of the buck pre-regulator stage and conduction losses and core losses of the buck pre-regulator stage are in balance includes detecting a load current below a predetermined threshold. Temporarily and intermittently disabling the switches of the buck pre-regulator stage and the resonant bus converter stage can include disabling the switches of the buck pre-regulator stage and the resonant bus converter stage simultaneously or can include disabling the switches of the buck pre-regulator stage first and subsequently disabling the resonant bus converter stage. BRIEF DESCRIPTION OF DRAWINGS

[0011] Figure 1 An embodiment of a two-stage converter including a buck pre-regulator and a bus converter is shown.

[0012] Figure 2 An alternative embodiment of a two-stage converter including a buck pre-regulator and a bus converter is shown.

[0013] Figure 3 Yet another alternative embodiment of a two-stage converter including a buck pre-regulator and a bus converter is shown, the bus converter including an inductor in an output filter stage.

[0014] Figure 4 A schematic diagram of an improved two-stage power converter based on a buck pre-regulator and a bus converter is shown.

[0015] Figure 5 An equivalent circuit of a resonant bus converter is shown.

[0016] Figure 6 and Figure 7 Certain waveforms depicting operation of the resonant bus converter 500 are shown. Figure 5

[0017] Figure 8 A full-bridge resonant converter is shown.

[0018] Figure 9 Another embodiment of a two-stage converter including a buck pre-regulator and a bus converter is shown.

[0019] Figure 10 Various possible relationships of reverse current of the buck pre-regulator stage to input voltage are shown.

[0020] Figure 11 Some specific operating waveforms of the buck pre-regulator are shown.

[0021] Figure 12 A simplified schematic diagram of a two-stage converter including a buck pre-regulator and a bus converter is shown.

[0022] Figure 13 A timing diagram showing on-off / pulse train mode operation as the power output is gradually reduced to the knee point is shown. DETAILED DESCRIPTION

[0023] In the following description, for purposes of explanation, numerous specific details are set forth in order to provide a thorough understanding of the disclosed concept. As part of this description, some of the diagrams in the drawings of the present disclosure are represented as block diagrams illustrating the architecture, functionality, and operation of possible implementations. In this regard, each block can represent a module, a procedure, a or a function. These blocks, and any other components of the present disclosure, can be implemented as electronic hardware, software, firmware, special-purpose computers, or combinations thereof. Moreover, each block of the diagrams can be implemented in hardware, software, firmware, or a combination thereof and can be implemented as a single device or multiple devices. In some embodiments, the blocks can be implemented as computer-executable instructions that are stored in non-transitory computer-readable media and that are executed by one or more processors that are a part of a computing device.

[0024] ​Various embodiments of the disclosed concepts are illustrated by way of example, and not by way of limitation, in the accompanying drawings, wherein like reference numbers refer to similar elements. For simplicity and clarity, reference numbers have been repeated in the description, where appropriate, to indicate different embodiments of the same or similar elements. Moreover, many specifics have been set forth in order to provide a thorough understanding of the embodiments described herein. In other instances, details have been omitted where such details would be understood by those skilled in the art. References to "one," "another," or "the" embodiment throughout this disclosure are not necessarily to the same or different embodiments, and this means at least one. A given figure can be used to illustrate more than one embodiment or kind of feature of the present disclosure, and not all elements in a figure can be required for a given embodiment or kind. When provided in a given figure, reference numbers refer to the same elements throughout several figures, but they can not be repeated in every figure. The drawings are not to scale, and the proportions of certain parts can be exaggerated for better illustration of details and features of the present disclosure.

[0025] Two-stage power converters can include a buck pre-regulator and an isolated bus converter. The isolated bus converter can be an isolated stage with a fixed gain set by the transformer turns ratio. In such converters, the buck pre-regulator can control the input voltage to the bus converter in order to achieve the desired voltage regulation at the output.

[0026] Figure 1 One embodiment of such a two-stage converter including a buck pre-regulator 101 and a bus converter 102 is shown. The output voltage of the buck pre-regulator 101 appears across capacitor C2, which is the input to the bus converter 102. Figure 2 One embodiment of a similar converter is shown. Figure 2 Converter 200 in Figure 1 Converter 100 in is operated substantially the same as converter 200, except that the buck pre-regulator control MOSFET Ql can be placed on the ground side for easier control. The bus converter 202 (or 102) can be driven using an isolated driver such as a driver transformer or a level shift driver. Figure 3 A schematic diagram of converter 300 is shown, which is yet another variation of the architecture and includes an inductor L2 in the output filter stage.

[0027] Bus converter switches Q2 and Q3 operate 180 degrees out of phase and close to 50% duty cycle, with a short dead time between switch transitions. This dead time can be introduced to avoid cross-conduction of Q2 and Q3, and also to allow zero voltage switching (ZVS). The buck output voltage across capacitor C2 must be lower than the minimum expected input voltage, which is limited by the maximum operating duty cycle of the buck controller.Figure 1 and Figure 2 The bus converter 102 / 202 in FIGS. 1-2 can be a resonant bus converter operating in a substantially boundary mode with near sinusoidal load current. The resonant frequency can be determined by the transformer and the parasitic leakage inductance of split capacitors C3 and C4. Energy stored in the magnetizing inductance can be used for zero voltage switching (ZVS) conversion over the entire load range. Conversely, Figure 3 The bus 302 converter depicted in FIG. 3 can operate with a substantially trapezoidal load current waveform. Zero voltage conversion can depend on energy stored in the leakage inductance, and in some embodiments, can be achieved at higher loads.

[0028] In each of the foregoing embodiments, the bus converter transformer TX1 can use a selected turns ratio to deliver a desired target output voltage. The transformer TX1 can also provide galvanic isolation, which can be necessary in certain applications. A control circuit (not shown) can compare the output voltage +V_OUT to a fixed reference, and can control the buck converter MOSFET Q1 to achieve output voltage regulation.

[0029] Such systems can suffer from certain limitations. For example, the buck converter can operate in conversion or boundary mode, but it achieves zero voltage conversion only when the buck converter operates at a duty cycle much higher than 50%. Furthermore, if such a converter is to be operated directly in a wide range off-line application such as a low power adapter (e.g., 90 Vac to 264 Vac input range), the efficiency of the buck converter can be poor or even quite poor. For example, at the lowest ac input voltage of 90 Vac and at full load, +V_IN can be as low as 75 VDC close to the valley of the bulk voltage. This in turn requires the buck converter output voltage to be very low, e.g., in the range of 60 V to 70 V. The low line efficiency can be improved to some extent by using a Q1 device with a very low rated on-resistance; however, the efficiency at high line voltage can be adversely affected due to the corresponding higher capacitance hard switching losses.

[0030] Another limitation of such a circuit is that the bus converter transformer TX1 experiences a fixed core loss due to operating at a fixed duty cycle and a fixed input voltage. As a result, the fixed power consumption of the bus converter can adversely affect the efficiency and input power at very low load conditions, which is a common mode of operation for ac / dc adapters in consumer electronics.

[0031] Figure 4A schematic diagram of an improved two-stage power converter 400 based on a buck pre-regulator 401 and a bus converter 402 is shown. The converter 400 can be made an "enhanced" two-stage power converter by including one or more of at least two aspects selected and configured so as to deliver high performance in low power AC / DC power supply applications. These two aspects can include (1) a resonant bus converter with controlled burst operation and (2) an adaptive soft-switching boundary mode buck pre-regulator, both of which are described in more detail below.

[0032] Resonant bus converter

[0033] The intrinsic leakage inductance of the bus converter transformer TX1 together with the external layout parasitic inductance of the converter can be used to build a resonant bus converter 500 that can operate at substantially unity gain. The equivalent circuit of such a resonant bus converter is shown in Figure 5 The inductor Lr need not be a discrete inductor, but can be a combination of transformer leakage inductance, primary side layout parasitic inductance, and reflected secondary stray inductance. Additional inductance can be provided, for example, in the form of a discrete inductor in series with the primary winding. Also, the circuit layout can be designed to provide the required level of stray and / or parasitic inductance. In any case, this inductance Lr in combination with the input split capacitors Cr1 and Cr2 can form a resonant tank circuit. The resonant time of the tank is given by:

[0034]

[0035] The MOSFET Qa and MOSFET Qb can operate 180 degrees out of phase and at substantially 50% duty cycle, except for a short dead time to prevent cross-conduction of the two MOSFETs. This dead time can also be used for zero voltage switching (ZVS) conversion. The resonant bus converter 500 is also sometimes simply described as a "bus converter" in the following description.

[0036] In some example practical implementations, the MOSFET Qa and MOSFET Qb can operate with a dead time of about 100 ns to 500 ns, with the particular value selected based on switching frequency and other circuit constraints. When Qa or Qb is on, Lr resonates with the combination of C3 and C4 to generate sinusoidal currents in the primary winding of the transformer Tx1 and the corresponding secondary winding. The values of Cr1 and Cr2 can be selected so that the half-resonant time of the circuit is substantially equal to the turn-on time of the MOSFET. This will result in boundary mode operation at the resonant frequency in each half cycle. For performance optimization, the circuit can be designed to operate slightly above or below the resonant frequency in each half cycle.

[0037] As a result of the above described switching operation, a triangular magnetizing current can also be established in the primary winding magnetizing inductance Lm. The primary inductance can be selected such that sufficient magnetizing current is built up in the primary to achieve zero voltage switching transitions during the dead time period, while discharging the total "effective capacitance" across the respective MOSFETs. This "effective capacitance" includes the two MOSFETs' parasitic drain-source capacitance, the reflected secondary side rectifier parasitic capacitance, and any leakage capacitance of the transformer. Thus, when each MOSFET turns off, the reflected secondary current has already dropped to zero or substantially close to zero, and the magnetizing current is at its peak.

[0038] The turns ratio of the transformer TX1 can be selected to match any desired output voltage based on the target buck output voltage.

[0039] Because the bus converter 500 operates in zero voltage switching (ZVS) mode, it does not experience capacitive turn-on switching losses. In addition, at turn-off, the current in each MOSFET is very low (i.e., the magnetizing current). Thus, the turn-off switching losses of the bus converter 500 are also very low, to the point of being almost non-existent. Thus, the bus converter 500 effectively has only resistive losses, which can allow operation at very high frequencies. Furthermore, the voltage transitions during the switching dead time period can have very slow rise times, which can significantly reduce common mode noise emissions.

[0040] Figure 6 and Figure 7 Certain waveforms are shown that depict the operation of the resonant bus converter 500 as Figure 5 shown in FIG. 6. Figure 6 The primary reflected load current 601 is shown, which has sinusoidal properties and reaches the zero level during the turn-on time of the two MOSFET switches. More specifically, one half cycle of the sinusoidal load current corresponds to the turn-on time of each respective switching device (shown by the switching drive signals 603 and 604). When the switch is active, the primary magnetizing current 602 ramps up in a triangular fashion. Figure 7 The ZVS operation of the switches is shown (particularly for switch Qb, although switch Qa is substantially similar but out of phase by 180 degrees). More specifically, Figure 7 The drive signal 704 for switch Qb is depicted (corresponding to the drive signal 604 shown in Figure 6 FIG. 6). Figure 7 The drain to source voltage 702 of the switch is also shown, which can be seen to go below zero at turn-on.

[0041] The aforementioned resonant bus converter 500 is a half-bridge resonant bus converter. However, the resonant bus converter can also be constructed as a full-bridge resonant bus converter 800 (as Figure 8MOSFET Qa and MOSFET Qal (i.e., the corresponding switches in the opposite bridge leg) can be driven simultaneously in one half-cycle, while MOSFET Qb and MOSFET Qbl can be driven simultaneously in the other half-cycle. The drain inductance Lr and series resonant capacitor Cr form a resonant tank. The circuit operates substantially similar to that described above with respect to Figures 5 to 7 The half-bridge resonant bus converter is described. In some embodiments, such as those delivering lower power levels, the half-bridge resonant bus converter can be preferred due to reduced circuit complexity and part count; however, either embodiment can be used depending on the requirements of a particular application.

[0042] As described above, the illustrated resonant bus converter produces an output voltage that is a fixed multiple of its input voltage (the multiple being determined by the turns ratio of transformer Tx1). Because such resonant bus converters do not have the ability to regulate their output voltage independently of their input voltage, a buck pre-regulator 901 can be combined with the resonant bus converter 902 to form a two-stage converter 900, as Figure 9 MOSFET Q1 and MOSFET Q2, along with diode D3, inductor LI, and capacitor C2, form the buck pre-regulator stage 901. Switch Q1 functions as a power control switch in buck converter operation, and switch Q2 performs the functions of reverse current control and / or ZVS control. The buck converter 901 can operate in peak current control mode with boundary mode operation. Diode D3 functions as a freewheeling rectifier for the buck converter. In some embodiments, switch Q2 can also function as an additional freewheeling element / synchronous rectifier. In some such embodiments, D3 can be omitted, depending on the particular semiconductor technology used, for example. However, for at least some embodiments, it can be advantageous to retain diode D3.

[0043] The operation of the buck converter 901 using the above-described peak current mode control can be as follows. When switch Q1 is turned on, current +V IN from the input source ramps up through inductor LI, capacitor C2, the load impedance reflected through the resonant bus converter, and finally to current sense resistor R1, back to the input source. The buck controller converter 903 can use the current sense signal voltage across R1 (CS1) to turn off switch Q1 when the current value reaches the peak current limit set by the control circuit. Error amplifier 904 can be configured to compare the converter output voltage (+V OUT) to a fixed reference (+VREF) to set the peak current limit of the buck converter, which can be signal 905 provided to controller 903. In some embodiments, error amplifier 904 and associated circuitry can be part of controller 903. These components together form the control circuit for the two-stage converter.

[0044] When switch Q1 is turned off by controller 903 (at the peak current limit set by error amplifier 904), the current flowing through inductor L1 is transferred to a parallel combination of diode D3 and the intrinsic body diode of switch Q2 (which has a resistor R2 connected in series with it). Once diode D3 begins to conduct, switch Q2 can be turned on, which is a zero-voltage switching (ZVS) transition. Therefore, switch Q2 does not experience any capacitive switching losses. In some embodiments, the size of switch Q2 can be set slightly smaller than that of switch Q1, and therefore it can have a higher on-resistance. Thus, diode D3 can carry most of the inductor current. However, as mentioned above, if switch Q2 has a sufficiently low on-resistance, it can be used as a synchronous rectifier, and in at least some embodiments, the freewheeling diode D3 can be omitted. However, such a choice may depend on the design goals and cost constraints of a particular application.

[0045] Once the energy stored in the buck inductor L1 is depleted, the current flowing through diode D3 drops to zero. The current in the inductor then reverses due to the energy stored in capacitor C2, driving a reverse current through switch Q2 (and series resistor R2). The voltage drop across resistor R2 (i.e., signal CS2) can be used to provide a reverse current (I_rev) sensing signal to controller 903. When the reverse current sensing signal CS2 reaches a predetermined threshold (corresponding to the desired level of reverse current), switch Q2 can be turned off by controller 903. The energy stored in inductor L1 due to the reverse current causes the drain node of switch Q2 to rise until the effective capacitance across switch Q1 discharges. Switch Q1 can then be turned on by control circuitry 903 during ZVS switching. It should be understood that allowing reverse current to flow through inductor L1 is for achieving ZVS switching on switch Q1.

[0046] In many cases, power converters designed for low-power consumer electronics applications are configured to operate over a wide range of practical input AC voltages. For example, it is not uncommon for "universal" adapters to operate with input voltages ranging from 90Vac to 265Vac. Therefore, the DC bus voltage, after rectification and filtering by the body capacitor, can vary widely. For example, the DC bus voltage can vary between 80Vdc and 375Vdc when considering the low-frequency ripple across the body capacitor. The lowest input DC bus voltage across the body capacitor can, for example, correspond to a 90Vac supply at maximum load, as the valley of the DC bus voltage. The lowest DC bus voltage can be manipulated to some extent based on the selection of the body filter capacitor value. For applications employing… Figure 9The illustrated embodiment of the two-stage converter 900, the step-down pre-regulator 901 will always have a gain less than unity. In other words, the output voltage of the step-down pre-regulator 901 will always be lower than its input voltage, and thus for a "universal" adapter as described above, its output will be lower than the lowest input DC voltage, e.g., < 80 Vdc. In some embodiments, the maximum duty cycle of the step-down converter can be kept in the range of 90% to 95% in order to better control regulation. Thus, for some embodiments, the output voltage of the step-down pre-regulator can be less than about 72 V. Some embodiments that employ a relatively low capacitance value for the DC bus bulk capacitor can see even lower voltages. When deciding on the operating step-down output voltage for an application of the two-stage converter as illustrated, such factors should be considered. Figure 9

[0047] In the following discussion, an exemplary embodiment will be used to explain the operation of the two-stage converter 900. In the example illustrated, the DC input voltage can be in the range of 80 Vdc to 375 Vdc, and the DC output voltage can be 12 Vdc at a 5 A maximum (i.e., a 60 W converter), which can be used as an adapter for a mid-range consumer laptop or similar load. Because the half-bridge resonant bus converter 900 operates with substantially unity gain, its transformer TX1 can be selected to have a turns ratio Np:Ns = 3: 1, where Np is the primary turns and Ns is the secondary turns. Since the primary winding sees half of the input voltage due to the split capacitors Cr3 and Cr4, the effective gain of the half-bridge resonant converter 902 is 6. Thus, to achieve a 12 V output voltage, the expected voltage regulation level of the step-down pre-regulator 901 is about 72 V. In practice, due to voltage drops in various resistive circuit elements and output rectifier diodes, the voltage can be slightly higher than 72 V. However, for the purposes of the following explanation, such voltage drops are ignored.

[0048] Step-down pre-regulator

[0049] ​In a boundary mode buck regulator, power is delivered to the load and energy is stored in the buck inductor LI during the active (i.e., "on") time of the power control switch Ql. When switch Ql is turned off, the energy stored in LI is delivered completely to the load. Once the buck inductor LI is completely discharged, in the absence of switch Q2, a reverse current naturally builds up in inductor LI by drawing current from the energy stored in capacitor C2 due to the resonant action between the inductor LI and the effective parasitic capacitance seen at the input node of the buck inductor LI. Such parasitic capacitance can include the intrinsic capacitance of the freewheeling diode D3 and the intrinsic "drain to source" capacitance of MOSFET Ql. If the voltage across capacitor C2 is higher than half the buck regulator input voltage (i.e., the buck converter duty cycle is greater than 50% or 0.5), the resonant ringing can discharge the effective parasitic capacitance at the input node of inductor LI completely. In other words, the valley of the resonant ringing voltage across switch Ql can allow ZVS operation.

[0050] Accordingly, the control circuit 903 can be configured to compare the input voltage of the buck pre-regulator 901 to its output voltage appearing across capacitor C2. Whenever the output voltage is higher than half the input voltage (i.e., duty cycle > 0.5), the operation of the reverse current control switch Q2 can be disabled. In this mode, the buck pre-regulator power control switch Ql can be turned on when its drain to source voltage drops to zero volts due to natural resonant ringing. By definition, this is a zero voltage switching (ZVS) event. This operation can be achieved by sensing the zero voltage event directly or indirectly, or by using a timing signal based on the resonant time of the circuit after detecting that the buck inductor LI is discharged.

[0051] Alternatively, in at least some embodiments, instead of disabling the operation of the reverse current control switch Q2, a small, predetermined reverse current (I rev th) can instead be allowed to be carried by the reverse current control switch Q2 when the buck converter is operated with a duty cycle > 0.5, where I rev th > 0. This reverse current flowing in switch Q2 can be sensed by a signal voltage (I rev) dropping across resistor R2. This can significantly simplify the control operation. Accordingly, when the set minimum reverse current is established in the buck inductor LI based on the CS2 signal sensed across resistor R2, the reverse current control switch Q2 can be turned off. When the reverse current control switch Q2 is turned off, this interruption of the reverse current in the buck inductor LI can cause the drain voltage of the reverse current control switch Q2 to rise until the capacitance at the input node of the buck inductor LI is completely discharged, allowing the power control switch Ql to turn on under ZVS conditions.

[0052] Since the input voltage in the example above varies between 80VDC and 375VDC, the allowable reverse current in the buck inductor L1 can be kept at a fixed low level when the input voltage is significantly below 144V, as this would result in a buck duty cycle >50%. In practical implementations, when the reverse current is insufficient to achieve ZVS operation for the target output voltage, the precise voltage level can be found through bench testing. This can be identified as the threshold voltage (Vth), above which the reverse current needs to be increased. Typically, the threshold voltage Vth will correspond to the point where the buck converter's duty cycle drops below 50%. From this level Vth upwards, the reverse current increases with increasing input voltage. The reverse current in the buck inductor L1 needs to be at its highest when the buck pre-regulator is operating at the maximum DC input voltage.

[0053] This increase in reverse current can have a linear or nonlinear relationship with the input voltage from the threshold point Vth, where the slope is determined by the nature of the capacitive behavior at that node. Figure 10 Various possible relationships between reverse current I_rev and input voltage V_in are illustrated. Typically, MOSFETs and diodes exhibit decreasing capacitance as the reverse voltage across them increases. Conversely, parasitic circuit capacitances associated with printed circuit board (PCB) layout and transformers remain constant over such voltage changes. In some implementations, the slope of the increasing reverse current curve can be simplified and linearized based on hardware testing. Even finer-grained reverse current measurement can be achieved by implementing a digital controller.

[0054] like Figure 9 The operating sequence of the two-stage converter 900 shown can be explained as follows. For the purposes of the following description, it is assumed that all operating bias voltages are available before the startup sequence. During initialization, the buck converter switches Q1 and Q2 remain off, while the resonant bus converter switches Q3 and Q4 are started. These two switches can be switched at a predetermined frequency and duty cycle, which can be determined based on the principles described above. In summary, the resonant capacitors C3 and C4 can be selected such that the half-resonant frequency of the resonant tank circuit consisting of C3, C4, and the total leakage inductance of the transformer TX1 (including external parasitic inductance, given by the equation above) is substantially equal to the turn-on times of Q3 and Q4, which can operate 180 degrees out of phase. Furthermore, a dead time, i.e., the duration for which both switches Q3 and Q4 are off, can be set so that ZVS operation of Q3 and Q4 is achieved using the magnetizing current of TX1. This may require an iterative process to determine the required primary inductance of TX1 to achieve ZVS within a dead time, which may be expected, for example, to be about 2% to 5% of the total switching cycle.

[0055] When switches Q3 and Q4 start operating as described above, there will be no voltage across capacitor C2 because the buck pre-regulator power control switch Ql has remained in the off state. Therefore, no current will flow through the resonant tank or transformer. After the operation of the resonant bus converter switches Q3 and Q4 is initiated, the buck pre-regulator can be allowed to start with a "soft start." In other words, the buck converter can start operating in peak current mode control, gradually increasing the peak current until the desired output voltage is reached.

[0056] As described above, the error amplifier 904 can be configured to compare the output voltage +V OUT to a fixed reference (+VREF). When the output voltage reaches the reference (e.g., 12Vdc regulated voltage level), the error signal 905 can be fed to the controller 903 to control the peak current of the buck inductor LI through an isolation mechanism such as an optocoupler. Furthermore, as described above, the buck pre-regulator 901 can operate in boundary or critical conduction mode with a controlled reverse current (I rev) to achieve ZVS operation of the power control switch Ql. Therefore, the buck converter 901 effectively operates in a variable frequency boundary mode. The peak current can be lower at lighter loads, which results in a higher operating frequency for a given fixed input voltage. Likewise, for a fixed peak current, a lower input voltage requires a longer Ql on-time, which corresponds to a lower operating frequency.

[0057] Buck pre-regulator losses

[0058] Figure 11 Some relevant operating waveforms of the buck pre-regulator 901 are shown. By comparing the drain-to-source voltage curve 1101 to the power control switch gate drive signal 1102, it can be seen that the power control switch Ql operates in ZVS conditions. (That is, the switching transitions of the power control switch Ql correspond to the times when the drain-to-source voltage across the switch is zero.) Otherwise, the power losses in the buck pre-regulator 901 can be classified into three categories:

[0059] (1) Conduction losses: These losses are primarily resistive in nature and decrease as the output power decreases. Conduction losses follow an "I 2 R" relationship, where R is the resistivity of the circuit and I is the RMS current in the conducting element. Exemplary resistive elements in the buck pre-regulator 901 include the switches, the inductor windings, the circuit layout resistance, and the current sensor. Furthermore, diode losses are partially resistive due to the inherent resistance of the PN junction and partially linear due to the forward voltage drop.

[0060] (2) Switching Losses: Due to the above ZVS operation, these losses are largely non-existent. Thus, the switching losses of the converter can be ignored. Because the ZVS operation is not completely lossless (because of the lossy nature of the switching and other circuit capacitances), the associated losses can increase slightly as the power delivered to the load decreases (which corresponds to an increased operating frequency).

[0061] (3) Magnetization Losses in the Inductor: These include hysteresis losses in the ferrite core. Because the buck pre-regulator 901 operates in peak current control mode, the core losses decrease as the output power delivered decreases.

[0062] (4) Bias Power Losses: Because the operating frequency increases as the load decreases, the circuit bias power and switching device drive power can increase slightly at lower loads.

[0063] As can be seen from the above loss descriptions, as the converter power throughput decreases, the conduction losses and the core losses also decrease, while the miscellaneous switching losses and bias power losses increase. Thus, starting from the maximum rated power and continuously decreasing the output power will result in the total buck pre-regulator losses decreasing to a minimum point, and then starting to increase. At this inflection point, a balance is achieved between "increasing switching losses and bias power losses" and "decreasing conduction losses and core losses". Further reduction of the power losses below this balance point can be achieved by "on-off" or "burst mode" type of control of the power throughput. In this mode, the peak current can be kept fixed at the above inflection point, and the entire buck pre-regulator can be alternately disabled and enabled to maintain output regulation. This mode can also be characterized as an "energy control mode".

[0064] Resonant bus converter power losses

[0065] The nature of the power losses in the resonant bus converter 902 is somewhat different from that in the buck pre-regulator 901. Because the resonant bus converter 902 operates at a substantially fixed input voltage, a fixed frequency, and a fixed on-time, it experiences fixed power losses due to the core losses in the magnetizing transformer, the bias power losses, and the capacitance losses. Although the resonant bus converter operation is ZVS (as described above), some capacitance losses can still be experienced due to the lossy nature of the parasitic capacitors.

[0066] In addition to these fixed losses, the resonant bus converter 902 can also operate in an "I 2The R-relationship involves resistive losses. These losses decrease significantly as the load decreases towards no-load conditions due to the square relationship with respect to the current flowing through the converter. In other words, the resonant bus converter 902 does not see the inflection point seen by the buck pre-regulator 901 when internal power losses begin to increase. This fact can have a significant impact on the overall power supply efficiency at very low loads. Furthermore, the resonant bus converter 902 can experience higher "no-load input power," which may be unacceptable in some consumer electronics applications. Therefore, in some implementations, it may be desirable to operate the resonant bus converter in an on-off or "burst" mode to reduce the average losses in the power converter at very low loads. However, this can be a tricky proposition, at least for the reasons described below.

[0067] Enhanced two-stage converter operation

[0068] The following describes methods for reducing and Figure 9 The diagram illustrates the control technique for the total loss associated with the two-stage converter 900. The converter 900 can be simply represented as follows: Figure 12 The converter 900' shown (ignoring the current isolation associated with transformer TX1). Because the buck pre-regulator 901 operates in boundary mode with peak current mode control, the current supplied by the buck pre-regulator 901 is proportional to the output load current drawn by the load resistor R_load. Therefore, the buck pre-regulator stage 901 in Figure 12 This is represented as current source I1 (901'). Similarly, the resonant bus converter 902' can be represented by switch S1, which simply connects and disconnects current source I1 with output capacitors C_out and R_load. Of course, bus converters 902 / 902' have a fixed gain based on the turns ratio of transformer TX1, but this gain is ignored for the purposes of this discussion, as there is no need to explain the operational concepts. It should be noted that bus converter 902', represented by switch S1, operates essentially with a 100% duty cycle, ignoring the short dead time between the switching of the two control switches.

[0069] Current sources must not be allowed to operate in open-circuit mode, as this could result in severely high voltages. Figure 12As shown, the capacitor C_buck can allow for short dead-time / open circuit conditions of the bus converter 902' / SI. In other words, the capacitor C_buck can absorb the energy delivered by the current source II during the short off-time interval of SI without catastrophic conditions. This will increase the voltage across the capacitor and appear at the input of the bus converter 902'. However, such open circuit conditions are fairly short, e.g., no more than 5-10% of the duty cycle, so the voltage rise can be acceptable. However, in practice, it would be better to not allow the bus converter 902' to turn off (thus creating an open circuit) whenever the buck pre-regulator 901' is operating.

[0070] A control technique is described below that allows the buck pre-regulator 901 / 901' and the resonant bus converter 902 / 902' to operate safely in a burst mode under light load conditions. By characterizing the hardware of the buck pre-regulator 901 / 901', a knee point can be determined at which the first power stage should enter a burst mode. For example, the knee point would correspond to a particular output current. At this point, the minimum peak current command of the buck pre-regulator 901 / 901' can be set by the control signal (V_comp) generated by the error amplifier 904( Figure 9 ) When the peak current control signal reaches a preset lower threshold "V_burst_l", the buck pre-regulator 901 / 901' can be disabled. Subsequently, the resonant bus converter 902 / 902' can also be disabled after completing the ongoing last switching period of the control switches Q3 and Q4( Figure 9 ).

[0071] Once both stages (i.e., the buck pre-regulator 901 / 901' and the resonant bus converter 902 / 902') are disabled, the associated losses in both power stages will become zero. The output capacitor C_out can continue to deliver power to the load until the output voltage drops below the set regulation level. The output voltage below the set regulation level can cause the control signal 905 delivered by the error amplifier 904 to increase. When the peak current control signal reaches a preset higher threshold "V_burst_2", the buck pre-regulator 901 / 901' and the resonant bus converter 902 / 902' can be re-enabled. The difference between the two peak current control thresholds V_burst_l and V_burst_2 provides a hysteresis for the burst mode, which can improve the stability and predictability of the operation. In addition, the above technique prevents the buck pre-regulator 901 / 901' from operating in an open circuit mode, thus preventing catastrophic high voltage conditions.

[0072] Variations of the above control technique are also possible. For example:

[0073] (1) The buck pre-regulator 901 / 901' and the resonant bus converter 902 / 902' can be disabled simultaneously when a lower control voltage (V_comp) threshold "V_burst_1" is sensed. The two converters can be enabled again when a higher control voltage threshold "V_burst_2" is reached.

[0074] (2) The buck pre-regulator 901 / 901' can be disabled first when a lower control voltage threshold "V_burst_1" is sensed, with the resonant bus converter 902 / 902' being disabled after a fixed number of switching cycles M is allowed, where M is greater than or equal to zero. The resonant bus converter 902 / 902' can be enabled first when a higher control voltage threshold "V_burst_2" is reached, and then the buck pre-regulator 901 / 901' can be enabled after N switching cycles of the bus converter, where N is greater than or equal to zero.

[0075] Figure 13 A timing diagram is shown that illustrates the on-off / pulse train mode operation when the power output is gradually decreased to the knee point. V_comp is the control voltage delivered by the error amplifier 904 that sets the peak current level of the buck pre-regulator 901 / 901'. Bus_PWM1 and Bus_PWM2 are the two complementary drive signals for the resonant bus converter switches Q3 and Q4. Figure 13 The behavior of V_comp at the knee point when the on-off / pulse train control mode is activated is shown. The duration taken for V_comp to increase from V_burst_1 to V_burst_2 (or vice versa) is a function of the output load on the two-stage converter and the speed of the error amplifier control loop. In addition, to prevent unipolar DC flux in transformer TX1, the on duration of the first pulse of the associated bus converter switch can be reduced to half of its normal duration, for example, by blanking the first half of its normal on duration. This can be seen in Figure 13 in the first pulse of BUS_PWM1 when it is enabled during the on-off mode.

[0076] Thus, when operated as described above, the two-stage converter can deliver high efficiency over a wide range of load conditions. Since all of its power switches operate in ZVS mode, higher switching frequency operation can be employed, which can advantageously allow new bandgap semiconductor devices, such as gallium nitride (GaN) devices, to be applied.

[0077] The foregoing describes exemplary embodiments of two-stage power converters employing a buck pre-regulator and a resonant bus converter. Such systems can be used in a variety of applications, but can be particularly advantageous when used in conjunction with power adapters for consumer electronics devices. Additionally, while a number of specific features and various embodiments have been described, it will be understood that various features and embodiments can be combined with each other, in particular implementations, unless otherwise explicitly stated as being mutually exclusive. Accordingly, the various embodiments described above are provided by way of example only, and should not be construed as limiting the scope of the disclosure. Various modifications and alterations to this disclosure will become apparent to those of ordinary skill in the art without departing from the scope and without departing from the scope of the claims.

Claims

1. A two-stage power converter, the two-stage power converter comprising: a buck pre-regulator stage configured to receive an input voltage and produce an intermediate voltage lower than the input voltage; a resonant bus converter stage configured to receive the intermediate voltage produced by the buck pre-regulator stage and produce an output voltage that is a fixed multiple of the intermediate voltage; and a control circuit coupled to the buck pre-regulator stage and the resonant bus converter stage, wherein the control circuit is configured to perform at least one of: adaptive soft-switching boundary mode operation of the buck pre-regulator stage; and controlled burst mode operation of the resonant bus converter; wherein the control circuit is configured to perform controlled burst mode operation of the resonant bus converter by: detecting a point at which switching losses and bias power losses of the buck pre-regulator stage and conduction losses and core losses of the buck pre-regulator stage are in balance or a load condition below the point; and temporarily and intermittently disabling and enabling switching of the buck pre-regulator stage and the resonant bus converter stage in response to the load condition.

2. The two-stage power converter of claim 1, wherein the resonant bus converter is a half-bridge, the half-bridge comprising: first and second switching devices coupled in series across the intermediate voltage; first and second resonant capacitors coupled in series across the intermediate voltage; and a transformer having a primary winding coupled between a junction of the first and second switching devices and a junction of the first and second resonant capacitors, the transformer further having at least one secondary winding coupled to an output of the two-stage converter through one or more rectifier devices; wherein: the first and second switching devices are alternately operated 180 degrees out of phase with approximately 50% duty cycle with dead time to prevent cross conduction, producing a sinusoidal current in the transformer primary winding; and the resonant bus converter is configured to achieve zero voltage switching of the first and second switching devices.

3. The two-stage power converter of claim 2, wherein the first and second resonant capacitors are configured to resonate with a parasitic inductance of the converter.

4. The two-stage power converter of claim 2, wherein the first and second resonant capacitors are configured to resonate with a discrete inductor.

5. The two-stage power converter of claim 1, wherein the resonant bus converter is a full-bridge, the full-bridge comprising: first and second switching devices coupled in series across the intermediate voltage; ​ ​ ​ a third switching device and a fourth switching device coupled in series across the intermediate voltage; and a resonant circuit including at least a resonant capacitor and a transformer primary winding coupled between the junction of the first and second switching devices and the junction of the third and fourth switching devices, wherein the transformer also has at least one secondary winding coupled to an output of the two-stage converter through one or more rectifier devices, wherein: the first and third switching devices and the second and fourth switching devices are alternately operated 180 degrees out of phase with a duty cycle of approximately 50% with dead time to prevent cross conduction, resulting in a sinusoidal current in the transformer primary winding; and the resonant bus converter is configured to enable zero voltage switching of the first and second switching devices.

6. The two-stage power converter of claim 5, wherein the resonant circuit includes a discrete inductor.

7. The two-stage power converter of claim 1, wherein the buck pre-regulator stage includes: a first switching device and a second switching device coupled in series across the input voltage, wherein the first switching device is a power control switch and the second switching device is a reverse current control switch; a buck inductor coupled between the junction of the first and second switching devices and an input of the resonant bus converter; a freewheeling diode coupled in parallel with the reverse current control switch; a first current sensor coupled to the control circuit and configured to sense a buck pre-regulator current during a turn-on time of the first switching device; and a second current sensor coupled to the control circuit and configured to sense a freewheeling current during a turn-off time of the first switching device.

8. The two-stage power converter of claim 7, wherein the first current sensor is a current sense resistor connected in series with at least the buck inductor during a turn-on time of the first switching device.

9. The two-stage power converter of claim 7, wherein the second current sensor is a current sense resistor connected in series with the second switching device.

10. The two-stage power converter of claim 7, wherein the control circuit is configured to perform adaptive soft-switching boundary mode operation of the buck pre-regulator stage by: turning on the first switching device, resulting in current flowing through the buck inductor; comparing an output of the first current sensor to a peak current command derived by the control circuit from an output voltage of the two-stage converter; turning off the first switching device when the current flowing through the buck inductor reaches the peak current; turning on the second switching device after turning off the first switching device; and turning off the second switching device when the current flowing through the buck inductor reaches the peak current. detecting a reverse current flowing through the second switching device using the second current sensor; and in response to the reverse current, turning on the first switching device in a zero voltage switching transition.

11. The two-stage power converter of claim 1, wherein the control circuit is configured to detect a point at which switching losses and bias power losses of the buck pre-regulator stage and conduction losses and core losses of the buck pre-regulator stage are in balance or a load condition below the point by detecting a lower disable control signal and a higher enable control signal at a load current below a first predetermined threshold.

12. The two-stage power converter of claim 1, wherein the control circuit is configured to enter the controlled burst mode in response to a control signal reaching a first threshold, and wherein the control circuit is configured to exit the controlled burst mode in response to the control signal reaching a second threshold.

13. The two-stage power converter of claim 1, wherein temporarily and intermittently disabling and enabling the switches of the buck pre-regulator stage and the resonant bus converter stage comprises: simultaneously disabling and enabling the switches of the buck pre-regulator stage and the resonant bus converter stage in response to respective disable and enable control signals.

14. The two-stage power converter of claim 1, wherein temporarily and intermittently disabling and enabling the switches of the buck pre-regulator stage and the resonant bus converter stage comprises: first disabling the switches of the buck pre-regulator stage and subsequently disabling the resonant bus converter stage in response to a disable control signal, and first enabling the switches of the resonant bus converter and subsequently enabling the buck pre-regulator stage in response to an enable control signal.

15. A method of operating a two-stage converter having a first buck pre-regulator stage and a second resonant bus converter stage, the method comprising: adaptive soft switching boundary mode operation of the buck pre-regulator stage, including: turning on a power control switching device of the buck pre-regulator stage such that current flows through a buck inductor of the buck pre-regulator stage; comparing a sensed current flowing through the buck inductor to a peak current command derived from an output voltage of the two-stage converter; turning off a first switching device when the current flowing through the buck inductor reaches the peak current; turning on a reverse current control switching device of the buck pre-regulator stage after turning off the first switching device; detecting a reverse current flowing through the reverse current control switching device; and in response to the reverse current, turning on the power control switching device in a zero voltage switching transition.

16. The method of claim 15, further comprising performing a controlled burst mode operation of the resonant bus converter, including: detecting a point at which switching losses and bias power losses of the buck pre-regulator stage and conduction losses and core losses of the buck pre-regulator stage are in balance or a load condition below the point; and in response to this load condition, temporarily and intermittently disabling and enabling the switches of the buck pre-regulator stage and the resonant bus converter stage.

17. The method of claim 16, wherein detecting a point at which switching losses and bias power losses at the buck pre-regulator stage and conduction losses and core losses at the buck pre-regulator stage are in equilibrium or a load condition below the point comprises: detecting a lower disable control signal and a higher enable control signal at a load current below a first predetermined threshold.

18. The method of claim 16, further comprising entering the controlled burst mode in response to a control signal reaching a first threshold, and exiting the controlled burst mode in response to the control signal reaching a second threshold.

19. The method of claim 16, wherein temporarily and intermittently disabling and enabling the switches of the buck pre-regulator stage and the resonant bus converter stage comprises: simultaneously disabling and enabling the switches of the buck pre-regulator stage and the resonant bus converter stage in response to respective disable and enable control signals.

20. The method of claim 16, wherein temporarily and intermittently disabling and enabling the switches of the buck pre-regulator stage and the resonant bus converter stage comprises: first disabling the switches of the buck pre-regulator stage and subsequently disabling the resonant bus converter stage in response to a disable control signal, and first enabling the switches of the resonant bus converter and subsequently enabling the buck pre-regulator stage in response to an enable control signal.

21. A method of operating a two-stage converter having a first buck pre-regulator stage and a second resonant bus converter stage, the method comprising: controlled burst mode operation of the resonant bus converter, including: detecting a point at which switching losses and bias power losses of the buck pre-regulator stage and conduction losses and core losses of the buck pre-regulator stage are in balance or a load condition below the point; and temporarily and intermittently disabling and enabling the switches of the buck pre-regulator stage and the resonant bus converter stage in response to the load condition; wherein detecting the point at which switching losses and bias power losses of the buck pre-regulator stage and conduction losses and core losses of the buck pre-regulator stage are in balance or the load condition below the point includes detecting a lower disable control signal and a higher enable control signal at a load current below a first predetermined threshold.

22. A method of operating a two-stage converter having a first buck pre-regulator stage and a second resonant bus converter stage, the method comprising: controlled burst mode operation of the resonant bus converter, including: detecting a point at which switching losses and bias power losses of the buck pre-regulator stage and conduction losses and core losses of the buck pre-regulator stage are in balance or a load condition below the point; and temporarily and intermittently disabling and enabling the switches of the buck pre-regulator stage and the resonant bus converter stage in response to the load condition; wherein temporarily and intermittently disabling and enabling the switches of the buck pre-regulator stage and the resonant bus converter stage includes simultaneously disabling and enabling the switches of the buck pre-regulator stage and the resonant bus converter stage in response to respective disable and enable control signals.

23. A method of operating a two-stage converter having a first buck pre-regulator stage and a second resonant bus converter stage, the method comprising: controlled burst mode operation of the resonant bus converter, including: detecting a point at which switching losses and bias power losses of the buck pre-regulator stage and conduction losses and core losses of the buck pre-regulator stage are in balance or a load condition below the point; and temporarily and intermittently disabling and enabling the switches of the buck pre-regulator stage and the resonant bus converter stage in response to the load condition; wherein temporarily and intermittently disabling and enabling the switches of the buck pre-regulator stage and the resonant bus converter stage includes first disabling the switches of the buck pre-regulator stage and subsequently disabling the resonant bus converter stage in response to a disable control signal, and first enabling the switches of the resonant bus converter and subsequently enabling the buck pre-regulator stage in response to an enable control signal.

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