A transimpedance amplifier circuit for measuring tunnel junction current signals in low-temperature STM

Abstract

The present invention discloses a transimpedance amplifier circuit for measuring tunnel junction current signals in a low-temperature STM, which belongs to the cross-field of STM measurement technology and amplifier technology. The core of this circuit is a large-bandwidth special preamplifier composed of HEMT field-effect transistors, which is powered by a single power supply with a current stabilization function, so that the HEMT can work safely and stably, especially placing the HEMT in a low-temperature area very close to the STM tunnel junction, reducing the capacitance C of the signal feeder. I , which increases the bandwidth of the circuit and solves the problem of capacitor C when the STM operating frequency is increased. I The serious problem of rapidly increasing circuit noise has been solved, which enables STM to complete scanning and spectrum measurement tasks at high speed and high precision. It also enables STM to avoid the interference of low-frequency 1 / f noise to measure weak signals such as tunnel junction shot noise, which provides conditions for using low-temperature STM to discover new phenomena and new laws of the movement of microscopic matter.

Description

Technical Field

[0001] The invention belongs to the cross field of low temperature scanning tunnel microscope measurement technology and amplifier circuit technology, and in particular relates to a transimpedance amplifier circuit with input for measuring tunnel junction current related signals in a low temperature scanning tunnel microscope. Background Art

[0002] The samples measured by the scanning tunneling microscope (STM) are crystals such as metals and semiconductors, quantum dots on the surface of substrates, single molecules adsorbed on metal surfaces, etc. During measurement, the probe is pointed at a certain point on the sample surface and the needle tip is kept at a very small distance from the sample, so that a "tunnel junction" is formed between the needle tip and the sample. A fixed DC bias V is applied to this tunnel junction, and the current I flowing through the tunnel junction can be measured. The probe is scanned at various points on the sample surface, and a morphological image reflecting the microstructure of the sample surface can be obtained by measuring the tunnel junction current I. This work is hereinafter referred to as "STM scanning". When the probe is fixed to point to a certain point on the sample surface, a DC bias V is applied to the tunnel junction, and a small voltage "sinusoidal modulation signal" with an amplitude of ΔV and a frequency of f is superimposed. The tunneling current change ΔI corresponding to ΔV is measured, and the differential conductivity spectrum point dI / dV corresponding to the DC bias V can be obtained. By changing the DC bias V multiple times to measure dI / dV, the differential conductivity spectrum line dI / dV-V can be obtained, and then the second-order differential spectrum line d 2 I / dV 2 -V; differential conductivity spectra and second-order differential spectra related to tunneling current all belong to "scanning tunneling spectra" (i.e. STS spectra); by scanning the probe at various points on the sample surface and measuring the STS spectra at each point, the STS spectrum of the sample surface can also be obtained. In addition, under appropriate conditions, the shot noise and capacitance of the tunnel junction can be measured by measuring the tunnel junction current. By scanning the probe at various points on the sample surface, shot noise spectra, capacitance spectra, etc. can be obtained. These spectra and spectra also contain information on the microscopic structure and physical properties of the sample surface. These works are hereinafter referred to as "STM spectroscopy". STM spectroscopy is a very effective means to study the microscopic physical properties of materials such as crystals, quantum dots, and single molecules.

[0003] When using STM to complete scanning and spectrum measurement, the current flowing through the tunnel junction needs to be amplified by a transimpedance amplifier (abbreviated as "TIA"), and the voltage signal is output after amplification. Then, the measuring instrument is connected to the output end of the TIA to complete the measurement. In existing low-temperature STMs, the probe and sample are generally placed in a low-temperature zone cooled by liquid helium, and the temperature of the tip-sample is 4.2K. Some low-temperature STMs have a tip-sample temperature below 0.5K. However, the preamplifier of the traditional TIA in the low-temperature STM is composed of bipolar transistors. This type of transistor cannot work in the low-temperature zone and can only be placed in the room temperature zone. Therefore, the TIA must be connected to the tip-sample in the low-temperature zone with a very long signal feeder (about 1 to 2 meters, generally a coaxial line).

[0004] STM measures tunnel junction current related signals, using a "TIA with input", whose basic structure and working principle are as follows: Figure 1 As shown in the figure, the circuit in the dotted rectangle is TIA, which consists of two parts: (i) Figure 1 The operational amplifier (abbreviated as "op amp") shown in the dotted triangle has two input terminals, P and N, and an output terminal O. The positive input terminal P is grounded. The equivalent input resistance R of this op amp is A and equivalent input capacitance C A connected in parallel to the two input terminals, (ii) there is a "feedback circuit" outside the op amp from the op amp output terminal O to its inverting input terminal N, Figure 1 Shown is only a feedback resistor R connected between O and N F , the actual feedback circuit may also be the feedback resistor R F Connect a circuit consisting of a "bandwidth compensation network" composed of resistors and capacitors. Figure 1 The circuit shown on the left outside the dotted rectangular frame can be called the "signal input circuit" connected to the TIA. Its structure is as follows: the tunnel junction TJ formed by the needle tip and the sample is between points A and B, which is connected in series to the voltage source of "DC bias superimposed modulation signal voltage" (abbreviated as "bias-modulation signal voltage source", abbreviated as BMS, represented by the series connection of bias power supply and modulation signal voltage source in the figure), and one end A is connected to the inverting input terminal N of the operational amplifier through a signal feeder, and the other end G is grounded. There is a so-called "feeder capacitor C" between the signal feeder AN and the ground. I ", the differential resistance of the tunnel junction is denoted as R J , its reciprocal is the differential conductance, denoted as G J , which is one of the contents of spectrum measurement; the needle tip and the sample also form a J The parallel "tunnel junction capacitance C J ", because it is very small, it can be considered that C J=0. The entire circuit formed by connecting the "signal input circuit" and the TIA may be called a "TIA with input", also referred to as a "TIA circuit" or a "transimpedance amplifier circuit" (note that it is not the same entity as the "TIA"). In this circuit, the ground potential is always 0, and the voltages at other points are relative to the ground. Due to the "virtual short" effect of the op amp, the DC potential of its inverting input terminal N can be considered to be 0 in terms of the DC working state, so the DC bias voltage V provided by the "bias-modulation signal voltage source BMS" i It is just applied to both ends of the tunnel junction, so that there is a DC voltage component V at the output of the op amp. o When measuring the spectrum, the modulation signal voltage provided by the BMS is a sinusoidal alternating voltage with a preset frequency f, which is expressed as (It is related to the DC bias voltage V i Phase superposition, denoted as ), when the whole circuit is in the sinusoidal steady state, there is a The corresponding sinusoidal alternating voltage output is expressed as At this time, the voltage at the N terminal is recorded as The voltage at the P terminal If is 0, Here a A is the open-loop voltage gain of the op amp, which is a function of the frequency f of the sinusoidal alternating signal, denoted by a A =a A (f) (when the frequency f approaches 0, it approaches a0). This function can be obtained by measuring the op amp, and it is usually given in the data sheet of the op amp; for example, a A =a0 / [(1+jf / f b )(1+jf / f b2 )](f b <f b2 ), where –2πf b and –2πf b2 They are respectively A The first and second poles of Considered as the input voltage of "TIA with input", the output voltage and Ratio It is called the voltage gain of the circuit and is denoted by A v ,for Figure 1 The circuit shown in the figure shows that a simple calculation shows that

[0005]

[0006] Therefore, the output of the TIA is measured by the instrument The differential resistance R of the tunnel junction can be calculated from the above formula J, thus obtaining the differential conductance G J When the circuit is used to scan the image, the modulation signal voltage zero, and the DC bias voltage V i Take it as a constant value, and the current I flowing through the tunnel junction can be known from the voltage measured at the output end of the TIA. If the tip constant height mode is used to scan the sample surface, the current I at each point of the sample can be detected to obtain information on the sample surface morphology; the tunnel junction constant current mode can also be used to scan the sample surface, which also relies on the measurement of the tunnel junction current I, and obtains an image of the sample surface morphology through special signal processing. Because the position of the tip relative to the sample surface must be first determined for STM spectrum measurement, the STM spectrum measurement work must be based on a certain degree of scanning accuracy.

[0007] The TIA with input cannot work if it is not stable; the so-called "instability" means that the circuit will self-oscillate, that is, under various excitations, there will be equal-amplitude oscillations different from the sinusoidal steady state, or wide-amplitude oscillations will occur, causing the circuit to enter a nonlinear saturation state. The following is based on the discussion on "stability" in Chapter 8 of [Reference 1] "Circuit Design Based on Operational Amplifiers and Analog Integrated Circuits" (4th edition, written by [US] Sergio Franco, published by Xi'an Jiaotong University Press in 2017) to discuss Figure 1 To improve the stability of the circuit shown in Reference 1, the loop gain T L The amplitude of |T L | (i.e., modulus) and argument ∠(T L ) (i.e. phase angle) characteristics to describe the stability of the circuit, which can be briefly described as follows: Since the output resistance of the op amp in the TIA used in STM is r o , and the feedback resistor R F and the load impedance behind the TIA |Z L |Comparison is at least 3 orders of magnitude smaller, so we might as well take r o = 0 for discussion, Figure 1 In the example, the bias-modulation signal voltage source BMS is removed and its two input terminals are short-circuited. The wire between points O and F is removed and replaced with the wire with broken ends O′ and F′ as indicated by the dotted line on the left. This forms a measuring T L Circuit, apply "measurement T" at point F' L Signal", which is a sinusoidal voltage with a frequency of f When measuring T L When the circuit is in a sinusoidal steady state, the sinusoidal voltage at point N can be calculated (or measured) (The voltage at point P ),remember So we get the voltage at point O' We put It is called the "return ratio factor". It is called "loop gain" (also known as "return ratio"), and it can be known from simple calculation that

[0008]

[0009] T L =a A β=a A / (1 / β), denoted as T L =T L (f); (f0-3)

[0010] Theoretically, if the circuit does not exist for all f∈(0,+∞) “|T L (f)|≥1 and ∠(T L (f)) = -180°”, the circuit is stable, and the further away from this condition, the higher the degree of stability; for example, if |T L (f)| and ∠(T L (f)) are all monotonically decreasing functions of f, and there exists a frequency f x Make |T L (f x )| dB =20lg|T L (f x )| is less than –6dB and ∠(T L (f x )) is greater than –150°, then in the frequency band (0,f x ]∠(T L (f)) is at least 30° higher than –180° (i.e., has at least 30° of amplitude margin in terms of stability) and is within the frequency band [f x ,+∞)on|T L (f)| dB Ratio(1) dB =0 is at least 6dB lower (i.e., there is at least 6dB of amplitude margin in terms of stability), then the circuit is considered to have "sufficient stability".

[0011] In the following, we refer to the frequency of the modulation signal used in STM spectrum measurement as the "operating frequency of the STM", and the frequency in the passband involved in signal processing when the STM scans the image as the "operating frequency of the STM".

[0012] The upper limit of the operating frequency of the conventional TIA circuit in the existing low-temperature STM (the TIA is represented by the FEMTO product DLPCA-200) is about 1kHz when scanning and measuring the spectrum. This frequency range is suitable for a large number of research topics on the microstructure and state of matter; however, for some research topics, the operating frequency f wThe upper limit of 1kHz is too low. For example (A): f w The corresponding period is T w =1 / f w It takes several T to measure one STS spectrum point. w It takes a lot of time to make an STS spectrum line. And some high-precision research often requires repeated measurement of hundreds of spectrum lines and averaging to get a spectrum line with high enough accuracy, which takes a lot of T. w The time required is several hours or even dozens of hours to measure. However, during such a long time, the object on the sample may have moved or changed its physical state, making it impossible to complete accurate measurement. w If the frequency is increased by 10 times, for example to 10kHz, the effective spectrum measurement time can be shortened by 10 times, so that accurate measurement can be completed before the sample "changes shape". Another example (B): the quantum state of the sample being measured changes rapidly. To measure the state at the beginning, middle and end of the change process, the measurement must be completed quickly, which also requires a sufficiently high operating frequency. Another example (C): the operating frequency f w When the frequency is below 1kHz, the low-frequency 1 / f noise voltage component of the TIA circuit output signal is particularly large compared with the useful signal voltage component. In some research projects, this noise will seriously interfere with the measurement of useful signals. w By increasing the frequency to above 10kHz and avoiding the interference of 1 / f noise, useful signals can be accurately measured. Special mention should be made of the measurement of the shot noise signal of the tunnel junction, which is urgently needed for studying major topics such as the Kondo effect and Andre reflection. However, the shot noise signal is relatively weak and is masked by the 1 / f noise in the low frequency band. To observe it, the TIA circuit must be able to achieve high-gain amplification in the high frequency band.

[0013] To increase the operating frequency, it is necessary to consider the J The parallel tunnel junction capacitance C J According to literature reports, the capacitor C J It is about 0.1 to 10 fF (see [Reference 2] "Measurement and Application of Scanning Probe Microscope Tip Capacitance" ("Electronic Measurement Technology" Vol. 32, No. 2, 2009), and [Reference 3] "Sensing the quantum limit in scanning tunneling spectroscopy" ("Nature Communications" October 2015)). It is estimated that R J C J About 0.1~1μs, if f w is 10kHz, then CJ Capacitive reactance | Z CJ |=1 / (2πf w C J ) is about R J 10 to 100 times of C J The influence on the spectrum measurement can be ignored. If the operating frequency is further increased to 300kHz~500kHz, the current flowing through C J The alternating current flowing through R J The alternating current flowing through C J The current signal can also extract information about the sample surface morphology, and can also measure the capacitance C J , making STM a scanning capacitance microscope. Therefore, increasing the bandwidth of the TIA circuit in the STM to 10kHz~500kHz does not lose its significance due to the existence of the tunnel junction capacitance.

[0014] However, for the low-temperature STM TIA circuit, increasing the operating frequency f w , there will be a problem that the circuit noise seriously interferes with the useful signal, which is described in detail as follows: There are many "noise sources" that generate noise in the TIA circuit (the noise generated by the tunnel junction is not counted here, because it belongs to the useful signal being measured; the noise generated by the "bias-modulation signal voltage source BMS" is also not counted). If the total effect generated by these noise sources is equivalent to the effect generated by a virtual noise current source connected to both ends of the tunnel junction, then the noise current generated by this virtual source is called the "equivalent input noise current" of the TIA circuit. Compared with the useful signal current flowing through the tunnel junction, the larger the former, the greater the interference to the measurement of the latter. The size of the equivalent input noise current of the TIA circuit is equal to the "equivalent input noise current power spectral density ” is the square root of the integral over the frequency band [f1, f2] to which the measurement relates, and It is called “equivalent input current noise density”. The components contributed by various noise sources in the circuit, such as One of the components is the feedback resistor R F The thermal noise contributes to (where T is the Kelvin temperature of the feedback resistor), this component is independent of the frequency f; There is also a capacitor C I The relevant ingredients are in is the equivalent input voltage noise density of the op amp in the TIA; Another major component is It is equal to the equivalent input current noise density of the op amp in the TIA. Taking the traditional TIA circuit in the existing low-temperature STM as an example (TIA is represented by FEMTO's product DLPCA-200), TIA and R F Both are in the room temperature region of T = 300K, the tip-sample is in the low temperature region of 4.2K, R F =1GΩ, C I =100pF, C A ≤5pF, C I +C A ≈C I According to FEMTO's product manual, (Because the equivalent input voltage noise density of the TIA and At f = 1kHz (because At f = 1kHz );then, And when f = 1kHz It can be seen from Table 1 that when f = 1kHz, The feedback resistor R F Contribution As the main component, when f = 10kHz, and C I Related ingredients Exceeded When f = 100kHz, Far more than Therefore, when using this circuit to measure the spectrum, even if the bandwidth of the TIA is expanded, it is allowed to increase the operating frequency f w , but in the relevant frequency band [f w -Δf / 2,f w +Δf / 2], the increased noise will also seriously affect the accuracy of spectrum measurement. If this circuit is used to quickly scan the image, the useful signal will be interfered by the large noise in the high frequency band, which will also seriously affect the clarity of the scanned image.

[0015] Table 1 Equivalent input current noise density of traditional TIA circuit in STM at different frequencies

[0016]

[0017] People have long recognized that the most direct and effective way to increase the bandwidth of TIA circuits without causing a sharp increase in circuit noise is to shorten the length of the signal feeder to reduce the feeder capacitance C. I , which requires moving the op amp or the preamplifier in the op amp to the low temperature area as much as possible.

[0018] [Reference 4] "Fast low-noise transimpedance amplifier for scanning tunneling microscopy and beyond" (Rev. Sci. Instrum. 91, 074701 (2020)) reported their TIA circuit for low-temperature STM, which contains three op amps, of which the first op amp is AD8615, which is composed of CMOS transistors. The nominal operating temperature of the product is -40 to 125 ° C, but they found that it can still work at liquid nitrogen temperature. It is placed in a liquid nitrogen temperature thermostat, which is not far from the liquid helium temperature zone where the needle tip and the sample are located, so that the capacitance C of the cable connecting the tunnel junction I is reduced to 35pF, making the equivalent input current noise density of the circuit at 10kHz: They expanded the bandwidth of the circuit to 50kHz for scanning, achieving high-speed scanning, and the circuit noise was not too high; however, the literature did not report the use of the circuit for spectrum measurement. Obviously, it is difficult for the AD8615 op amp placed at liquid nitrogen temperature to get closer to the needle tip-sample in the liquid helium temperature zone.

[0019] [Reference 5] "Atomic scale shot-noise using cryogenic MHz circuitry" (Rev. Sci. Instrum. 89, 093708 (2018)) reports a very special amplifier circuit for low-temperature STM. The traditional TIA circuit used for tracing and spectrum measurement in STM was retained, and this amplifier for measuring tunnel junction signals was connected in parallel. It is composed of a dedicated preamplifier (referred to as "dedicated preamplifier") placed at a temperature of 5K and a room-temperature amplifier in cascade (but it does not constitute an operational amplifier and has no feedback circuit). This dedicated preamplifier is made of a low-temperature HEMT transistor. This HEMT is a major research and development achievement of the French CNRS / LPN laboratory with ultra-low noise and power consumption. They also used a superconducting coil and a cable line capacitor C IA LC resonator is formed in parallel to connect the dedicated preamplifier and the tunnel junction in the 2K temperature zone. The circuit has a resonant frequency of 1MHz. This structure makes the circuit have a high amplification gain only for signals with frequencies near the resonant frequency, so that the STM can detect the shot noise of the tunnel junction tunneling current; they used this STM to study the shot noise properties of the tunneling current between the needle tip and the superconducting film, and achieved very outstanding scientific research results. They also used this STM to measure the STS spectrum. This work must be carried out when the modulation signal frequency is basically the same as the circuit resonant frequency. They measured the differential conductivity spectrum of a sample with a modulation frequency of 1MHz, and measured it again with a traditional TIA at a modulation frequency of 429Hz. The spectra measured by the two are roughly consistent in profile, but the former lacks many details and is much less accurate than the latter. The reason may be that the modulation frequency of 1MHz makes the tunnel junction capacitance C J The small capacitance of the circuit interferes with the STS spectrum measurement. It may also be due to the equivalent input current noise density of the circuit. This large circuit noise interferes with the STS spectrum measurement, or it is because when the DC bias is switched, the spectrum point measurement is started without waiting for the circuit to enter the steady state that can effectively measure the spectrum. When using this circuit to measure the differential conductivity spectrum line, when the DC bias is switched, the time it takes for the circuit to transition to the steady state that can effectively measure the spectrum (i.e., the "establishment time" of the spectrum measurement steady state) is quite long. There are two reasons for this. First, the LC resonator is an electromagnetic energy storage device, and its energy leakage and replenishment are much slower than pure capacitance. Second, this circuit does not have a feedback circuit like an op amp as an energy transmission channel (they temporarily add a resistor whose channel function is limited, but it increases the circuit noise); therefore, even if the increase in modulation frequency shortens the time for effective spectrum measurement, the "establishment time" of the spectrum measurement steady state is greatly extended, so that the total time to complete the entire spectrum line measurement is not shortened much. In summary, the circuit of Reference 5 does not have much advantage over the traditional TIA circuit in measuring STS spectrum.

[0020] To increase the bandwidth of the TIA circuit, if the feedback resistor R F Larger parasitic capacitance C in parallel with it F The influence of cannot be ignored. For example, if the feedback circuit is only a resistor R F , R F =1GΩ, C F is about 0.3pF, so its impedance is |Z F |=|R F / (1+j(2πf)C F R F )|equals The frequency is 530Hz, and the bandwidth of this feedback circuit is too small. Therefore, in the feedback circuit connected to the op amp in the TIA circuit, when the feedback resistor RF When it is large, the feedback resistor R F Connect a "bandwidth compensation network" composed of resistors and capacitors. The other end of the bandwidth compensation network is connected to the output of the op amp, and the feedback resistor R F The other end is connected to the inverting input of the op amp. There are many literatures reporting various structures of this feedback circuit with bandwidth compensation. The one reported in [Reference 6] below is the simplest and most commonly used one.

[0021] [Reference 6] The feedback circuit shown in "Low-temperature compatible IV converter" (Ultramicroscopy 42-44 (1992) 1647) is as follows: Figure 2 As shown, its structure is: the output terminal O of the op amp in the TIA is connected to a variable resistor R k , and then connect the feedback resistor R in series F , and then connected to the inverting input terminal N of the op amp, and then at R k With R F The connection point M is connected to a capacitor C c To the ground, here R k and R F There are parasitic capacitances, denoted by C k and C F , R k and C c They are called bandwidth compensation resistor and capacitor respectively. Assume that the output voltage of the op amp is Then a current flows to point N. Due to the "virtual short" effect of the op amp, the potential difference between the positive input terminal P and the negative input terminal N of the op amp is relative to the voltage The size can be considered to be 0, and the potential of the grounded positive input terminal P is 0. On this basis, the calculation is carried out to obtain:

[0022]

[0023] Take k as 10000~30000, take C c About kC F , so that R k In R F / k can be changed around, usually C F >C k , so R in the numerator of the second term on the right side of formula (f0-4) k C c R k C k At least 10,000 times larger, so

[0024]

[0025] If you adjust R k , so that R k C c =R F C F , then the above formula can be rewritten as

[0026]

[0027] In terms of equivalence, it can be considered is the impedance of the feedback circuit, denoted as Z F =Z F (f). Also note f F =1 / (2πR k C k ), then when f=f F Sometimes f F It can be called the bandwidth of the feedback circuit.

[0028] In reference 6, R F =10GΩ, C F About 0.35pF, R k =0.8MΩ, C k About 0.2pF, take C c =4.4nF, so that R k C c =R F C F , then f F =1 / (2πR k C k ) is about 1MH z； The experimental results they reported also showed that Z F =Z F (f) and R F In the frequency domain (0, 0.4MHz], it is close to the following degree: ||Z F (f) / R F | dB |≤1dB and -10°≤∠(Z F (f))≤0°.

[0029] By the way, if the TIA described in document 6 is used for STM spectrum measurement, its circuit will be unstable when the tunnel junction resistance is not particularly small; in addition, the JFET field effect transistor used in the circuit cannot operate at liquid helium temperature, and its offset voltage is too large; document 4 also mentioned that the TIA described in document 6 is not suitable for STM.

[0030] The feedback circuit of TIA in Reference 4 has a resistor R FThe "bandwidth compensation network" connected between them is much more complicated than that in Reference 6, and is also designed to make the impedance of the feedback circuit close to the pure resistance value R in a large frequency domain. F There are a lot of literatures introducing various bandwidth compensation networks of feedback circuits in TIA, which belong to the prior art and will not be described in detail here. Summary of the invention

[0031] The main purpose of the present invention is to provide a transimpedance amplifier circuit for measuring tunnel junction current related signals in a low-temperature STM, which can limit the increase of circuit noise in the high frequency band, so that the STM scanning and spectrum measurement work can be carried out based on a larger bandwidth, especially so that a higher operating frequency can be used when the STM spectrum is measured and excessive interference from circuit noise can be avoided, and the establishment time spent on establishing a spectrum measurement steady state is shortened (the establishment time of a DC bias power supply to establish a DC steady state is not included here), creating conditions for fast and high-precision STM spectrum measurement.

[0032] In the description of the technical solution of the present invention, certain special nouns, terms and symbols will be used, and their meanings are as agreed upon in the above “Background Technology”.

[0033] The basic idea of ​​the present invention is to use a "High Electron Mobility Field Effect Transistor" (HEMT) that can work at very low temperatures to assemble a "special preamplifier", place the HEMT in the "special preamplifier" in the low temperature zone where the tunnel junction is located, so that it is as close to the tip of the STM as possible, and then cascade the special preamplifier with very small output resistance with the "op amp" in the room temperature zone to form a high-gain "macro op amp", and then add a feedback circuit from its output end to the inverting input end, by reducing the capacitance C of the signal feeder connecting the HEMT and the tip I , to reduce the TIA circuit and C I The related noise can reduce the establishment time of spectrum measurement steady state by reducing the RC time constant of the related resistor-capacitor structure, and at the same time improve the gain of the TIA circuit in the high frequency band, thereby increasing the circuit bandwidth.

[0034] The basic contents of the technical solution of the present invention are described in (i) to (v) below.

[0035] (I) A transimpedance amplifier circuit for measuring tunnel junction current related signals in a low temperature STM, characterized in that:

[0036] (1) The circuit consists of four parts, namely, the "dedicated preamplifier" part, the "post-op amp circuit" part, the "feedback circuit" part, and the "signal input circuit" part;

[0037] (1.1) The "dedicated preamplifier" part, referred to as "dedicated preamplifier", includes two parts, namely, a "differential circuit" and a "constant current power supply" for supplying power to the differential circuit;

[0038] The structure of the "differential circuit" of the dedicated preamplifier is as follows: a pair of "high electron mobility field effect transistors" H1 and H2 with the same specifications are connected as follows: the two sources of the field effect transistors H1 and H2 are connected together, and then a resistor R g The other end of the resistor is grounded, and a capacitor C of more than 5μF is connected in parallel to the resistor. g The two drains of field effect transistors H1 and H2 are connected to drain load resistors R1 and R2 respectively, where R1 = R2 = R L , the other ends of R1 and R2 are denoted as L1 and L2 respectively, L1 and L2 are connected to capacitors C1 and C2 respectively, and the other ends of C1 and C2 are grounded, where C1=C2=C u , C u is 5μF or more; L1 and L2 are connected to a potentiometer R 12 The inactive ends at both ends of the potentiometer are used as the "power supply current input end" of the differential circuit. 12 The resistance between the two ends is R L / 10 or less; the two drains of the two field effect tubes H1 and H2 are respectively called the "positive phase output terminal" and "negative phase output terminal" of the dedicated preamplifier, and the two gates of H1 and H2 are respectively called the "negative phase input terminal" and "positive phase input terminal" of the dedicated preamplifier, and the positive phase input terminal is grounded;

[0039] The "constant current power supply" of the dedicated preamplifier includes a "current stabilization network" composed of transistors and resistors and a "positive voltage source" for supplying power to the current stabilization network. The negative electrode of the positive voltage source is grounded, and the positive electrode is connected to the current stabilization network. The "current output end" of the current stabilization network is connected to the power supply current input end of the differential circuit. The constant current is input to the differential circuit through the regulation of the current stabilization network. The differential circuit is only powered by this constant current power supply. The equivalent input capacitance of the preamplifier powered by a single power supply is recorded as C A , its equivalent input resistance is denoted as R A ;

[0040] (1.2) The “post-op amp circuit” is composed of a ready-made operational amplifier, called “post-OPA”, whose equivalent input capacitance and equivalent input resistance are denoted as C a and R a , connect a feedback resistor R from the output of the post OPA f To its inverting input, two cables are connected to the two inputs of the post-OPA respectively; the cable connected to the non-inverting input of the post-OPA has a capacitance to ground denoted as Ci1 The end of the cable is called the "positive input of the post-op amplifier circuit", which is connected to the positive output of the dedicated preamplifier; the cable connected to the inverting input of the post-OPA, its capacitance to ground is recorded as C i2 The end of the cable is called the "inverting input of the post-amplifier circuit", which is connected to the inverting output of the dedicated preamplifier; here C i1 =C i2 =C i ; The output end of the post OPA is the output end of the post op amp circuit;

[0041] The circuit formed by connecting the "dedicated preamplifier" and the "post-op amp circuit" is called a "macro operational amplifier", abbreviated as "macro op amp"; the inverting input terminal and the non-inverting input terminal of the dedicated preamplifier are respectively called the inverting input terminal and the non-inverting input terminal of this macro op amp, and the output terminal of the post-op amp circuit is also called the output terminal of this macro op amp; the voltage of the non-inverting input terminal of this macro op amp relative to the inverting input terminal is called the "input voltage of the macro op amp", and the voltage of the output terminal of this macro op amp relative to the ground is called the "output voltage of the macro op amp", and the ratio of the latter to the former is called the "open-loop voltage gain of the macro op amp", denoted as a A or a A (f);

[0042] (1.3) The “feedback circuit” is composed of: a feedback resistor R F A "bandwidth compensation network" consisting of a resistor and a capacitor is connected, and the other end of the bandwidth compensation network is connected to the output end of the macro op amp, and the feedback resistor R F The other end of the feedback circuit is connected to the inverting input of the macro op amp; the impedance of the feedback circuit as a function of the signal frequency f is recorded as Z F =Z F (f);

[0043] The circuit formed by connecting the macro op amp and the feedback circuit is called a "transimpedance amplifier", or "TIA" for short;

[0044] (1.4) The “signal input circuit” is composed as follows: a tunnel junction is formed between the tip of the STM and the sample, and a “bias-modulation signal voltage source” is abbreviated as BMS. One end of the BMS is grounded, and the other end is connected to the sample end of the tunnel junction. The tip of the tunnel junction is connected to a “signal feeder”. The other end of the signal feeder serves as the signal output end of the “signal input circuit” and is connected to the inverting input end of the macro op amp in the TIA. Here, the capacitance of the signal feeder to ground is denoted as C. I ;

[0045] The circuit formed by connecting the signal input circuit and the TIA is called a "TIA with input", also referred to as a "TIA circuit" or "transimpedance amplifier circuit";

[0046] (2) The tip-sample in the signal input circuit is in the low temperature zone set for it by the low temperature STM, all the field effect transistors in the differential circuit and the two drain load resistors R1 and R2, as well as the feedback resistor R in the feedback circuit F , are placed in the low temperature zone where the needle tip-sample is located.

[0047] (ii) The transimpedance amplifier circuit according to (i), characterized in that the structure-performance parameters of the macro operational amplifier, feedback circuit and signal input circuit meet the following conditions:

[0048] (1) For the open-loop voltage gain a of the macro op amp A =a A (f), whose modulus is denoted by |a A (f)|, and its phase angle is denoted as ∠(a A (f)), there is a frequency f in the frequency domain [5kHz, +∞) -60 Make ∠(a A (f -60 ))=-60°,at (0,f -60 ] has ∠(a A (f))≥-60°, and in [f -60 ,+∞) has |a A (f)|≤|a A (f -60 )|;

[0049] (2) The bandwidth compensation of the feedback circuit makes the frequency range [100Hz, f -60 ]Inside||Z F (f) / R F | dB |≤1dB;

[0050] (3) Let a1 = |a A (f -60 )|, let C=C I +C A , let f y =2a1 / (2πR F C), parameter f y So that the following inequality holds:

[0051] 1kHz≤f y ≤f -60 ;

[0052] (4) The frequency f of the modulation signal used by STM for spectrum measurement w ∈[100Hz,f y ];

[0053] (5)R A>10R F .

[0054] (III) The transimpedance amplifier circuit according to (I) or (II), characterized in that: the "constant current power supply" of the dedicated preamplifier has the following structure: two silicon-based PNP bipolar transistors T1 and T2 are used to form a "current stabilization network" in the following manner: a resistor R is connected between the emitter and base of transistor T1 b , and the collector of T1 is connected through a resistor R t The emitter of transistor T2 is connected to the base of T1, and the base of T2 is connected to the collector of T1; the collector of transistor T2 is connected to the power supply current input of the differential circuit as the current output terminal of the constant current power supply; the negative electrode of the "positive voltage source" that supplies power to the current stabilization network is grounded, and the positive electrode is connected to the emitter of transistor T1.

[0055] (IV) A transimpedance amplifier circuit for measuring tunnel junction current related signals in a low temperature STM, characterized in that: it is a partial replacement of the transimpedance amplifier circuit described in (I) or (II) above, that is, the "differential circuit" therein is replaced while the other parts remain unchanged, and the "differential circuit" therein is replaced with the following structure: four "high electron mobility field effect transistors" H11, H21, H12, and H22 of the same specifications are used and connected as follows: the drain of the field effect transistor H11 is connected to the source of the field effect transistor H12, the drain of the field effect transistor H21 is connected to the source of the field effect transistor H22, and the gates of the two field effect transistors H12 and H22 are respectively connected to the sources of the two field effect transistors H11 and H21; the two sources of the field effect transistors H11 and H21 are connected together, and then connected to a resistor R g The other end of the resistor is grounded, and a capacitor C of more than 5μF is connected in parallel to the resistor. g The two drains of the two field effect transistors H12 and H22 are connected to the drain load resistors R1 and R2 respectively, where R1 = R2 = R L , the other ends of R1 and R2 are denoted as L1 and L2 respectively, L1 and L2 are connected to capacitors C1 and C2 respectively, and the other ends of C1 and C2 are grounded, where C1=C2=C u , C u is 5μF or more; L1 and L2 are connected to a potentiometer R 12 The inactive ends at both ends, and the active end in the middle of the potentiometer is used as the "power supply current input end" of the differential circuit, connected to the current output end of the constant current power supply. 12 The resistance between the two ends is R L / 10 or less; the two drains of the two field effect transistors H12 and H22 are respectively called the "positive phase output terminal" and "negative phase output terminal" of the dedicated preamplifier, and the two gates of the two field effect transistors H11 and H21 are respectively called the "negative phase input terminal" and "positive phase input terminal" of the dedicated preamplifier, and the positive phase input terminal is grounded; the preamplifier formed by connecting the differential circuit with the constant current power supply has an equivalent input capacitance denoted as C A , its equivalent input resistance is denoted as R A .

[0056] (V) The transimpedance amplifier circuit according to (IV) is characterized in that: the structure of the "constant current power supply" is the same as the constant current power supply in the transimpedance amplifier circuit described in (III) above.

[0057] In the above technical solution, the circuit formed by connecting the "dedicated preamplifier" and the "post-op amp circuit" is actually a new operational amplifier, so it is called a "macro op amp", which can be compared to the op amp in the TIA circuit mentioned in the "background technology" (see Figure 1 ). The technical solution says that a special preamplifier should be formed by using a "high electron mobility field effect transistor" (HEMT), because this transistor can work at very low temperatures; the technical solution says that the HEMT should be placed in the low temperature area where the tip of the low-temperature STM-sample is located, which can shorten the signal feeder between the HEMT and the tip from one or two meters to one or two centimeters, making the capacitance C of the signal feeder I Therefore, the capacitor C I The related noise will be greatly reduced, or in other words, if the operating frequency of the STM is increased to a certain extent, the influence of the circuit noise will not increase much. From the formula (f0-1), it can be seen that the capacitor C I +C A The decrease in capacitance C makes the circuit gain not decrease much when the frequency f increases by a certain amplitude, which means that the bandwidth of the circuit increases. I +C A The decrease in J and R F The charging and discharging time is reduced, and the capacitor C i +C a The charging and discharging speed is very fast through the small output resistance of the differential circuit (and the small feedback resistance of the post-op-amp circuit), which shortens the time spent on establishing the spectrum measurement steady state. This creates conditions for the low-temperature STM to scan the spectrum quickly and with high precision.

[0058] In order to give full play to the positive benefits of the technical solution of the present invention, the technical solution is further elaborated below, and the beneficial effects brought about are further explained.

[0059] In the "Background Technology", it is mentioned that although it is limited by the tunnel junction capacitance, the operating frequency of STM scanning and spectrum measurement can still be increased to 10kHz~500kHz, which does not lose its significance due to the existence of tunnel junction capacitance. In the following design of TIA circuit, it is assumed that the bandwidth of the macro op amp should reach 0.5~1MHz, which requires that the two components of the macro op amp, the "dedicated preamplifier" and the "post op amp circuit", have at least the same bandwidth.

[0060] (I) The design of a dedicated preamplifier in a macro op amp is described below.

[0061] 1-1. First look at the "differential circuit" part of the dedicated preamplifier.

[0062] In the above technical solution, the reason why "high electron mobility field effect transistor" (HEMT) is selected to form a dedicated preamplifier is that this transistor can work at very low temperatures, such as the temperature of liquid helium 4.2K, or even below 0.5K, and its very low power consumption and heat generation when working at low temperatures will not affect the temperature distribution of the surrounding temperature field. Therefore, they can be placed very close to the needle tip-sample in the STM low temperature zone. The signal feed line from the needle tip to the HEMT gate can be as short as less than 2 cm. Instead of using a coaxial line, a gold wire is used, which can make the capacitance C of the signal feed line I In fact, there are various types of quartz crystals that can work normally below 4.2K and have stable transconductance g in the frequency domain of more than 30MHz. m and conductivity g d A HEMT can be selected as long as its gate-source capacitance C s and gate-drain capacitance C d Not too large, the bandwidth of the differential circuit made with them according to the described structure will generally reach above 10MHz.

[0063] In order to make the differential circuit of the dedicated preamplifier have super superior performance, a low-temperature HEMT developed by the French CNRS / LPN laboratory is particularly recommended, see [Reference 7] "Specific HEMTs for deep cryogenic high-impedance ultra low low-frequency noise read-out electronics" (2014 12th IEEEInternational Conference on Solid-State and Integrated Circuit Technology), hereinafter referred to as CNRS-HEMT. The first inventor of this patent has participated in the research and development of CNRS-HEMT. CNRS-HEMT is known as the low-temperature transistor with the lowest low-frequency noise in the world. It can work below 0.5K and its power consumption is only 0.1mW. The specific performance parameters are shown in Table 2, where the source-drain voltage V ds and source-drain current I ds is the commonly used operating point of this transistor, and They represent the equivalent input voltage noise density and equivalent input current noise density of the transistor respectively. Their corresponding values ​​at frequencies f=10kHz and f=100kHz are listed in the table.

[0064] Another HEMT developed by CNRS / LPN in France is recommended, namely PHEMT, see [Reference 8] "Investigation on the low-power and low-frequency noise performance of pHEMT at 4.2K" (J.Phys.IV France 12 (2002)), hereinafter referred to as CNRS-PHEMT. It can operate at a temperature of 4.2K, and its power consumption and noise are slightly larger than those of CNRS-HEMT. The specific performance parameters are shown in Table 2.

[0065] Table 2 Performance parameters of CNRS-HEMT and CNRS-PHEMT

[0066]

[0067] In the following specific embodiments, these two types of HEMTs will be mainly used to form the differential circuit, and the bandwidth of the differential circuit can reach more than 30MHz. The bandwidth of the "post-op amp circuit" cascaded after the preamplifier is generally below 1MHz. Since the bandwidth of the preamplifier is much greater than 1MHz, the bandwidth of the entire macro op amp is determined by the bandwidth of the post-op amp circuit. The voltage of the preamplifier's positive input terminal relative to the negative input terminal is called the preamplifier's input voltage, and the voltage of the preamplifier's positive output terminal relative to the negative output terminal is called the preamplifier's output voltage. The ratio of the latter to the former is called the preamplifier's voltage gain, denoted as A. vP ; The voltage gain of the front-end with a bandwidth of more than 10MHz in the frequency domain below 1MHz is A vP It can be regarded as a constant that does not change with frequency.

[0068] The dedicated preamplifier of the TIA circuit described in technical solutions (I), (II), and (III), wherein the "differential circuit" is composed of two HEMTs, is called a "HEMT×2 differential circuit". A simple calculation shows that the voltage gain A of the preamplifier is vP The constants are:

[0069] A vP =g m / (1 / R L +g d ), (f1-1)

[0070] The equivalent input capacitance C of the preamplifier A for

[0071] C A =C s +(1+A vP )C d ; (f1-2)

[0072] The equivalent input resistance of the preamplifier is R A It is actually equal to the input resistance of a single HEMT, which is generally 50GΩ~100TΩ, R A >>10R F , so it can be regarded as infinite and its influence can be ignored; in addition, the output resistance R of the preamplifier is calculated d for

[0073] R d =1 / (1 / R L +g d ), (f1-3)

[0074] R d It is also the "signal source resistance" of the post-op amp circuit cascaded behind the preamplifier. For example, R L =1kΩ CNRS-HEMT×2 differential circuit A vP=20, C A =26pF, R d =500Ω.

[0075] The dedicated preamplifier of the TIA circuit described in technical solution (IV) (V), wherein the "differential circuit" is composed of "a pair of HEMT dual tubes connected in series", is called "HEMT×4 differential circuit". A simple calculation shows that the voltage gain of the preamplifier is

[0076] A vP =g m R L ; (f1-4)

[0077] The equivalent input capacitance of the preamplifier is

[0078] C A =C s +(2+g d R L )C d ; (f1-5)

[0079] The equivalent input resistance of the preamplifier is R A is actually equal to the input resistance of a single HEMT, since R A >>10R F , its influence can be ignored; in addition, the output resistance of the preamplifier is

[0080] R d =R L , (f1-6)

[0081] R d It is also the "signal source resistance" of the post-op amp circuit cascaded behind the preamplifier. For example, R L = 0.5kΩ CNRS-HEMT × 4 differential circuit A vP =20, C A =7.5pF, R d =500Ω.

[0082] The biggest advantage of the HEMT×4 differential circuit is that it eliminates the Miller effect of the capacitor, making the equivalent input capacitance C of the amplifier A For example, compared with the CNRS-HEMT×2 differential circuit, C A = 26pF, the CNRS-HEMT×4 differential circuit is reduced to C A =7.5pF. C A The significant reduction of the TIA circuit and C A The related noise is reduced (see (VI) and (VII) below); in addition, the capacitance C = C I +C AThe reduction of C=C reduces the time for charging and discharging through the resistor, thereby reducing the time for establishing the effective spectrum measurement state (see (VIII) below); in particular, the capacitance C=C I +C A The reduction of the feedback resistor R F It can take a larger value, for example, R F =1GΩ increases to R F =3GΩ, which makes the transimpedance gain of the entire TIA circuit larger, so that the STM can measure a larger tunnel junction differential resistance R J (See (V) below).

[0083] From the above formula, we know that after selecting HEMT, we need to select the drain load resistor R L , you can adjust the voltage gain A vP and C A , R d Adjust the resistor R connected to the HEMT source g , that is, the gate bias and the static operating point of the HEMT can be adjusted. In the following specific implementation, the specific parameters of each component in the differential circuit will be given as an example. For general technicians working in the field of "op amp and application design", according to the calculation formula and following the example, the specific design and production of the differential circuit can be completed without creative labor.

[0084] 1-2. Let’s look at the “constant current power supply” section in the dedicated preamplifier.

[0085] The dedicated preamplifier is powered by a single power supply, that is, a "constant current power supply" including a positive voltage source and a current stabilization network is connected to the "power supply current input terminal" of the differential circuit to realize power supply. As long as the DC current output by the "constant current power supply" is constant, the DC potential of the drain and source of each HEMT remains constant when the potentials of the inverting and non-inverting input terminals of the differential circuit remain unchanged. Various "constant current power supplies" can be used here, and the one described in technical solution (III) is a very common one. The differential circuit of the differential amplifier of the traditional structure is powered by a dual power supply: the source of the two field effect tubes in the differential circuit is connected to the constant current power supply to obtain the negative voltage provided, and the drain load resistor connected to the drain of the two field effect tubes is connected to the positive voltage stabilization source; in this structure, the voltage fluctuation of the positive voltage stabilization source will directly affect the potential of the drain, making the fluctuation of the potential of the drain almost as large as the fluctuation of the power supply. However, in the present invention, only a single "constant current power supply" is used to power the differential circuit. Even if the positive voltage source in the constant current power supply that supplies power to the current stabilization network has a large voltage fluctuation, the DC current output by the current stabilization network remains almost unchanged. For example, the constant current power supply described in technical solution (III), in which the silicon-based PNP bipolar transistor T1 is connected to a DC voltage V between the emitter and the base. ebis the “on-state voltage”, which is approximately 0.7V. In the current stabilization network, there is a feedback mechanism (due to the resistor R t and R b ), so that the DC voltage between the emitter and base of T1 is stably maintained at V eb A fixed value V′ near eb , so it flows through the resistor R between the emitter and the base b The DC current V' eb / R b Stable and unchanged, it is basically the DC current output to the differential circuit through transistor T2. The stability of the DC current stabilizes the operating voltage of the HEMT. The simulation shows that even if the voltage fluctuation of the positive voltage source supplying power to the current stabilization network reaches 10mV, the fluctuation of the source-drain voltage of the HEMT will not exceed 1mV. This small voltage fluctuation is very important for the normal operation of the HEMT with a source-drain voltage of only 100-300mV. The temperature of the transistor in the current stabilization network can also be controlled by a temperature control device TEC to make the temperature fluctuation less than 0.5℃, which will achieve a better current stabilization effect.

[0086] The positive voltage source in the constant current power supply that supplies power to the current stabilization network is a noise source, and the noise generated by it is about The current-stabilizing network composed of transistors connected in series with the positive voltage source has a DC resistance of about several hundred ohms, but its micro-variable resistance is usually as high as 100,000 ohms or more; this micro-variable resistor and the two large capacitors C1 and C2 (the other ends of the two capacitors are grounded) connected to the power supply current input end of the differential circuit form a low-pass filter, so that the high-frequency components of the noise generated by the positive voltage source are basically filtered out, and not much noise current is input to the differential circuit. The positive voltage source of the traditional differential amplifier is also through a resistor R p It is connected to the differential circuit to realize power supply, and a large capacitor of more than 5μF is also connected to the ground at the connection point, but in order to reduce the voltage drop, this resistor R pIt cannot be too large, and is only a few thousand ohms at most. Therefore, the low-pass filter formed has a very weak filtering effect on the high-frequency components of the noise. The noise that is not filtered out is a common-mode signal for the differential circuit. Theoretically, if the two branches of the differential circuit are completely symmetrical, the common-mode noise signal has no effect on the output of the differential circuit. However, if the differential circuit is composed of HEMT, due to the characteristics of the HEMT structure and preparation process, even HEMT devices made in the same batch have very different performances, which makes the input common-mode noise signal The impact on the useful signal output by the differential circuit cannot be ignored. However, in the present invention, the dedicated preamplifier is only powered by a constant current power supply, and the powerful filtering of the low-pass filter prevents the common-mode noise signal generated by the power supply from entering the differential circuit, thus avoiding its influence from the root, and the symmetry of the differential circuit does not need to be particularly demanding. In addition, the resistor R connected to the source of the HEMT of the dedicated preamplifier here g A large capacitor C of more than 5μF is also connected in parallel g , which makes the resistor R g The noise generated is also basically filtered out. It should also be pointed out that the special preamplifier here is not as good as the traditional differential amplifier in common mode rejection of common mode noise signals from the two input terminals, but the two input terminals here, one is connected to the STM needle tip with a short feed line, and the other is grounded, which does not attract much common mode noise signals.

[0087] It should be pointed out that the source-drain voltage of HEMTs such as CNRS-HEMT is only 0.1V. HEMTs are very sensitive to voltage surges and there is a freezing problem of electrons captured by the DX center during the cooling process. If it is used to make a traditional dual-power supply differential amplifier, the positive and negative power supplies need to be synchronously boosted according to the source and drain potentials of the two HEMTs when powered on, involving many monitoring instruments and moving parts. The voltage adjustment is very inconvenient and has poor repeatability. Minor abnormalities may cause the HEMT to burn out, and it is difficult to expect additional protection circuits to solve the problem because this will greatly increase the circuit noise. The present invention adopts a single power supply in the dedicated preamplifier, and one important reason is to solve the above problems.

[0088] (II) The design of the post-op amp circuit in the macro op amp is as follows.

[0089] The "post-op amp circuit" described in the technical solution, in which the "post-OPA" used, can be selected from a large number of existing operational amplifier products with high "gain-bandwidth product" and low offset current and voltage, such as TI's THS4021, OPA657, LMH6624, OPA846, OPA847, etc., all of which have a gain-bandwidth product of more than 1GHz and other good performance. In the technical solution, the structure of the post-op amp circuit has been given. It is a general op amp circuit with nothing special. The following will give the calculation of its voltage gain A from its various component parameters. va The formula is given, and the specific values ​​of the component parameters will be given in the specific implementation mode below as an embodiment; therefore, for a general technician in the relevant field, according to the calculation formula and following the embodiment, it is easy to design and make such a circuit.

[0090] The voltage of the non-inverting input terminal of the post-op amplifier circuit relative to the inverting input terminal is called its input voltage, and the voltage of its output terminal relative to the ground is called its output voltage. The ratio of the latter to the former is its voltage gain A. va , the output resistance of the preamplifier R d is the signal source resistance of the post-op amp circuit. A simple calculation shows that the voltage gain A va for

[0091]

[0092] Among them, a a is the open-loop voltage gain of the post-OPA, R a and C a The equivalent input resistance and capacitance of the post-OPA are respectively. The capacitance of the two cables connected to the preamplifier from the positive and negative input terminals of the post-OPA to the ground is C i1 and C i2 , C i1 =C i2 =C i , R f R is the feedback resistor connecting the output of the post OPA to its inverting input. f It is in the MΩ level, which is larger than the large feedback resistor R in the TIA circuit. F 3 orders of magnitude smaller, R f There is a parasitic capacitance of 0.1pF, which affects R f It can be like R F A bandwidth compensation network consisting of resistors and capacitors is connected to offset the influence of its parasitic capacitance, making the feedback circuit basically equivalent to a pure resistor R f However, for R f No compensation is required, because R fThe existence of parasitic capacitance is beneficial to the stability of the post-op-amp circuit, but it will increase the feedback at the high frequency end, resulting in a decrease in the voltage gain of the post-op-amp circuit at the high frequency end. A detailed description of this situation can be found in Section 6.4 of Reference 1. Generally, R d is several hundred ohms, R a R d 20 to 10,000 times larger, so the term 2R in the denominator of (f2-1) f / (a a R a ) relative to R f / (a a R d ) can be omitted. The post-OPA usually selected, such as TI's THS4021, OPA657, LMH6624, etc., has an open-loop voltage gain a as can be seen from the product introduction. a In a wide frequency band, the surface can be approximated as a a =a a0 / (1+jf / f b ), so formula (f2-1) can be expressed as

[0093]

[0094] Replace j2πf with s, record ω b =2πf b , from the denominator of formula (f2-2) we can get A va The two extreme points s1 and s2 are:

[0095]

[0096] Adjust the parameter R in the subtrahend within the square root of the formula (f2-3) f The subtrahend can be made not to exceed the minuend, so that the two poles s1 and s2 are negative real numbers. The subtrahend can also be made closer to the minuend, so that f1=―s1 / (2π) and f2=―s2 / (2π) are both far from 0, which helps to increase the bandwidth of the macro op amp. If the subtrahend within the square root is larger than the minuend, the two poles s1 and s2 are two conjugate imaginary numbers, which may cause the transimpedance amplifier circuit to have "ringing" or step response "overshoot". Although it is still possible to satisfy the stability conditions described in technical solution (II), if the "ringing" or "overshoot" is more serious, it will affect the accuracy of the STM measurement and prolong the time it takes for the circuit to enter the effective measurement steady state. The description of this situation can be found in Chapter 8 of Reference 1.

[0097] (III) The composition of the macro op amp is as follows.

[0098] The macro op amp is formed by cascading the preamplifier and the post-amplifier circuit. The differential circuit in the preamplifier has two types: HEMT×2 and HEMT×4. Therefore, the corresponding macro op amps are called "macro op amp containing HEMT×2" and "macro op amp containing HEMT×4" respectively. The open-loop voltage gain of the macro op amp is a A is the voltage gain of the preamplifier A vP And the voltage gain A of the post op amp circuit va The product of A =A vP A va . Open loop voltage gain a A =a A (f) can be calculated as a function of the signal frequency f, or it can be simulated using software. In addition, the equivalent input capacitance C of the preamplifier A and resistor R A It is the equivalent input capacitance and resistance (R A Very large, its influence can be ignored); the voltage gain A of the post-op amp circuit calculated in (II) va The two poles s1 and s2 are also the open-loop voltage gain a of the macro op amp. A The extreme point.

[0099] Since the two output ends of the differential circuit of the preamplifier have DC common-mode voltage output, even if the potentials of the positive and negative input ends of the differential circuit are both 0, the output end of the post-op amp circuit may still have a non-zero DC voltage output. For this purpose, a potentiometer R 12 To adjust the DC potential at the output of the post-op amplifier circuit to zero. In addition, the feedback circuit in the entire TIA circuit also has the function of maintaining the DC potential at the output of the post-op amplifier circuit to zero.

[0100] In addition, if the maximum output voltage of the aforementioned macro op amp is not large enough, another op amp can be added to make a new macro op amp.

[0101] (IV) Regarding the bandwidth compensation of the feedback circuit, see below.

[0102] In the technical solution, the "feedback circuit" in the TIA is composed of a feedback resistor R F A "bandwidth compensation network" composed of resistors and capacitors is connected between the output terminal and the inverting input terminal of the macro op amp. As explained in the previous "background technology", the bandwidth compensation of this feedback circuit belongs to the prior art. There are a large number of documents introducing various specific structures, most of which can be used in this technical solution, such as those described in Documents 4 and 6. The structure described in Document 6 is the simplest and most commonly used one, which is repeated as follows: Figure 2 As shown, the output terminal O of the op amp in the TIA is connected to a variable resistor Rk , and then connect the feedback resistor R in series F , and then connected to the inverting input terminal N of the op amp, and then at R k With R F The connection point M is connected to a capacitor C c To the ground, R k and R F The parasitic capacitances are denoted as C k and C F ; Take k as 10000~30000, take C c About kC F , so that R k In R F / k is variable, adjust R k Make R k C c =R F C F , the equivalent impedance Z of the feedback circuit F =Z F (f) The formula is given in “Background Technology”:

[0103]

[0104] Remember f F =1 / (2πR k C k ), then when f=f F Sometimes So we call f F is the bandwidth of the feedback circuit.

[0105] In reference 6, R F =10GΩ, C F About 0.35pF, R k =0.8MΩ, C k About 0.2pF, take C c =4.4nF, so that R k C c =R F C F , then f F =1 / (2πR k C k ) is about 1MHz; their reported experimental results also show that Z F =Z F (f) and R F In the frequency domain (0, 0.4MHz], it is close to the following degree: ||Z F (f) / R F | dB |≤1dB and -10°≤∠(Z F (f))≤0°.

[0106] In STM, the tunnel junction differential resistance R is usually measured. J The highest is 1~3GΩ, and R F Generally not less than R J If the structure described in Reference 6 is used, the compensation resistor R k is about 100kΩ, and its parasitic capacitance C k Generally it does not exceed 0.2pF, so f F =1 / (2πR k C k ) is above 8MHz, so it can be realized in the frequency domain (0,3MHz]||Z F (f) / R F | dB |≤1dB and -10°≤∠(Z F (f))≤0°; if we examine it in the frequency domain (0,1MHz], we have And |Z F (f)|≤R F , so it can be considered that Z F Equal to R F .

[0107] (V) The stability of the transimpedance amplifier circuit is described below.

[0108] Technical solution (II) shows that the structure-performance parameters of the TIA circuit must meet the five conditions (1) to (5). When designing the TIA circuit, only a series of operations are required to achieve these conditions. Once these conditions are achieved, the entire TIA circuit can be guaranteed to have sufficient stability, as described in detail below.

[0109] According to the method for determining the stability of the TIA circuit mentioned in the “Background Technology”, the loop gain of the TIA circuit here is

[0110] T L (f) = a A (f)β(f)=a A (f) / (1 / β(f)), (f5-1)

[0111] Where β(f) is the return ratio factor, and its reciprocal is

[0112] in

[0113] The open-loop voltage gain of the macro op amp is a A =a A (f) can be calculated, simulated or measured; it can be obtained that ∠(a A(f)) = frequency f of -60° -60 .f -60 This is roughly the upper cutoff frequency of the macro op amp. -60 The circuit is designed for 0.5 to 1 MHz.

[0114] Generally, the object under test R J =1 / G J No more than R F , and not small R F / 1000, corresponding to the usual measured object R J The range of R F It is usually taken as 200MΩ~3GΩ. I , C A Correlated noise, C I and C A The smaller the better, the signal feeder capacitance C I can be reduced to below 0.5pF, C A Generally 5 to 30 pF, so C = C I +C A ≈C A In addition, the input resistance of the preamplifier R A Generally, it is 50GΩ~100TΩ, R A >10R F , so R A >10R J (That is, condition (ii) (5) is satisfied), so the 1 / R in the brackets in the first equation in (f5-2) is A Much smaller than 1 / R J , we can convert 1 / R A As described in (IV) above, the resistor R used for compensation in the feedback circuit k It can be taken as about 100kΩ, and its parasitic capacitance C k does not exceed 0.2pF, so R k C k ≤2×10 -8 s, calculated according to the latter formula in (f5-2), in the frequency domain (0,1MHz], we have And |Z F (f)|≤R F , so it can be considered that Z F Equal to R F . Because f -60 It is set to 0.5MHz~1MHz, so it can be considered that in (0,f -60 ] Up Z F Equal to R F , so in (0,f -60 ]

[0115]

[0116] In addition, from (f5-2) we know Where R F C is of the order of 10 -2 s, than R k C k ≤2×10 -8 s is 5 orders of magnitude larger, so |1 / β(f)| in [f -60 ,+∞) increases monotonically with respect to f. Strictly speaking, when the frequency f is greater than 10MHz, C A It is no longer constant but slightly reduced, but in general, at least in [f -60 ,10f -60 ]On this basis, |1 / β(f)| is monotonically increasing with respect to f.

[0117] According to condition (3) in technical solution (II), y =2a1 / (2πR F C) and f y ≤f 60 , and formula (f5-3) shows

[0118]

[0119] Thus, a1=|a A (f -60 )|Know

[0120]

[0121] Condition (II) (1) When it comes to [f -60 ,+∞) has |a A (f)|≤|a A (f -60 )| (no special design is required, this condition will generally be achieved naturally), and |1 / β(f)| in [f -60 ,10f -60 ] is monotonically increasing with respect to f, so from formula (f5-4) we know that in [f -60 ,10f -60 ]

[0122] |T L (f)| dB =|a A (f)| dB -|1 / β(f)| dB ≤|a A (f -60 )| dB -|1 / β(f -60 )|dB =|T L (f -60 )| dB <-6dB,

[0123] In general, when f ≥ 10f -60 Time|a A (f) |At least greater than a1 = |a A (f -60 )| is 20dB lower, and |1 / β(f)| will not be lower than |1 / β(f -60 )|, so in [f -60 ,10f -60 ]∪(10f -60 ,+∞)=[f -60 ,+∞)

[0124] |T L (f)| dB =|a A (f)| dB -|1 / β(f)| dB ≤|a A (f -60 )| dB -|1 / β(f -60 )| dB =|T L (f -60 )| dB <-6dB, (f5-5)

[0125] That is, the TIA circuit is in [f -60 ,+∞), the amplitude margin of its stability is greater than 6dB.

[0126] On the other hand, condition (ii) (1) says that in (0,f -60 ] has ∠(a A (f))≥-60° (no special design is required, this condition will generally be achieved naturally), so from formula (f5-3) we know that at (0,f -60 ]

[0127] ∠(T L (f))=∠(a A (f))-∠(1 / β(f))>-60°-90°=-150°, (f5-6)

[0128] That is, the TIA circuit is in (0,f -60 ] Its stability angular margin is greater than 30°.

[0129] From the conclusions of equations (f5-5) and (f5-6), it can be determined that the TIA circuit has sufficient stability. Figure 4 An intuitive representation of .

[0130] Condition (2) of the technical solution (II) states that “the bandwidth compensation of the feedback circuit is such that in the frequency range [100Hz, f -60 ]Inside||Z F (f) / R F | dB |≤1dB", this condition is more stringent than the above "at (0,f -60 ] Up Z F Equal to R F " is a little weaker, but it can still be considered to ensure a certain degree of stability of the TIA circuit.

[0131] To ensure stability, you need to select R F C and a1 make condition (ii) (3) 1kHz≤f y =2a1 / (2πR F C)≤f -60 is satisfied, you can select R F C and then consider choosing a1=|a A (f -60 )| to achieve this condition, the selected a1 can be obtained by selecting the feedback resistor R in the post-op amplifier circuit f To adjust the voltage gain A of the circuit va and a A =A vP A va to achieve.

[0132] For example, for a TIA circuit containing a CNRS-HEMT×2 differential circuit, its C A =26pF, assuming that the capacitance C I has been reduced to below 0.5pF, the feedback resistor R F Press R J It is usually necessary to take 1GΩ. If a1=2.5×10 4 , so that f y =310kHz, so that the modulation signal frequency f of STM spectrum measurement can be w For another example, for a TIA circuit including a CNRS-HEMT×4 differential circuit, its C A =7.5pF, to maintain f y =2a1 / (2πR F C) and a1 are the same as in the previous example, and R F =3GΩ; Larger R FThis means that the transimpedance gain of the TIA circuit is large, which enables the STM to measure a larger tunnel junction differential resistance R J .

[0133] (VI) About the equivalent input voltage noise density of the macro op amp and equivalent input current noise density See below.

[0134] The macro op amp is formed by cascading a preamplifier and a post op amp circuit. Depending on whether the differential circuit in the preamplifier is a "HEMT×2 differential circuit" or a "HEMT×4 differential circuit", they are respectively called "macro op amp including HEMT×2" and "macro op amp including HEMT×4".

[0135] The equivalent input voltage noise density of the macro op amp containing HEMT×2 is and equivalent input current noise density The calculation formula is as follows:

[0136]

[0137]

[0138] in, and They are the equivalent input noise voltage PSD and equivalent input noise current PSD of a single HEMT (PSD is the abbreviation of "power spectral density"), and They are the equivalent input noise voltage PSD and equivalent input noise current PSD of the post-OPA. In the post-op amplifier circuit here, the general feedback resistor R f MΩ level, R f Temperature T f =300K, C i About 100pF, C i +C a ≈C i The square root in formulas (f6-1) and (f6-2) is If we calculate more precisely, One more thing should be added For the working frequency f below 300kHz used in STM spectrum measurement, C i The capacitive reactance is 1 / (2πfC i ) is greater than 5kΩ, and R d It is only a few hundred ohms, so in both equations No additional The relative error caused by this item is less than 1 / 100.

[0139] If the post OPA uses TI's THS4021, LMH6624 or other similar op amps, then and G m It is about 30~60mS, from which we can infer that and Respectively At least 1 and 2 orders of magnitude smaller, while Compare At least 3 orders of magnitude smaller, so formulas (f6-1) and (f6-2) can be approximated as:

[0140]

[0141]

[0142] The above approximation reflects the macro op amp and The contribution of the front-stage amplifier circuit is mainly due to the noise, and the contribution of the rear-stage amplifier circuit can be ignored. In other words, the noise generated by the rear-stage circuit has a very small impact on the useful signal amplified by the front-stage circuit.

[0143] The equivalent input voltage noise density of the macro op amp containing HEMT×4 is and equivalent input current noise density The calculation formula of (f6-1) and (f6-2) is different from that of (f6-1) and (f6-2). d It needs to be multiplied by 2 again, so the calculation formula can also be approximated as (f6-3) and (f6-4).

[0144] If the HEMT used is CNRS-HEMT, Table 2 shows that at f = 10kHz and For the macro op amp containing “CNRS-HEMT×2”, its C A =26pF, we can see that at f=10kHz and For the macro op amp containing “CNRS-HEMT×4”, its C A Reduced to 7.5pF, it can be seen that at f = 10kHz and It is slightly less than the macro op amp containing “CNRS-HEMT×2”.

[0145] The macro op amp in the TIA circuit in Reference 4 is composed of three cascaded op amps (the macro op amp does not include the feedback circuit and signal input circuit in the TIA circuit). and It is almost the same as the first-stage op amp AD8615. From the product manual, we know that at f = 10kHz and The noise characteristics of this macro op amp are much worse than those of the macro op amp including "CNRS-HEMT×2".

[0146] (VII) Equivalent input current noise density of transimpedance amplifier circuit See below.

[0147] The equivalent input noise current of the so-called "transimpedance amplifier circuit" is defined as follows: Remove the "bias-modulation signal voltage source" in the circuit and short-circuit its two input terminals, leaving the tunnel junction as the resistance R in the circuit. J , but it is not considered as a noise source when calculating the circuit noise (because the noise generated by the tunnel junction is a useful signal). In this case, the total noise voltage generated by all noise sources in the circuit at the output end of the circuit is obtained. This total noise voltage is equivalent to the noise voltage generated by a virtual noise current source with the two ends A and B of the tunnel junction as input ends at the output end of the circuit (when making this conversion, it is assumed that there is only this virtual noise current source in the circuit, and all other noise sources are removed). This virtual noise current source introduced in this way (with R J is the internal resistance of the current source), and the noise current generated is called the equivalent input noise current of the TIA circuit. The equivalent input noise current power spectrum density of the TIA circuit is recorded as Its square root It is called the equivalent input current noise density. People pay attention to this equivalent input noise current because it can be easily compared with the useful signal current flowing through the tunnel junction to confirm the degree of interference. For details, please refer to [Reference 9] "Electronic Noise and Low Noise Design" (edited by Gao Jinzhan, published by Tsinghua University Press in 2016).

[0148] For a TIA circuit containing "CNRS-HEMT×2", its equivalent input noise current PSD The calculation formula is as follows

[0149]

[0150] Where T is R F For a TIA circuit containing "CNRS-HEMT×4", it is only necessary to replace C d Multiply by 2. Ignore the small amount contributed by the post-op amp circuit, and use the approximate expressions of formulas (f6-3) and (f6-4), and we get

[0151]

[0152] From (f7-2), we can see that the equivalent input current noise density of the TIA circuit is There are four basic ingredients: Respectively referred to as "and capacitor C I The noise density component related to the input current noise density of the macro op amp is equal to the component of the equivalent input current noise density of the macro op amp, and the feedback resistor R F The noise density component contributed by the resistor R J Correlated noise density components".

[0153] For a TIA circuit including “CNRS-HEMT×2”, its C A =26pF, if C I =0.5pF, R F =1GΩ, R J =0.001R F , T = 4.2K, calculation shows that at f = 10kHz Total

[0154] For a TIA circuit including “CNRS-HEMT×4”, its C A =7.5pF, if C I =0.5pF, R F =3GΩ, R J =0.001R F , T = 4.2K, calculation shows that at f = 10kHz Total Its noise is half that of the TIA circuit containing "CNRS-HEMT×2".

[0155] For the TIA circuit in Reference 4, the calculation can still be performed according to formula (f7-2), where C A =6.7pF, C I =35pF, R F =1GΩ, let R J =0.001R F , and TIA and R F The temperature is T = 80K, and we know that at f = 10kHz Total If we ignore the resistance R J The noise density components associated with the J >0.01R F In the case of This is consistent with the literature 4 Figure 8The experimental results shown are the same. Compared with the TIA circuit including "CNRS-HEMT×2", the TIA circuit of Document 4 has 15 times more noise.

[0156] From the above description, it can be seen that the use of the TIA circuit of the present invention in a low-temperature STM can significantly reduce circuit noise while increasing the operating frequency, which enables the STM to complete scanning and spectrum measurement at high speed and high precision.

[0157] (VIII) The establishment time for entering the effective spectrum measurement state is as follows.

[0158] When measuring STS spectrum with STM, each time the bias voltage of the "bias-modulation signal voltage source" is changed, it needs to quickly transition to the effective spectrum measurement state in order to implement effective spectrum measurement based on the modulation signal. The time to establish the effective spectrum measurement state depends on the charging and discharging time of the energy storage capacitor. Capacitor C A Determined by the differential circuit, such as C of the "CNRS-HEMT × 2 differential circuit" A is 26pF, while the C A is 7.5 pF. The capacitance C of the signal feeder in the present invention is I can be reduced to less than 0.5pF, which makes C = C I +C A Greatly reduced (C≈C A ), so that the capacitor C passes through R J and R F The charging and discharging time is reduced. In addition, in the post-op amp circuit, the capacitor C i +C a The signal source resistor R d and feedback resistor R f The resistance is charged and discharged. Since the resistance here is very small, the charging and discharging speed is very fast. In the following specific implementation, the response of the TIA circuit to the step signal will be used to illustrate the speed of capacitor charging and discharging. The simulation shows that the step response reaches an adjustment time t with an error of less than 0.1%. s It is approximately less than 4μs, which is almost equal to the period of the modulation signal. BRIEF DESCRIPTION OF THE DRAWINGS

[0159] Figure 1 It is a schematic diagram of the structure and working principle of the transimpedance amplifier circuit in STM.

[0160] Figure 2 This is a diagram illustrating the bandwidth compensation of the feedback circuit in the transimpedance amplifier circuit.

[0161] Figure 3 This is a circuit diagram of a TIA circuit including a HEMT×2 differential circuit.

[0162] Figure 4 yes Figure 3 Circuit embodiment ① is a diagram illustrating the open-loop voltage gain of the macro op amp and the stability of the TIA circuit.

[0163] Figure 5 yes Figure 3 Circuit embodiment ① is an illustration of the voltage gain of the TIA circuit.

[0164] Figure 6 yes Figure 3 Embodiment ① of the circuit is an illustration of the response of the TIA circuit to a step signal.

[0165] Figure 7 yes Figure 3 Embodiment ④ of the circuit is a diagram illustrating the open-loop voltage gain of the macro op amp and the stability of the TIA circuit.

[0166] Figure 8 This is a circuit diagram of a dedicated preamplifier including a HEMT×4 differential circuit.

[0167] Fig. 9 is included Figure 8 Embodiment ⑥ of the preamplifier is an illustration of the open-loop voltage gain of the macro op amp and the stability of the TIA circuit.

[0168] Fig.10 is included Figure 8 An illustration of the voltage gain of the TIA circuit of the preamplifier embodiment ⑥.

[0169] Fig.11 is included Figure 8 An illustration of the response of the TIA circuit of the preamplifier embodiment ⑥ to a step signal. DETAILED DESCRIPTION

[0170] The following describes the specific implementation of the technical solution of the present invention in conjunction with the accompanying drawings.

[0171] (I) First describe the specific implementation methods of technical solutions (I), (II), and (III).

[0172] Figure 3 This is a circuit diagram of a TIA circuit including a HEMT×2 differential circuit. Figure 3In the figure, the dotted box 1.1 is the HEMT×2 differential circuit in the dedicated preamplifier, and the dotted box 1.2 is the constant current power supply circuit of the dedicated preamplifier. The current output of the constant current power supply is connected to the power supply current input of the differential circuit, forming a dedicated preamplifier; the inverting input of the dedicated preamplifier in the figure is point S, which is also the inverting input of the macro op amp. The dotted box 2 is the post-op amp circuit, which includes two cables. These two cables connect the positive and negative inputs of the post-OPA to the positive output O1 and negative output O2 of the preamplifier respectively. The capacitance of the two cables to ground is C i1 and C i2 , C i1 =C i2 =C i The dotted box 3 is the feedback circuit in the TIA, where C F and C k Respectively represent the feedback resistance R F and bandwidth compensation resistor R k The parasitic capacitance of R F and R k The connection point is also connected to the bandwidth compensation capacitor C c to ground; the feedback circuit connects the output terminal O of the post-OPA and the inverting input terminal S of the preamplifier. The feedback circuit here can also be replaced with the structure described in document 4. The dotted box 4 is the signal input circuit, where the tunnel junction TJ formed by the STM tip and sample is between points A and B, and points B and G are the input ports of the "bias-modulation signal voltage source" (i.e., BMS). Here, BMS is represented by a DC bias source in series with a modulation signal voltage source, but it may also be other ways to generate the superposition of two signal voltages. Point G is grounded, and the tip of the STM is connected to the inverting input terminal S of the preamplifier through a signal feeder. The capacitance of the signal feeder to the ground is C I In addition, the dashed box 5 shows a measuring instrument (such as a lock-in amplifier) ​​connected to the output terminal O of the post-OPA via a coaxial line. o and capacitor C o In parallel and with one end grounded, the resistor R o and capacitor C o They are approximately 10MΩ and 100pF respectively. In particular, the equivalent input capacitance C of the preamplifier is A It is related to the capacitance parameters inside the HEMT. Here it is drawn as an independent capacitor outside the HEMT. The equivalent input capacitance C of the post-OPA a and resistor R a Also drawn outside the post-OPA; voltage source B p and P p and P n Either a battery or other regulated power supply with a certain accuracy can be used.

[0173] Here, the HEMT tube used in the differential circuit of the TIA circuit can be selected from various existing HEMTs in the world that can work below 4.2K and have stable transconductance and conductance at a bandwidth above 30MHz; for example, the two HEMTs recommended above, developed by CNRS / LPN in France: CNRS-HEMT and CNRS-PHEMT, have performance parameters shown in Table 2. In low-temperature STM, the tip-sample is generally in a low-temperature region below 4.2K. Since HEMT can work normally at temperatures below 4.2K (CNRS-HEMT can even work at temperatures below 0.5K), the HEMT in the preamplifier (as well as R1, R2 and R F ) is placed very close to the needle tip, and the signal feeder connecting the two can be replaced with a gold wire, so that the capacitance C of the feeder can be I Reduced to below 0.5pF.

[0174] In order to make a specific embodiment, the specific parameters of each part of the TIA circuit are given first: Table 3 is Figure 3 The parameters of the post-op amplifier circuit are different due to the different post-OPA models and feedback resistor R f are also different, so they are divided into four groups, which are listed in four rows according to the row numbers in the table. a0 and f b The meaning of is as described in (II) in the previous "Summary of the Invention"; Table 4 is Figure 3 The parameters of the differential circuit and constant current power supply of the dedicated preamplifier are listed in two rows according to the specifications of the HEMT; Table 5 is Figure 3 Parameters of the feedback circuit and signal input circuit.

[0175] Table 3 Parameters of post-op amp circuit

[0176]

[0177] Table 4 Parameters of dedicated preamplifier circuit

[0178]

[0179] Table 5 Parameters of feedback circuit and signal input circuit

[0180]

[0181] Table 6 is Figure 3 Circuit parameters of five embodiments of “TIA circuit including HEMT×2” are shown, and the parameters of each part of each embodiment are taken from the specified rows in Tables 3, 4, and 5.

[0182] Table 6 Circuit parameters of five embodiments of TIA circuits including HEMT×2

[0183]

[0184] Table 7 lists the poles s1 and s2 calculated by formula (f2-3) for the macro op amps in embodiments ① to ④, and the corresponding poles s1 and s2 according to formula a A =A vP A va and A vP and A va The circuit performance parameters are calculated using the expression (f1-1)(f2-2).

[0185] Table 7 Performance parameters of the macro op amps in Examples ① to ④ calculated by the formula (f1 = –s1 / (2π), f2 = –s2 / (2π))

[0186]

[0187]

[0188] Figures 4 to 7 Table 8 is the result of simulating the performance of the TIA circuit in the above embodiment by software. The simulation software used is TINA-TI of TI. The simulation uses the performance parameter calculation models of three operational amplifiers, THS4021, OPA657 and LMH6624, which have been embedded in TINA-TI software. These three operational amplifiers are also products of TI. The simulation also uses the original parameters of CNRS-HEMT and CNRS-PHEMT listed in Table 2.

[0189] Figure 4 is an explanatory diagram of the open-loop voltage gain of the macro op amp and the stability of the TIA circuit in Example ①; a curve obtained by simulating the macro op amp using TINA-TI software A | dB and ∠(a A ) is a curve showing the modulus and phase angle of the open-loop voltage gain of the macro op amp as a function of frequency f; from the curve, we can measure f -60 =664kHz, |a A (f -60 )| dB = 89dB. For f1 = –s1 / (2π) = 694kHz and f2 = –s2 / (2π) = 2460kHz calculated by formula (f2-3), |a A (f1)| dB =88.3dB, ∠(a A (f1))=-62°,|a A (f2)| dB =77.3dB, ∠(a A(f2)) = -121°. Comparing the data measured by the curve with the results calculated by the formula listed in Table 7, the two are basically consistent, which is a verification of the correctness of the formula used. Curve |1 / β| dB and ∠(1 / β) are obtained by taking the following parameters according to formula (f5-2): R F =1GΩ, R k =100kΩ, C k =0.2pF, C = C A +C I =26.5pF, R J =0.001R F From the figure, we can see that: 2πfCR is satisfied F =2a1f y =320kHz, which is less than f -60 =664kHz, the stability condition of the TIA circuit described in technical solution (II) is met, and the loop gain T L (f) = a A The stability amplitude margin and angle margin shown by (f) / (1 / β(f)) far exceed the usual basic requirements.

[0190] Figure 5 is the voltage gain A of the TIA circuit of Example ① v The results obtained by simulation using TINA-TI software are as follows: v It means "in R J =R F =1GΩ" TIA circuit has an output AC voltage and an input modulation signal voltage ratio. The solid line and virtual image show the analog|A v | dB and phase angle ∠(A v ) changes with the modulation frequency f. From the figure, we can see that at frequency f y =320kHz, |A v | dB and phase angle ∠(A v ) are –3.0dB and 101° respectively. It can also be seen that curve |A v | dB As f decreases monotonically, there is no upward convex hull, which also reflects that the circuit is quite stable.

[0191] Figure 6 The step response performance of the TIA circuit in Example ① is simulated by TINA-TI software. Here, R J =R F=1GΩ. The dotted lines in the figure below and the figure above represent the upward and downward step voltage signals input at the modulation signal input port, respectively, while the solid lines represent the voltage change curves of the TIA output terminal in response to the step voltage signal; the time it takes for the voltage change to reach 0.1% error in the step signal voltage response is recorded as t s , called the “step response adjustment time”, both the lower and upper figures show t s = 3.5μs. The output voltage curve does not have "ringing" and "overshoot" characteristics, which also reflects that the circuit is quite stable.

[0192] Figure 7 The diagram is an illustration of the open-loop voltage gain of the macro op amp and the stability of the TIA circuit in Example ④; the last row of Table 7 shows the open-loop voltage gain a of the macro op amp. A =A vP A va There are two conjugate imaginary poles s1 and s2, which is different from the macro op amps in other embodiments where both poles are negative real numbers. The curve obtained by simulating the macro op amp in embodiment ④ using TINA-TI software |a A | dB and ∠(a A )and Figure 4 The curve in |a A | dB and ∠(a A ) are compared, the shape is not much different, but it drops faster at the high frequency end. From the curve, we can measure f -60 =955kHz, |a A (f -60 )| dB =87dB. Curve |1 / β| dB and ∠(1 / β) are also obtained by taking the following parameters according to formula (f5-2): R F =1GΩ, R k =100kΩ, C k =0.2pF, C = C A +C I =26.5pF, R J =0.001R F From the figure we can see: y =255kHz, which is less than f -60 =955kHz, the stability condition of the TIA circuit described in technical solution (II) is met, and the loop gain T L (f) = a A The stability amplitude margin and angle margin shown by (f) / (1 / β(f)) far exceed the usual basic requirements.

[0193] Table 8 is a summary of the results obtained by simulating the performance of the macro operational amplifier and the entire TIA circuit in Examples ① to ⑤.

[0194] Table 8 summarizes the performance parameters of the macro op amp and the entire TIA circuit simulation in Examples ① to ⑤

[0195]

[0196] As can be seen from Table 8, the bandwidth of the macro op amp of the TIA circuit here is approximately 0.7~1.2MHz, the maximum operating frequency of the TIA circuit is approximately 0.25~0.35MHz, and the adjustment time of the step response is approximately 0.35~4μs; the feedback resistor R in the circuit here is F =1GΩ, so the transimpedance gain of the circuit is generally high.

[0197] (ii) Further describe the specific implementation methods of technical solutions (iv) and (v).

[0198] Figure 8 This is a circuit diagram of a dedicated preamplifier including a HEMT×4 differential circuit. Figure 8 In the figure, the dotted box 1.1-C is the HEMT×4 differential circuit in the dedicated preamplifier, and the dotted box 1.2-C is the constant current power supply circuit of the dedicated preamplifier. Figure 3 The constant current power supply in the dotted box 1.2 is the same, and the current output end of the constant current power supply is connected to the power supply current input end of the differential circuit to form a dedicated preamplifier. Figure 8 The dedicated preamplifier circuit shown replaces Figure 3 The dedicated preamplifier circuit in the circuit diagram becomes the circuit diagram of the "TIA circuit including HEMT×4" here, which is referred to as the "circuit Figure 3 -C".

[0199] The HEMT tube used in the HEMT×4 differential circuit here can adopt the same specifications as the HEMT tube in the previous HEMT×2 differential circuit. F ) can be placed in a low temperature area below 4.2K, so that the signal feeder connected to the needle tip is as short as possible and a gold wire is used instead, which can reduce the capacitance C of the feeder. I Reduced to below 0.5pF.

[0200] To make a specific embodiment, the specific parameters of each part of the TIA circuit are given first: Circuit Figure 3 -C has the same parameters as those in Table 3; Table 9 is the circuit Figure 3The parameters of the differential circuit and constant current power supply of the dedicated preamplifier in -C are listed in two rows according to the specifications of the HEMT. The parameters listed in row 1 are compared with the parameters listed in row 1 of the previous table 4. Here, the drain load resistors R1 and R2 connected to the drains of the two field effect tubes in the differential circuit are R1=R2=R L =0.5kΩ, which is R in Table 4 L = half of 1kΩ, this is to make the voltage gain A vP Still 20, here the equivalent input capacitance C of the preamplifier A =7.5pF, approximately C in Table 4 A = 1 / 3 of 26pF, where the output resistance R d =R L =500Ω, which is similar to R in Table 4 d Same; Table 10 is the circuit Figure 3 -C, the parameters of the feedback circuit and signal input circuit, compared with the parameters in Table 5, here the feedback resistor R F The resistor R used to compensate the bandwidth increases from 1GΩ to 3GΩ. k Still 100kΩ, take such a large R F This TIA circuit can be used to measure R in the range of 0.003 to 3GΩ J .

[0201] Table 9 Parameters of dedicated preamplifier circuit

[0202]

[0203] Table 10 Parameters of feedback circuit and signal input circuit

[0204]

[0205] Table 11 is the circuit Figure 3 -C shows circuit parameters of the five embodiments of the “TIA circuit including HEMT×4”, where the parameters of each part of each embodiment are taken from the specified rows in Tables 3, 9, and 10.

[0206] Table 11 Circuit parameters of five embodiments of TIA circuits including HEMT×4

[0207]

[0208] Figures 9 to 11 Table 12 is the result of simulating the TIA circuit performance of the above embodiment using TI's TINA-TI software, and the description of the situation is the same as the previous Figures 4 to 6 , similar to Table 8.

[0209] Fig. 9This is a diagram for explaining the open-loop voltage gain of the macro op amp and the stability of the TIA circuit in Example ⑥, where the curve |1 / β| dB and ∠(1 / β) are obtained by taking the following parameters according to formula (f5-2): R F =3GΩ, R k =100kΩ, C k =0.2pF, C = C A +C I =8pF, R J =0.001R F .

[0210] Fig.10 is the voltage gain A of the TIA circuit of Example ⑥ v The results obtained by simulation with TINA-TI software are shown in the figure below. J =R F =3GΩ.

[0211] Fig.11 The step response performance of the TIA circuit in Example ⑥ is simulated by TINA-TI software. Here, R J =R F =3GΩ.

[0212] Table 12 is a summary of the results obtained by simulating the performance of the macro operational amplifier and the entire TIA circuit in Examples ⑥ to ⑩.

[0213] Table 12 summarizes the performance parameters of the macro op amp and the entire TIA circuit simulation in Examples ⑥ to ⑩

[0214]

[0215] It can be seen from Table 12 that the bandwidth of the macro op amp of the TIA circuit here is approximately 0.65~1.2MHz, the maximum operating frequency of the TIA circuit is approximately 0.3~0.4MHz, and the adjustment time of the step response is approximately 2.5~3.5μs; the feedback resistor R in the circuit here is F =3GΩ, so the transimpedance gain of the circuit is very high.

[0216] (iii) Compare the noise conditions of each transimpedance amplifier circuit.

[0217] The equivalent input noise density of the following TIA circuits used for low-temperature STM is compared:

[0218] TIA circuit embodiment ①, which includes a CNRS-HEMT×2 differential circuit,

[0219] TIA circuit embodiment ⑥, which includes a CNRS-HEMT×4 differential circuit,

[0220] The TIA circuit in Reference 4 has a macro op amp composed of three cascaded op amps, the first stage of which is AD8615.

[0221] FEMTO's TIA circuit includes a TIA model DLPCA-200 produced by FEMTO.

[0222] Field effect tube CNRS-HEMT its equivalent input voltage noise density At f = 10kHz and f = 100kHz respectively and Its equivalent input current noise density At f = 10kHz and f = 100kHz respectively and (See Table 2). For the TIA circuits of embodiments ① and ⑥, (VI) and (VII) in the previous "Summary of the Invention" give the noise calculation formulas (f6-3), (f6-4) and (f7-2), and calculate the values ​​of each noise density at f = 10kHz. According to these formulas, the values ​​of each noise density at f = 100kHz can be calculated. The calculation results are shown in Table 13.

[0223] The TIA circuit in Reference 4 has a macro op amp. and And the equivalent input current noise density of the entire TIA circuit The query and calculation results have been given in (VI) and (VII) above, and are found to be basically consistent with the experimental results reported in Reference 4. These results are also listed in Table 13.

[0224] FEMTO's TIA circuit, where the equivalent input voltage noise density of the macro op amp and equivalent input current noise density It can be deduced from the data in the DLPCA-200 product manual: at f = 1kHz, Then, according to formula (f7-2), the equivalent input current noise density of the entire TIA circuit can be calculated: The calculation results are shown in Table 13.

[0225] By the way, the calculation formulas for various noise densities of Examples ⑤ and ⑩ of CNRS-PHEM are the same as those of Examples ① and ⑥, and the calculation results are placed in square brackets in the grid of Table 13.

[0226] Table 13 Equivalent input noise of different transimpedance amplifier circuits

[0227]

[0228] It should also be pointed out that the amplifier circuit reported in Reference 5 for low-temperature STM measurements is capable of measuring tunnel junction shot noise signals. Their measurements were performed at the resonant frequency of their LC circuit, 1 MHz. At this frequency, the equivalent input current noise density of the amplifier circuit is Approx. In terms of the large inductance and high Q value of the superconducting coil in its LC circuit, it is difficult to reduce the resonant frequency, and if there is a strong magnetic field in the STM, the performance of the superconducting coil will be greatly affected. In fact, under normal circumstances, measuring the tunnel junction shot noise at 20kHz is already far away from the influence of 1 / f noise, so the TIA circuit of the present invention can measure the tunnel junction shot noise quite well, not only the operating frequency is suitable, but also the influence of the circuit inherent noise is very low. For example, the TIA circuit of the embodiments ① and ⑥ has an equivalent input current noise density of 100kHz. They are and Much lower than the amplifier circuit in reference 5

[0229] In summary, it can be seen from the above that the use of the TIA circuit of the present invention in a low-temperature STM can greatly increase the operating frequency and reduce circuit noise, so that the STM can complete scanning and spectrum measurement at high speed and high precision, and can also enable the STM to measure weak signals such as tunnel junction shot noise quite well, which provides conditions for using low-temperature STM to discover new phenomena and new laws of microscopic matter movement.

[0230] The aforementioned technical solutions (I) to (V) are the basic contents of the present invention. The aforementioned (I) to (VIII) and the specific implementation methods here further elaborate on this basic content and provide examples, but this is only a more detailed description of the technical solution, rather than a limitation on the protection scope of the present invention. Therefore, certain equivalent replacements, similar improvements, etc. made to the basic features of the technical solutions (I) to (V) of the present invention still fall within the protection scope of the present invention.

Claims

1. A transimpedance amplifier circuit for measuring tunnel junction current related signals in a low temperature STM, characterized in that: (1) The circuit consists of four parts, namely, the "dedicated preamplifier" part, the "post-amplifier circuit" part, the "feedback circuit" part, and the "signal input circuit" part; (1.1) The "dedicated preamplifier" part, referred to as "dedicated preamplifier", consists of two parts, namely, "differential circuit" and "constant current power supply" for supplying power to the differential circuit; The structure of the "differential circuit" of the dedicated preamplifier is as follows: a pair of "high electron mobility field effect transistors" H1 and H2 with the same specifications are connected as follows: the two sources of the field effect transistors H1 and H2 are connected together, and then a resistor R g The other end of the resistor is grounded, and a capacitor C of more than 5μF is connected in parallel to the resistor. g The two drains of field effect transistors H1 and H2 are connected to drain load resistors R1 and R2 respectively, where R1 = R2, and this resistance value is recorded as R L , the other ends of R1 and R2 are denoted as L1 and L2 respectively, L1 and L2 are connected to capacitors C1 and C2 respectively, and the other ends of C1 and C2 are grounded, where C1=C2=C u , C u is 5μF or more; L1 and L2 are connected to a potentiometer R 12 The inactive ends at both ends of the potentiometer are used as the "power supply current input end" of the differential circuit. 12 The resistance between the two ends is R L / 10 or less; the two drains of the two field effect tubes H1 and H2 are respectively called the "positive phase output terminal" and "negative phase output terminal" of the dedicated preamplifier, and the two gates of H1 and H2 are respectively called the "negative phase input terminal" and "positive phase input terminal" of the dedicated preamplifier, and the positive phase input terminal is grounded; The "constant current power supply" of the dedicated preamplifier includes a "current stabilization network" composed of transistors and resistors and a "positive voltage source" for supplying power to the current stabilization network. The negative electrode of the positive voltage source is grounded, and the positive electrode is connected to the current stabilization network. The "current output end" of the current stabilization network is connected to the power supply current input end of the differential circuit. The constant current is input to the differential circuit through the regulation of the current stabilization network. The differential circuit is only powered by this constant current power supply. The dedicated preamplifier powered by the constant current power supply has an equivalent input capacitance denoted as C A , its equivalent input resistance is denoted as R A ; (1.2) The "post-op amp circuit" is composed of the following: a ready-made operational amplifier, called "post-OPA", whose equivalent input capacitance and equivalent input resistance are denoted by C a and R a , connect a feedback resistor R from the output of the post OPA f To its inverting input, two cables are connected to the two inputs of the post-OPA respectively; the cable connected to the non-inverting input of the post-OPA has a capacitance to ground denoted as C i1 The end of the cable is called the "positive input of the post-op amplifier circuit", which is connected to the positive output of the dedicated preamplifier; the cable connected to the inverting input of the post-OPA has a capacitance to ground denoted as C i2 The end of the cable is called the "inverting input of the post-amplifier circuit", which is connected to the inverting output of the dedicated preamplifier; here C i1 =C i2 ; The output end of the post OPA is the output end of the post op amp circuit; The circuit formed by connecting the "dedicated preamplifier" and the "post-op amp circuit" is called a "macro operational amplifier", abbreviated as "macro op amp"; the inverting input terminal and the non-inverting input terminal of the dedicated preamplifier are respectively called the inverting input terminal and the non-inverting input terminal of this macro op amp, and the output terminal of the post-op amp circuit is also called the output terminal of this macro op amp; the voltage of the non-inverting input terminal of this macro op amp relative to the inverting input terminal is called the "input voltage of the macro op amp", and the voltage of the output terminal of this macro op amp relative to the ground is called the "output voltage of the macro op amp", and the ratio of the latter to the former is called the "open-loop voltage gain of the macro op amp", denoted as a A or a A (f); (1.3) The "feedback circuit" is composed of: a feedback resistor R F A "bandwidth compensation network" consisting of a resistor and a capacitor is connected, and the other end of the bandwidth compensation network is connected to the output end of the macro op amp, and the feedback resistor R F The other end of the feedback circuit is connected to the inverting input of the macro op amp; the impedance of the feedback circuit as a function of the signal frequency f is recorded as Z F =Z F (f); The circuit formed by connecting the macro op amp and the feedback circuit is called a "transimpedance amplifier", or "TIA" for short; (1.4) The "signal input circuit" is composed as follows: there is a tunnel junction between the tip of the STM and the sample, the "bias-modulation signal voltage source" is abbreviated as BMS, one end of the BMS is grounded, and the other end is connected to the sample end of the tunnel junction. The tip of the tunnel junction is connected to the "signal feeder", and the other end of the signal feeder serves as the signal output end of the "signal input circuit", which is connected to the inverting input end of the macro op amp in the TIA; the capacitance of the signal feeder to ground is denoted as C I ; The circuit formed by connecting the signal input circuit and the TIA is called "TIA with input", also referred to as "TIA circuit" or "transimpedance amplifier circuit"; (2) The tip-sample in the signal input circuit is in the low temperature zone set for it by the low temperature STM, all the field effect transistors in the differential circuit and the two drain load resistors R1 and R2, as well as the feedback resistor R in the feedback circuit F , are placed in the low temperature zone where the needle tip-sample is located.

2. The transimpedance amplifier circuit according to claim 1, characterized in that: The structure-performance parameters of the macro operational amplifier, feedback circuit and signal input circuit meet the following conditions: (1) For the open-loop voltage gain a of the macro op amp A =a A (f), whose modulus is denoted by |a A (f)|, and its phase angle is denoted as ∠(a A (f)), there is a frequency f in the frequency domain [5kHz, +∞) -60 Make ∠(a A (f -60 ))=-60°,at (0,f -60 ] has ∠(a A (f))≥-60°, and in [f -60 ,+∞) has |a A (f)|≤|a A (f -60 )|; (2) The bandwidth compensation of the feedback circuit makes the frequency range [100Hz, f -60 ]Inside||Z F (f) / R F | dB |≤1dB; (3) Let a1 = |a A (f -60 )|, let C=C I +C A , let f y =2a1 / (2πR F C), parameter f y So that the following inequality holds: 1kHz≤f y ≤f -60 ; (4) The frequency f of the modulation signal used by STM for spectrum measurement w ∈[100Hz,f y ]; (5)R A >10R F 。 3. The transimpedance amplifier circuit according to claim 1 or 2, characterized in that: The structure of the "constant current power supply" of the dedicated preamplifier is as follows: two silicon-based PNP bipolar transistors T1 and T2 are used to form a "current stabilization network" in the following manner: a resistor R is connected between the emitter and base of transistor T1. b , and the collector of T1 is connected through a resistor R t The emitter of transistor T2 is connected to the base of T1, and the base of T2 is connected to the collector of T1; the collector of transistor T2 is connected to the power supply current input of the differential circuit as the current output terminal of the constant current power supply; the "positive voltage source" that supplies power to the current stabilization network has its negative electrode grounded and its positive electrode connected to the emitter of transistor T1.

4. A transimpedance amplifier circuit for measuring tunnel junction current related signals in a low temperature STM, characterized in that: It is a circuit formed by partially changing the transimpedance amplifier circuit described in claim 1 or 2, that is, the "differential circuit" is changed while other parts remain unchanged, and the "differential circuit" is changed into the following structure: four "high electron mobility field effect transistors" H11, H21, H12, and H22 with the same specifications are used, and connected as follows: the drain of the field effect transistor H11 is connected to the source of the field effect transistor H12, the drain of the field effect transistor H21 is connected to the source of the field effect transistor H22, and the gates of the two field effect transistors H12 and H22 are respectively connected to the sources of the two field effect transistors H11 and H21; the two sources of the field effect transistors H11 and H21 are connected together, and then connected to a resistor R g The other end of the resistor is grounded, and a capacitor C of more than 5μF is connected in parallel to the resistor. g The two drains of the two field effect transistors H12 and H22 are connected to the drain load resistors R1 and R2 respectively, where R1 = R2, and this resistance value is recorded as R L , the other ends of R1 and R2 are denoted as L1 and L2 respectively, L1 and L2 are connected to capacitors C1 and C2 respectively, and the other ends of C1 and C2 are grounded, where C1=C2=C u , C u is 5μF or more; L1 and L2 are connected to a potentiometer R 12 The inactive ends at both ends, and the active end in the middle of the potentiometer is used as the "power supply current input end" of the differential circuit, connected to the current output end of the constant current power supply. 12 The resistance between the two ends is R L / 10 or less; the two drains of the two field effect transistors H12 and H22 are respectively called the "positive phase output terminal" and "negative phase output terminal" of the dedicated preamplifier, and the two gates of the two field effect transistors H11 and H21 are respectively called the "negative phase input terminal" and "positive phase input terminal" of the dedicated preamplifier, and the positive phase input terminal is grounded; the preamplifier formed by connecting the differential circuit with the constant current power supply has an equivalent input capacitance denoted as C A , its equivalent input resistance is denoted as R A .

5. The transimpedance amplifier circuit according to claim 4, characterized in that: The "constant current power supply" described therein has the following structure: two silicon-based PNP bipolar transistors T1 and T2 are used to form a "current stabilization network" in the following manner: a resistor R is connected between the emitter and base of transistor T1. b , and the collector of T1 is connected through a resistor R t The emitter of transistor T2 is connected to the base of T1, and the base of T2 is connected to the collector of T1; the collector of transistor T2 is connected to the power supply current input of the differential circuit as the current output terminal of the constant current power supply; the "positive voltage source" that supplies power to the current stabilization network has its negative electrode grounded and its positive electrode connected to the emitter of transistor T1.

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