Signal processing method and device

Abstract

The present application discloses a signal processing method and device, which belongs to the field of radar technology and can be used in scenarios such as intelligent transportation (such as smart transportation, assisted driving or autonomous driving), smart home, and robots. The method includes: obtaining a first range-direction time domain signal of a first subband and a second range-direction time domain signal of a second subband adjacent to the first subband; synthesizing and superimposing the first range-direction time domain signal and the second range-direction time domain signal to obtain a third range-direction time domain signal; obtaining a first peak point and a second peak point of the third range-direction time domain signal; and determining a constant phase error θ between the first range-direction time domain signal and the second range-direction time domain signal based on the first peak point and the second peak point. err This method can achieve high-resolution radar imaging in the range direction and can be applied to vehicle-to-vehicle (V2V) systems, such as vehicle-to-everything (V2X), long-term evolution (LTE-V) and vehicle-to-vehicle (V2V).

Description

Technical Field

[0001] The present application relates to the field of radar technology, and more specifically, to a signal processing method and device. Background Art

[0002] With the development of society, smart transportation, smart homes, and smart robots are gradually becoming part of people's daily lives. Sensors play a crucial role in various smart electronic devices. Various radar sensors installed in these devices, such as millimeter-wave radar, lidar, and ultrasonic radar, can detect and identify targets, sense the surrounding environment, identify and track moving objects, and recognize static scenes. Sensors enhance the environmental perception capabilities of smart electronic devices, enabling intelligent transportation, smart homes, and smart robots.

[0003] In radar systems, high-resolution radar imaging can be used to achieve high-resolution target detection and recognition, such as in high-resolution synthetic aperture radar (SAR). High resolution in the radar's range dimension can be achieved by transmitting ultra-wideband linear frequency modulation (LCM) signals. Typically, a high resolution of 0.05 meters in the range dimension corresponds to a radar transmission bandwidth exceeding 3 GHz. In direct acquisition mode, the radar echo frequency response characteristics obtained from ultra-wideband signals are not ideal, introducing significant amplitude and phase errors, making it difficult to achieve ideal pulse compression results. Therefore, transmitting ultra-wideband signals above 3 GHz requires ensuring high requirements for the radar system environment and high linearity for wide-bandwidth signals. Furthermore, directly receiving such wide-bandwidth signals requires an ultra-high-speed analog-to-digital (A / D) converter and memory, which places significant pressure on the receiver's in-phase and quadrature (I / Q) detection, increasing system complexity and cost.

[0004] Considering the actual system complexity and cost, ultra-wideband signals can be synthesized by frequency-stepping multi-subband signals to achieve high-resolution imaging in the range direction. In a multi-subband radar system, phase mismatches between sub-band signals will lead to a deterioration of the range impulse response, affecting the effectiveness of sub-band coherent synthesis. Multi-subband bandwidth distribution includes three modes: sub-band overlap mode, adjacent mode, and sub-band spacing mode. Currently, interferometric phase extraction can be performed on the common portion of the overlapping sub-bands in the sub-band overlap mode to estimate the phase error between the sub-band signals, and compensation is performed based on this phase error to achieve sub-band splicing. However, this scheme cannot achieve sub-band splicing in the sub-band adjacent mode or sub-band spacing mode. Therefore, a bandwidth synthesis scheme suitable for the three modes of sub-band overlap, adjacent mode, and sub-band spacing is urgently needed. Summary of the Invention

[0005] The present application provides a signal processing method and apparatus, which can realize sub-band splicing in a sub-band overlapping mode, a sub-band adjacent mode, and a sub-band interval mode.

[0006] In a first aspect, a signal processing method is provided, comprising:

[0007] Acquire a first range-direction time domain signal of a first sub-band and a second range-direction time domain signal of a second sub-band adjacent to the first sub-band;

[0008] synthesizing and superimposing the first range-direction time domain signal and the second range-direction time domain signal to obtain a third range-direction time domain signal;

[0009] Acquire a first peak point and a second peak point of the third range time domain signal;

[0010] Determine the constant phase error θ between the first range time domain signal and the second range time domain signal based on the first peak point and the second peak point. err ,θ err ∈[0,2π].

[0011] Therefore, the embodiment of the present application obtains a spliced ​​bandwidth composite signal by synthesizing and superimposing the distance time domain signals of adjacent sub-bands, and obtains two peak points in the spliced ​​bandwidth composite signal, such as a first peak point and a second peak point. Since the two peak points in the bandwidth composite signal are related to the constant phase between the two adjacent sub-bands before splicing, the embodiment of the present application can obtain the constant phase error between the adjacent sub-bands based on the two peak points. Since the embodiment of the present application does not involve the overlapping sub-band common spectrum portion of adjacent sub-bands in the process of determining the constant phase error, the embodiment of the present application can improve spectrum utilization and can be applied to three modes: sub-band overlapping mode, sub-band adjacent mode, and sub-band spacing mode.

[0012] In conjunction with the first aspect, in certain implementations of the first aspect, the first peak point is a peak point corresponding to a main lobe of the third range-direction time-domain signal, and the second peak point is a peak point corresponding to a first sidelobe adjacent to the main peak of the third range-direction time-domain signal, wherein the peak point corresponding to the first sidelobe is higher than the peak point corresponding to the second sidelobe adjacent to the main peak of the third range-direction time-domain signal. In other words, the second peak point is a peak point corresponding to the second highest peak in the third range-direction time-domain signal.

[0013] Wherein, the constant phase error θ between the first range-direction time domain signal and the second range-direction time domain signal is determined based on the first peak point and the second peak point. err ,include:

[0014] According to the difference between the first peak point and the second peak point, the residual constant phase error Δθ of the third range-direction time domain signal is determined, wherein the third range-direction time domain signal is a constant phase error θ err Compensation of the first compensation value θ is performed, Δθ=θ-θ err , Δθ∈[0,2π], θ∈[0,2π];

[0015] Determine the constant phase error θ according to the residual constant phase error Δθ and the first compensation value θ err .

[0016] Among them, the residual constant phase error Δθ has a mapping relationship with the difference between the first peak point and the second peak point. err There is also a mapping relationship between them, so the constant phase error θ err The difference between the first peak point and the second peak point also has a mapping relationship. Here, the constant phase error θ can be err This mapping relationship between the difference between the first peak point and the second peak point is called a main lobe splitting operation model.

[0017] Therefore, the embodiment of the present application obtains the difference between the main lobe and the first side lobe (i.e., the side lobe corresponding to the second peak) in the bandwidth synthesis signal, and according to the mapping relationship between the difference between the main lobe and the first side lobe in the bandwidth synthesis signal and the residual constant phase error Δθ, and the residual constant phase error Δθ and the constant phase error θ err The mapping relationship is based on the main lobe splitting inverse operation model to obtain the constant phase error θ between sub-bands. err .

[0018] In combination with the first aspect, in certain implementations of the first aspect, when the peak point corresponding to the first side lobe is to the left of the peak point corresponding to the main lobe, the value range of the residual constant phase error Δθ is [0,π], that is, at this time the first side lobe is the side lobe adjacent to the left of the main lobe.

[0019] When the peak point corresponding to the first side lobe is on the right side of the peak point corresponding to the main lobe, the value range of the residual constant phase error Δθ is [π, 2π]. That is, at this time, the first side lobe is the side lobe adjacent to the right side of the main lobe.

[0020] In this way, the value range of the residual constant phase error Δθ can be further obtained based on the position of the first side lobe relative to the main lobe, that is, whether the first side lobe is the left adjacent side lobe or the right adjacent side lobe, which can help to more accurately determine the corresponding residual constant phase error Δθ based on the difference between the peak point of the main lobe and the peak point of the first side lobe.

[0021] In combination with the first aspect, in some implementations of the first aspect, determining the residual constant phase error Δθ of the third range-direction time-domain signal based on the difference between the first peak point and the second peak point includes:

[0022] When the difference between the peak point corresponding to the main lobe of the third range-direction time-domain signal and the peak point corresponding to the first side lobe is a minimum value, the residual constant phase error Δθ is determined to be π. Here, the minimum value of the difference includes the difference being 0 and the difference being approximately 0, which is not limited in this embodiment of the present application.

[0023] Therefore, the embodiment of the present application obtains the minimum value of the difference between the main lobe and the first side lobe in the bandwidth synthesis signal (i.e., 0 or approximately 0), and when the difference between the main lobe and the first side lobe in the bandwidth synthesis signal is the minimum value, the remaining constant phase error Δθ is π, at this time θ err =θ-π, and the compensation value θ corresponding to the bandwidth synthesis signal is substituted into it to obtain the constant phase error θ between sub-bands. err .

[0024] In conjunction with the first aspect, in certain implementations of the first aspect, the first range time domain signal is expressed as the following formula:

[0025]

[0026] The second range time domain signal is expressed as follows:

[0027]

[0028] The third range time domain signal is expressed as follows:

[0029]

[0030] Among them, r1(t q )=sin c(2γTt q ), r2(t q )=sin c(γTt q )sin(πγTt q ), R d (t q ; θ) represents the third distance to the time domain signal, R i (t q ) indicates that the i-th subband is at distance t q The distance to the time domain signal at the moment, where i∈[1,I], I represents the number of sub-bands that need to be synthesized, q∈[1,Q] represents the time of distance discrete sampling, Q represents the total time of distance discrete sampling, sin c(γTt q) represents the signal range envelope signal, γ represents the range linear frequency modulation slope, T represents the radar transmission time period, Indicates the signal distance phase information, Represents the phase error information of the inter-subband signal distance.

[0031] In combination with the first aspect, in some implementations of the first aspect, the first peak point is a peak point corresponding to a sidelobe adjacent to the left side of a main lobe of the third range-direction time domain signal, and the second peak point is a peak point corresponding to a sidelobe adjacent to the right side of a main lobe of the third range-direction time domain signal;

[0032] Wherein, the constant phase error θ between the first range-direction time domain signal and the second range-direction time domain signal is determined based on the first peak point and the second peak point. err ,include:

[0033] According to the difference between the first peak point and the second peak point, and the difference and the constant phase error θ err The first mapping relationship determines the constant phase error θ err .

[0034] Here, the first mapping relationship between the constant phase error and the difference between the first peak point and the second peak point may be referred to as a left-right sidelobe equalization model.

[0035] Therefore, the embodiment of the present application obtains the difference between the left adjacent side lobe and the right adjacent side lobe of the main lobe in the bandwidth synthesis signal, and calculates the difference between the left adjacent side lobe and the right adjacent side lobe of the main lobe in the bandwidth synthesis signal and the constant phase error θ between adjacent sub-bands. err The mapping relationship is that based on the left and right sidelobe equalization models, the constant phase error θ between subbands is obtained. err .

[0036] In combination with the first aspect, in some implementations of the first aspect, the constant phase error θ between the first range-direction time domain signal and the second range-direction time domain signal is determined based on the first peak point and the second peak point. err Previously, it also included:

[0037] Get the phase error θ between the adjacent side lobe on the left side of the main lobe and the constant err The second mapping relationship;

[0038] Get the phase error θ between the adjacent side lobe on the right side of the main lobe and the constant err The third mapping relationship;

[0039] The first mapping relationship is determined according to the first mapping relationship and the second mapping relationship.

[0040] Therefore, after obtaining the third range-direction time domain signal, the embodiment of the present application can respectively obtain the left adjacent side lobe of the main lobe of the third range-direction time domain signal and the constant phase error θ err The second mapping relationship, and the adjacent side lobe on the right side of the main lobe and the constant phase error θ err The third mapping relationship is determined, and then the first mapping relationship is determined according to the second mapping relationship and the third mapping relationship.

[0041] In conjunction with the first aspect, in some implementations of the first aspect, the first mapping relationship is shown in the following formula:

[0042]

[0043] Among them, P(θ err ) represents the difference between the first peak point and the second peak point.

[0044] In conjunction with the first aspect, in certain implementations of the first aspect, the first range time domain signal is expressed as the following formula:

[0045]

[0046] The second range time domain signal is expressed as follows:

[0047]

[0048] The third range time domain signal is expressed as follows:

[0049]

[0050] The side lobe Q adjacent to the left side of the main lobe of the third range time domain signal l (θ err ) satisfies the following formula:

[0051]

[0052] The side lobe Q adjacent to the right side of the main lobe of the third range time domain signal r (θ err ) satisfies the following formula:

[0053]

[0054] Among them, Q(t q ) represents the third range time domain signal, R i (t q ) indicates that the i-th subband is at distance t qThe distance to the time domain signal at the moment, where i∈[1,I], I represents the number of sub-bands that need to be synthesized, q∈[1,Q] represents the time of distance discrete sampling, Q represents the total time of distance discrete sampling, sin c(γTt q ) represents the signal range envelope signal, γ represents the range linear frequency modulation slope, T represents the radar transmission time period, Indicates the signal distance phase information, Represents the phase error information of the inter-subband signal distance.

[0055] In conjunction with the first aspect, in some implementations of the first aspect, the method further includes:

[0056] According to the constant phase error θ err , determine the constant phase error compensation function;

[0057] Compensating the first range-direction time-domain signal or the second range-direction time-domain signal according to the constant phase error compensation function;

[0058] The compensated first range time domain signal and the second range time domain signal are synthesized and superimposed to obtain a fourth range time domain signal.

[0059] Therefore, after obtaining constant phase error compensation between adjacent subbands, the embodiments of the present application can compensate for the adjacent subbands based on the constant phase error and perform bandwidth synthesis on the compensated adjacent subbands to obtain a high-resolution radar image in the range direction. Because the common spectrum of the overlapping subbands of adjacent subbands is not involved in the constant phase error determination process, the bandwidth synthesis scheme of the embodiments of the present application can improve spectrum utilization and is applicable to three modes: overlapping subband mode, adjacent subband mode, and spaced subband mode.

[0060] In conjunction with the first aspect, in certain implementations of the first aspect, before synthesizing and superimposing the first range-direction time domain signal and the second range-direction time domain signal to obtain a third range-direction time domain signal, the method further includes:

[0061] performing channel amplitude calibration on the first range-direction time domain signal and the second range-direction time domain signal respectively;

[0062] Rearranging the first range-direction time domain signal and the second range-direction time domain signal in sequence according to the order of carrier frequencies;

[0063] respectively compensating for high-order phase errors within sub-bands of the first range-direction time-domain signal and the second range-direction time-domain signal;

[0064] A first-order phase error between the first range-direction time-domain signal and the second range-direction time-domain signal is compensated.

[0065] Since the example of the present application calibrates the amplitude and phase characteristics within the adjacent sub-band, rearranges the signals of the adjacent sub-band in sequence according to the carrier frequency points, and compensates for the high-order phase error within the adjacent sub-band and the first-order phase error between adjacent sub-bands, before obtaining the constant phase error between adjacent sub-bands, the error between adjacent sub-bands is only the constant phase error. Therefore, the embodiment of the present application can accurately obtain the constant phase error between adjacent sub-bands based on the mapping relationship between the correlation peak point of the synthesized bandwidth synthesis result and the remaining constant phase error.

[0066] In a second aspect, a signal processing apparatus is provided, comprising:

[0067] An acquiring unit, configured to acquire a first range-direction time domain signal of a first sub-band and a second range-direction time domain signal of a second sub-band adjacent to the first sub-band;

[0068] a synthesis unit, configured to synthesize and superimpose the first range-direction time domain signal and the second range-direction time domain signal to obtain a third range-direction time domain signal;

[0069] The acquisition unit is further configured to acquire a first peak point and a second peak point of the third range time domain signal;

[0070] A determining unit, configured to determine a constant phase error θ between the first range-direction time domain signal and the second range-direction time domain signal according to the first peak point and the second peak point. err ,θ err ∈[0,2π].

[0071] In combination with the second aspect, in certain implementations of the second aspect, the first peak point is a peak point corresponding to a main lobe of the third range-direction time domain signal, and the second peak point is a peak point corresponding to a first side lobe adjacent to the main peak of the third range-direction time domain signal, wherein the peak point corresponding to the first side lobe is higher than the peak point corresponding to the second side lobe adjacent to the main peak of the third range-direction time domain signal;

[0072] The determining unit is specifically used for:

[0073] According to the difference between the first peak point and the second peak point, the residual constant phase error Δθ of the third range-direction time domain signal is determined, wherein the third range-direction time domain signal is a constant phase error θ err Compensation of the first compensation value θ is performed, Δθ=θ-θ err , Δθ∈[0,2π], θ∈[0,2π];

[0074] Determine the constant phase error θ according to the residual constant phase error Δθ and the first compensation value θ err .

[0075] In combination with the second aspect, in some implementations of the second aspect, when the peak point corresponding to the first side lobe is to the left of the peak point corresponding to the main lobe, the value range of the residual constant phase error Δθ is [0, π];

[0076] When the peak point corresponding to the first side lobe is on the right side of the peak point corresponding to the main lobe, the value range of the residual constant phase error Δθ is [π, 2π].

[0077] In conjunction with the second aspect, in some implementations of the second aspect, the determining unit is specifically configured to:

[0078] When the difference between the peak point corresponding to the main lobe of the third range time domain signal and the peak point corresponding to the first side lobe is a minimum value, the residual constant phase error Δθ is determined to be π.

[0079] In conjunction with the second aspect, in certain implementations of the second aspect, the first range time domain signal is expressed as the following formula:

[0080]

[0081] The second range time domain signal is expressed as follows:

[0082]

[0083] The third range time domain signal is expressed as follows:

[0084]

[0085] Among them, r1(t q )=sin c(2γTt q ), r2(t q )=sin c(γTt q )sin(πγTt q ), R d (t q ; θ) represents the third distance to the time domain signal, R i (t q ) indicates that the i-th subband is at distance t q The distance to the time domain signal at the moment, where i∈[1,I], I represents the number of sub-bands that need to be synthesized, q∈[1,Q] represents the time of distance discrete sampling, Q represents the total time of distance discrete sampling, sin c(γTt q) represents the signal range envelope signal, γ represents the range linear frequency modulation slope, T represents the radar transmission time period, Indicates the signal distance phase information, Represents the phase error information of the inter-subband signal distance.

[0086] In combination with the second aspect, in some implementations of the second aspect, the first peak point is a peak point corresponding to a sidelobe adjacent to the left side of a main lobe of the third range-direction time domain signal, and the second peak point is a peak point corresponding to a sidelobe adjacent to the right side of a main lobe of the third range-direction time domain signal;

[0087] The determining unit is specifically configured to:

[0088] According to the difference between the first peak point and the second peak point, and the difference and the constant phase error θ err The first mapping relationship determines the constant phase error θ err .

[0089] In conjunction with the second aspect, in some implementations of the second aspect, the acquiring unit is further configured to:

[0090] Get the phase error θ between the adjacent side lobe on the left side of the main lobe and the constant err The second mapping relationship;

[0091] Get the phase error θ between the adjacent side lobe on the right side of the main lobe and the constant err The third mapping relationship;

[0092] The first mapping relationship is determined according to the first mapping relationship and the second mapping relationship.

[0093] In conjunction with the second aspect, in some implementations of the second aspect, the first mapping relationship is shown in the following formula:

[0094]

[0095] Among them, P(θ err ) represents the difference between the first peak point and the second peak point.

[0096] In conjunction with the second aspect, in certain implementations of the second aspect, the first range time domain signal is expressed as the following formula:

[0097]

[0098] The second range time domain signal is expressed as follows:

[0099]

[0100] The third range time domain signal is expressed as follows:

[0101]

[0102] The side lobe Q adjacent to the left side of the main lobe of the third range time domain signal l (θ err ) satisfies the following formula:

[0103]

[0104] The side lobe Q adjacent to the right side of the main lobe of the third range time domain signal r (θ err ) satisfies the following formula:

[0105]

[0106] Among them, Q(t q ) represents the third range time domain signal, R i (t q ) indicates that the i-th subband is at distance t q The distance to the time domain signal at the moment, where i∈[1,I], I represents the number of sub-bands that need to be synthesized, q∈[1,Q] represents the time of distance discrete sampling, Q represents the total time of distance discrete sampling, sin c(γTt q ) represents the signal range envelope signal, γ represents the range linear frequency modulation slope, T represents the radar transmission time period, Indicates the signal distance phase information, Represents the phase error information of the inter-subband signal distance.

[0107] In combination with the second aspect, in some implementations of the second aspect, the determining unit is further configured to determine the phase error θ according to the constant phase error θ. err , determine the constant phase error compensation function;

[0108] a compensation unit, configured to compensate the first range time domain signal or the second range time domain signal according to the constant phase error compensation function;

[0109] The synthesis unit is further configured to synthesize and superimpose the compensated first range time domain signal and the second range time domain signal to obtain a fourth range time domain signal.

[0110] In conjunction with the second aspect, in some implementations of the second aspect, the method further includes:

[0111] a channel amplitude calibration unit, configured to perform channel amplitude calibration on the first range-direction time domain signal and the second range-direction time domain signal respectively;

[0112] a spectrum shifting unit, configured to rearrange the first range-direction time domain signal and the second range-direction time domain signal in sequence according to the order of carrier frequency points;

[0113] a high-order phase error compensation unit, configured to compensate for high-order phase errors in sub-bands of the first range-direction time domain signal and the second range-direction time domain signal respectively;

[0114] The first-order phase error compensation unit is used to compensate for the first-order phase error between the first range-direction time domain signal and the second range-direction time domain signal.

[0115] In a third aspect, a signal processing apparatus is provided, comprising: a processor, wherein the processor is configured to execute instructions stored in a memory, and when the processor executes the instructions stored in the memory, the execution causes the signal processing apparatus to perform the method of the first aspect or any possible implementation of the first aspect.

[0116] Optionally, the signal processing device also includes the above-mentioned memory.

[0117] According to a fourth aspect, a computer-readable medium is provided for storing a computer program, wherein the computer program comprises instructions for executing the method according to the first aspect or any possible implementation of the first aspect.

[0118] In a fifth aspect, a computer program product comprising instructions is provided, which, when run on a computer, enables the computer to execute the first aspect or any possible implementation of the first aspect.

[0119] In the sixth aspect, a chip is provided, comprising a processor and a communication interface, wherein the processor is used to call and run instructions from the communication interface, and when the processor executes the instructions, the method in the above-mentioned first aspect or any possible implementation method in the first aspect is implemented.

[0120] It should be understood that the beneficial effects achieved by the second to sixth aspects of the present application and the corresponding implementation methods can be referred to the beneficial effects achieved by the first aspect of the present application and the corresponding implementation methods, and will not be repeated here. BRIEF DESCRIPTION OF THE DRAWINGS

[0121] Figure 1 is a schematic block diagram of a radar device provided in an embodiment of the present application;

[0122] Figure 2 It is the comparison result of the range profile of a single sub-band signal before bandwidth synthesis of multi-sub-band signals and the broadband signal after synthesis;

[0123] Figure 3 is a schematic block diagram of another radar device provided in an embodiment of the present application;

[0124] Figure 4 is a schematic flow chart of a method for obtaining a radar image provided in an embodiment of the present application;

[0125] Figure 5 These are three schematic diagrams of bandwidth distribution of multi-subband echo signals;

[0126] Figure 6 This is a comparison diagram of the amplitude and phase characteristics within the sub-band before and after channel amplitude calibration;

[0127] Figure 7 There are three schematic diagrams showing that the multi-subband echo signals are arranged in sequence according to the carrier frequency points;

[0128] Figure 8 In the embodiment of the present application, r1(t q ) and r2(t q ) is a schematic diagram of the range impulse response function;

[0129] Figure 9 is the result of the synthesized bandwidth pulse pressure of the embodiment of the present application;

[0130] Figure 10 It is a mapping diagram of the residual constant phase error Δθ and the difference between the main lobe and the first side lobe of the synthesized bandwidth pulse compression result;

[0131] Figure 11 This is a comparison diagram of the distance pulse pressure envelope before and after bandwidth synthesis of the two sub-bands;

[0132] Figure 12 is a schematic flow chart of a signal processing method provided in an embodiment of the present application;

[0133] Figure 13 is a schematic block diagram of a signal processing device provided in an embodiment of the present application;

[0134] Figure 14 This is a schematic block diagram of another signal processing device provided in an embodiment of the present application. DETAILED DESCRIPTION

[0135] The technical solution in this application will be described below with reference to the accompanying drawings.

[0136] Figure 1The following is a schematic block diagram of a radar device 100 used in the signal processing method provided in an embodiment of the present application. By way of example, the radar device 100 may be a millimeter-wave radar, a lidar, an ultrasonic radar, a SAR radar, or the like, though this embodiment of the present application does not limit this. The radar device 100 can be applied to intelligent scenarios such as smart transportation, intelligent foreign object detection at airports, airborne and spaceborne radar imaging and mapping, smart homes, and smart robots.

[0137] For example, in an intelligent transportation scenario, the radar device 100 can be installed on intelligent monitoring equipment or intelligent transportation equipment. Intelligent monitoring equipment can be installed at smart intersections or high-speed gantry cranes, so its high resolution allows for high-resolution detection of vehicles on the road, enabling high-performance traffic supervision. Intelligent monitoring equipment can also be installed on roadside monitoring equipment and intelligent transportation equipment, identifying and tracking moving objects on the road, as well as identifying stationary objects (such as lane markings and signs) to improve overall road safety.

[0138] like Figure 1 As shown, the radar device 100 includes at least a transmitter 110, a receiver 120, a signal processor 130, and an antenna 140. The transmitter 110 is configured to transmit a radar wave signal via the antenna 140. The receiver 120 is configured to receive an echo signal of the radar wave signal via the antenna 140. Here, the echo signal of the radar wave signal is the signal generated by the radar wave signal reflecting off a target within the detection area of ​​the device 100. The signal processor 130 is connected to the receiver 120 and is configured to process the echo signal.

[0139] In one possible implementation, transmitter 110 may transmit an ultra-wideband radar wave signal, such as an ultra-wideband linear frequency modulation signal. For example, the transmission bandwidth of the radar signal may be greater than 4 GHz to obtain high-resolution radar imaging results with a resolution of less than 0.04 m in the range dimension (i.e., the range direction).

[0140] In another possible implementation, the transmitter 110 may also transmit a stepped-frequency multi-subband signal. Since the stepped-frequency multi-subband signal can be synthesized into an ultra-wideband signal, high-resolution radar imaging results in the range dimension can also be obtained by transmitting the stepped-frequency multi-subband signal.

[0141] In the embodiment of the present application, the receiver 120 can obtain echo signals of multiple sub-bands. By synthesizing the echo signals of the multiple sub-bands obtained by the receiver 120 into an ultra-wideband signal, high-resolution imaging in the range dimension can be achieved. Figure 2The comparison results of the range profiles of a single sub-band signal before bandwidth synthesis and the broadband signal after synthesis of the stepped frequency multi-sub-band signal are shown. It can be seen that the radar range dimension resolution of the target is higher after bandwidth synthesis, and adjacent structural components in the target can be effectively separated, which is beneficial to the subsequent target detection and recognition.

[0142] In one possible implementation, when transmitter 110 transmits an ultra-wideband radar wave signal, receiver 120 can receive the ultra-wideband echo signal via antenna 140. In this case, receiver 120 cannot directly acquire the wide-bandwidth signal due to the limited ADC acquisition rate. Instead, receiver 120 can filter the ultra-wideband echo signal to divide it into multiple sub-band signals, thereby obtaining multi-sub-band echo signals. For example, receiver 110 can filter the ultra-wideband echo signal using multiple filters with different filter bandwidths to obtain multi-sub-band echo signals.

[0143] In a possible implementation, when the transmitter 110 transmits a stepped-frequency multi-sub-band signal, the receiver 120 may receive an echo signal of the stepped-frequency multi-sub-band through the antenna 140 .

[0144] In an embodiment of the present application, the signal processor 130 is used to process the multi-subband echo signals obtained by the receiver 120, for example, to estimate and compensate for the phase error within the sub-band, estimate and compensate for the phase error between sub-bands, or perform coherent synthesis of the multi-subbands, radar imaging, and other processing, which is not limited in this embodiment of the present application.

[0145] Figure 3 FIG2 is a schematic block diagram of another radar device 300 to which the signal processing method provided in an embodiment of the present application is applied. The device 300 may be a specific example of the device 100.

[0146] like Figure 3 As shown, the apparatus 300 may include a transmitter 301, a local oscillator 302, a waveform generator 303, an antenna 304, a receiver 305, a low noise amplifier 306, a mixer 307, a filter 308, an orthogonal demodulator 309, an A / D sampler 310, and a signal processor 311. The transmitter 301, the local oscillator 302, and the waveform generator 303 may be included in the transmitter as Figure 1 A specific example of the transmitter 110 in FIG. The receiver 305, the low noise amplifier 306, the mixer 307, the filter 308, the orthogonal solution 309, the A / D sampling 310, etc. can be included in the receiver as Figure 1 A specific example of the receiver 120 in FIG. The signal processor 311 is Figure 1 A specific example of the signal processor 130 in FIG.

[0147] The waveform generator 303 can be used to generate a linear frequency modulation waveform for radar transmission, such as an ultra-wideband signal. The local oscillator 302 is used to provide a local oscillator signal with a fixed oscillation frequency. For example, the linear frequency modulation waveform can be up-converted to a corresponding transmission frequency using the local oscillator signal. The transmitter 301 is used to transmit the up-converted radar wave signal via the antenna 304.

[0148] Receiver 305 receives radar echo signals via antenna 304. Low-noise amplifier (LNA) 306 amplifies the high-frequency or intermediate-frequency (IF) echo signals received by receiver 305. Mixer 307 mixes the high-frequency echo signals received by receiver 305 with the local oscillator frequency to convert them to an IF. Filter 308 filters and divides the radar echo signals into subbands. Quadrature demodulator 309 converts the IF output signals into orthogonal baseband signals, namely, I and Q components, to obtain the amplitude and phase of the echo signals. A / D sampler 310 converts analog signals directly into digital signals. Signal processor 311 processes the digital signals generated by A / D sampler 311.

[0149] In one possible implementation, Figure 3 As shown, the number of filters 308 can be N, each filter being used to select echo signals of different frequency bands. Where N is a positive integer greater than 1. Accordingly, the number of orthogonal demodulators 309 and A / D samplers is the same as that of filters 308, and they are respectively used to perform orthogonal demodulation and A / D sampling on the sub-bands obtained by one filter.

[0150] exist Figure 1 or Figure 3 In the radar device shown, after obtaining the multi-subband echo signals, the signal processing unit needs to perform sub-band splicing on the multi-subband echo signals to synthesize the ultra-wideband echo signals to obtain high-resolution radar imaging results in the distance dimension. Figures 4 to 11 Describes the process of subband splicing.

[0151] Figure 4 4 shows a schematic flow chart of a method 400 for acquiring a radar image. As an example, the method 400 may be executed by a radar device, such as Figure 1 The radar device 100 or Figure 3 The radar device 300 in FIG. Furthermore, the method 400 may be executed by a signal processing unit in the radar device. Alternatively, in other embodiments, the method 400 may be executed by a processing unit external to the radar device, such as an onboard computing system or a cloud server, which is not limited in this application.

[0152] It should be understood that Figure 4The steps or operations of the method for obtaining a radar image are shown, but these steps or operations are only examples. The embodiment of the present application may also perform other operations or Figure 4 In addition, Figure 4 The steps in Figure 4 are executed in a different order than the ones presented, and may not be executed Figure 4 All operations in . Figure 4 As shown, method 400 includes steps 401 to 408 .

[0153] 401. Acquire multi-subband echo signals.

[0154] As an example, the signal processing unit may obtain multi-subband echo signals from a plurality of A / D samplers.

[0155] Figure 5 Three schematic diagrams of bandwidth distribution of multi-subband echo signals that can be obtained by embodiments of the present application are shown. The multi-subband signal in (a) is a multi-subband signal in a sub-band overlapping frequency band mode, the multi-subband signal in (b) is a multi-subband signal in a sub-band adjacent mode, and the multi-subband signal in (c) is a multi-subband signal in a sub-band spacing mode. Figure 5 It can be seen that the multi-subband signals in Figure (a) have overlapping frequency bands, while the multi-subband signals in Figures (b) and (c) do not have overlapping frequency bands.

[0156] 402: Perform channel amplitude calibration. Specifically, channel amplitude calibration may be performed on each sub-band echo signal in the multi-sub-band echo signals.

[0157] In some embodiments, due to errors in the radar signal's echo signal during RF transmission and conversion within radar hardware (e.g., receiver, low-noise amplifier, filter, etc.), the sub-band echo signal may have undesirable amplitude and phase characteristics within the sub-band. In this case, channel amplitude calibration can be performed on the sub-band echo signal to compensate for the envelope level of the sub-band echo signal.

[0158] As an example, a range spectrum distribution curve function can be obtained by superimposing the range frequency domain signals along the azimuth direction, performing statistical and high-order smoothing fitting. The spectrum distribution curve function can then be inverted to obtain a spectrum amplitude error compensation function. This compensation function can be used to compensate for the spectrum amplitude of the sub-band echo signals, achieving intra-channel amplitude calibration and obtaining an ideal radar response function, which can be, for example, a gate function.

[0159] Figure 6 Figure (a) shows an example of the amplitude and phase characteristics within a sub-band before channel amplitude calibration, where the spectrum amplitude of the sub-band is not flat and has severe fluctuations. Figure 6Figure (b) shows an example of the amplitude and phase characteristics within a subband after channel amplitude calibration. As can be seen, the spectral amplitude within the subband becomes flat after calibration. Therefore, by performing channel amplitude calibration on the subband echo signal, spectral amplitude errors within the subband can be compensated, achieving ideal subband amplitude and phase characteristics, which helps improve the subsequent subband splicing results. Conversely, if spectral amplitude errors within the subband are not compensated, the subsequent subband splicing results will be affected.

[0160] It should be noted that after channel amplitude calibration is performed on multiple sub-band echo signals, the amplitude of each sub-band echo signal is the same. In some possible implementations, after channel amplitude calibration is performed on each sub-band echo signal, the amplitudes of the multiple sub-band echo signals may be normalized so that the amplitudes of each sub-band echo signal are the same.

[0161] 403. Spectrum shifting is performed. Specifically, the multi-subband echo signals may be rearranged in sequence according to the order of carrier frequencies.

[0162] As an example, each subband echo signal can be upsampled multiple times. One possible implementation is to perform N-fold upsampling when N subband echo signals are obtained, where N is a positive integer greater than 1. Typically, the subband signal is upsampled by transferring it to the range-frequency domain and performing zero padding on both ends of the frequency domain (the zero padding length can be equal to the number of range points multiplied by (N-1)). The signal is then multiplied by a conventional transfer function, thereby rearranging the multiple subband signals in the order of the carrier frequency points, preparing for subsequent error estimation and subband splicing.

[0163] Figure 7 Three schematic diagrams are shown showing multiple subband echo signals arranged in order of carrier frequencies. (a) shows multiple subband signals in a subband overlapping mode, where the carrier frequencies corresponding to the subbands k=1, k=2, and k=3 increase in sequence, and there is an overlapping frequency band between the subband k=1 and the subband k=2, and between the subband k=2 and the subband k=3. (b) shows multiple subband signals in a subband adjacent mode, where the carrier frequencies corresponding to the subbands k=1, k=2, k=3, and k=4 increase in sequence, and the subbands k=1 and k=2, k=2 and k=3, and k=3 and k=4 are adjacent, i.e., there are no overlapping frequency bands. Figure (c) shows multiple sub-band signals in the sub-band spacing mode, where the carrier frequencies corresponding to the k=1 and k=2 sub-bands increase successively, and there is a frequency band spacing between the k=1 sub-band and the k=2 sub-band.

[0164] In some embodiments, after envelope-level compensation (i.e., step 402) and spectrum shifting (i.e., step 403) are performed on the sub-band echo signals, it is necessary to estimate and compensate for the phase-level error (i.e., phase error) of the sub-band echo signals. The phase error may include high-order phase errors within the sub-band and low-order phase errors between sub-bands. The low-order phase error between sub-bands includes a first-order phase error (also known as a first-order linear phase error) and a zero-order phase error (commonly referred to as a constant phase error). Steps 404 to 406 below describe the process of estimating and compensating for the high-order phase error, first-order phase error, and constant phase error of the sub-band echo signals, respectively.

[0165] 404. Estimation and compensation of high-order phase errors: Specifically, estimation and compensation of high-order phase errors in each sub-band echo signal may be performed.

[0166] In one possible implementation, a phase optimization algorithm based on range energy contrast enhancement can be used to estimate high-order phase errors. For example, the image energy contrast can be used as a measure of image focus, and the estimated value of the phase error can be continuously adjusted to perform an optimization operation. Accordingly, the range high-order phase error corresponding to the maximum image contrast function is the desired high-order phase error. This optimization process is an optimization problem with high-order phase error as the independent variable and image contrast as the cost function. As a specific example, contrast can be defined as the ratio of the amplitude variance of the data of each azimuth unit of the SAR image to the square of the mean. Through the contrast optimization operation, the range pulse pressure result of the single sub-band echo signal can be well focused (for example, it can be manifested as symmetrical low side lobes distributed around the main peak).

[0167] In another possible implementation, high-order phase errors can be estimated based on the radar device's internal calibration signal. In the radar device, a calibration loop can be configured in the radar transmitter. The radar device transmits an internal loop signal, which is then received by the radar receiver. The internal calibration signal passes through the radar's transceiver link and has the same transceiver link characteristics as the measured echo signal. Therefore, high-order phase errors can be extracted based on the received internal calibration signal, and high-order phase errors within the sub-band echo signal can be compensated.

[0168] 405. Estimation and compensation of first-order phase errors: Specifically, estimation and compensation may be performed on first-order phase errors between adjacent sub-bands in the multi-sub-band echo signal.

[0169] In one possible implementation, the presence of first-order linear phase error causes the envelopes of the same target after pulse compression in different sub-band echo signals to be at different distances, making coherent synthesis impossible. Therefore, the first-order linear phase error can be obtained by calculating the offset of the envelopes of two adjacent sub-band echo signals. For example, the offset of the envelopes of the same target in different sub-band echo signals can be estimated using energy correlation. This offset can then be converted into a first-order linear phase to compensate for the sub-band echo signals. For example, the offset can be compensated using an envelope shift function, thereby compensating for the first-order phase error within the low-order phase error of the sub-band echo signals.

[0170] In some possible designs, multiple strong scattering points can be selected and their offsets estimated using the energy correlation method. The resulting offsets are then averaged to obtain a mean offset. This mean offset can contain both integer and fractional components. By compensating for the first-order phase error between subbands, the same scattering point is distributed within the same range bin in adjacent subbands.

[0171] 406. Estimating and compensating for constant phase errors. Specifically, the constant phase errors between adjacent sub-bands in the multi-sub-band echo signal may be estimated and compensated separately.

[0172] The constant phase error between adjacent sub-bands can cause the mainlobe of the resulting bandwidth composite signal to split and the sidelobes to increase, directly impacting the quality of the sub-band splicing. In the most severe cases, when the constant phase error is large enough, the mainlobe of the resulting bandwidth composite signal can split into two mainlobes, resulting in a target with both a mainlobe and a split pseudo-mainlobe, severely impacting radar imaging quality.

[0173] As can be seen from this, when adjacent subbands with a constant phase error are synthesized and superimposed to obtain a spliced ​​bandwidth composite signal, the correlation peak points in the spliced ​​bandwidth composite signal (for example, the peak points corresponding to the main lobe or the left and right sidelobes) are correlated with the constant phase error between adjacent subbands before splicing. Based on this, the constant phase error between adjacent subbands can be estimated based on the relevant information about the main lobe or the left and right sidelobes in the spliced ​​bandwidth composite signal.

[0174] To estimate the constant phase error between adjacent subbands, the present invention proposes two computational models: a mainlobe splitting inverse computational model and a left and right sidelobe equalization model. The following describes the process of constant phase estimation using the two models.

[0175] (1) Inter-subband constant phase estimation based on the mainlobe splitting inverse operation model

[0176] The inverse mainlobe splitting model is used to characterize the mapping relationship between the difference between the peak point of the mainlobe (i.e., main peak) in the bandwidth composite signal and the peak point of the first sidelobe adjacent to the main peak (this difference can also be called the peak-to-sidelobe ratio) and the inter-subband constant phase error of adjacent subbands. Here, the peak point corresponding to the first sidelobe is higher than the peak point of the other second sidelobe adjacent to the main peak, that is, the first sidelobe is the second strongest peak. The first sidelobe can be the left sidelobe or the right sidelobe, without limitation.

[0177] In the inter-subband constant phase estimation scheme based on the mainlobe splitting inverse operation model, the difference between the peak point of the mainlobe and the peak point of the adjacent sidelobe (i.e., the second strongest peak) in the bandwidth synthetic signal after synthetic superposition of the target is substituted into the mainlobe splitting inverse operation model. The constant phase error between the subbands can be obtained through the mapping relationship between the difference between the peak point of the mainlobe and the peak point of the adjacent sidelobe and the inter-subband constant phase error.

[0178] Next, the process of obtaining a constant phase error based on the main lobe splitting inverse operation model will be described.

[0179] As an example, the first range-direction time domain signal of the first sub-band can be expressed as the following formula (1):

[0180]

[0181] The second range time domain signal of the second sub-band can be expressed as the following formula (2):

[0182]

[0183] Among them, R i (t q ) indicates that the i-th subband is at distance t q The distance to the time domain signal at the moment, where i∈[1,I], I represents the number of sub-bands that need to be synthesized, q∈[1,Q] represents the time of distance discrete sampling, Q represents the total time of distance discrete sampling, sin c(γTt q ) represents the signal range envelope signal, γ represents the range linear frequency modulation slope, T represents the radar transmission time period, Indicates the signal distance phase information, Represents the phase error information of the sub-band signal distance, θ err represents the phase error of the signal range between sub-bands, θ err ∈[0,2π].

[0184] It should be noted that, here, only the first range-direction time-domain signal is represented by formula (1) and the second range-direction time-domain signal is represented by formula (2) as an example for illustration, but the embodiments of the present application are not limited thereto. For example, the first range-direction time-domain signal may also be an equivalent transformation of formula (1) or other forms different from formula (1), and the second range-direction time-domain signal may also be an equivalent transformation of formula (2) or other forms different from formula (2), all of which fall within the scope of protection of the embodiments of the present application.

[0185] By directly performing bandwidth synthesis and superposition (e.g., coherent synthesis) on the first range-direction time-domain signal and the second range-direction time-domain signal containing a constant phase error, i.e., formula (1) and formula (2), a preliminary synthesis result can be obtained, as shown in the following formula (3):

[0186]

[0187] Among them, R d (t q ; θ) represents the bandwidth synthetic signal containing phase error, θ represents the constant phase error θ err The compensation value of θ∈[0,2π]. Therefore, in the above formula (3), θ is the independent variable, and the bandwidth composite signal R of the first sub-band and the second sub-band is d (t q θ) varies with the value of θ. That is, for each bandwidth composite signal, there is a specific value of θ.

[0188] Further, by deducing formula (3), we can obtain the following formula (4):

[0189]

[0190] in:

[0191] r1(t q )=sin c(2γTt q )

[0192] r2(t q )=sin c(γTt q )sin(πγTt q )

[0193] It can be understood that the above formula (3) and formula (4) both represent bandwidth synthesis signals, which can also be called third range-direction time domain signals.

[0194] In formula (4), r1(t q ) is the first set of range impulse response functions in the bandwidth synthesis signal, r2(t q ) is the second set of range impulse response functions in the bandwidth synthetic signal. Figure 8 Figure (a) shows r1(t q ) is a schematic diagram of the range pulse function response, (b) shows r2(t q ) is a schematic diagram of the distance impulse response function.

[0195] like Figure 8 As shown in Figure (a), r1(t q ) is the bandwidth of the simulation function of the original single-band signal R j (t q ) is twice the bandwidth γT, that is, 2γT. r1(t q ) is the ideal result after bandwidth synthesis. This item is the item that needs to be retained and its coefficient needs to be maximized, that is, When it is equal to 2, the independent variable θ is equal to the actual constant phase error θ err .

[0196] like Figure 8 As shown in Figure (b), r2(t q ) is a sub-band signal modulated by the sin(·) function, forming a notch at the zero position. This notch can be understood as splitting the bandwidth synthesis signal, causing the target signal to disperse, thereby reducing the signal-to-noise ratio of the image. q ) will affect the result of bandwidth synthesis and is the term we need to suppress. Suppressed to zero, the independent variable θ is equal to the actual constant phase error θ err In summary, when the independent variable θ is close to or equal to the actual constant phase error θ err When we can get the component r1(t q ), and can suppress the unwanted component r2(t q ).

[0197] In the above formula (4), due to different values ​​of θ, different (θ-θ err ) corresponding to R d (t q ; θ) result. Therefore, the above formula (4) can be rearranged to obtain the following formula (5):

[0198] R d (t q ; Δθ)=(1+e jΔθ )·r1(t q )-j·(1-e jΔθ )·r2(t q ) (5)

[0199] Among them, R d (t q; Δθ) also represents the bandwidth synthesis signal, which can also be called the third range time domain signal. Δθ=θ-θ err , represents the err The remaining constant phase error after compensation is Δθ∈[0,2π].

[0200] When Δθ increases gradually from 0, (1+e jΔθ ) gradually becomes smaller, (1-e jΔθ ) gradually increases; when Δθ gradually increases from π to 2π, (1+e jΔθ ) gradually increases, (1-e jΔθ ) gradually becomes smaller.

[0201] The following combination Figure 9 The simulation results of the bandwidth synthesis signal in the figure describe the process of R increasing from 0 to 2π. d (t q ; An example of the change of Δθ). Figure 9 The synthesized bandwidth pulse compression results in Figures (a) to (i) are the bandwidth synthesis signals, that is, the time domain signals in the third range direction.

[0202] Figure 9 Figure (a) shows the result of synthesized bandwidth pulse pressure when Δθ = 0. d (t q ;Δθ) in r1(t q ) coefficient (1+e jΔθ ) has an amplitude of 2, r1(t q ) has the largest effect ratio, r2(t q ) coefficient (1-e jΔθ ) has an amplitude of 0, r2(t q ) is the smallest. At this time, the main lobe of the synthesized range-direction time-domain signal is the largest. Figure 9 As shown in (a), the main lobe peak position is 0m at this time, and the difference between the peak point of the main lobe and the peak point of the adjacent side lobe is -13.26dB. That is, when Δθ=0, the constant phase error θ err The distance between adjacent sub-bands is compensated, and the time domain signals are perfectly synthesized and superimposed.

[0203] Figure 9 The figure (b) shows the result of synthesized bandwidth pulse pressure when Δθ=π / 4. When Δθ increases from 0 to π / 4, (1+e jΔθ ) is reduced from 2 to 1.85, r1(t q ) gradually decreases; (1-e jΔθ ) increases from 0 to 0.77, r2(t q) gradually increases. In this process, the left side lobe of the synthetic bandwidth pulse compression result gradually increases, and the main lobe energy gradually decreases. Figure 9 As shown in (b), the main peak position is 0.0183, and the difference between the peak point of the main lobe and the peak point of the adjacent left side lobe is -9.144dB.

[0204] Figure 9 Figure (c) shows the result of synthesized bandwidth pulse pressure when Δθ=π / 2. When Δθ increases from π / 4 to π / 2, (1+e jΔθ ) decreases from 1.85 to 1.41, r1(t q ) gradually decreases; (1-e jΔθ ) increases from 0.77 to 1.41, r2(t q ) gradually increases. In this process, the left side lobe in the result of the synthetic bandwidth pulse compression is further improved, and the main lobe energy is further reduced. Figure 9 As shown in (c), the main peak position is 0.0367 at this time, and the difference between the peak point of the main lobe and the peak point of the adjacent left side lobe is -5.781dB.

[0205] Figure 9 The figure (d) in the middle describes the result of synthesized bandwidth pulse pressure when Δθ=3π / 4. When Δθ increases from π / 2 to 3π / 4, (1+e jΔθ ) decreases from 1.41 to 0.77, r1(t q ) gradually decreases; (1-e jΔθ ) increases from 1.41 to 1.85, r2(t q ) gradually increases. In this process, the left side lobe in the result of the synthetic bandwidth pulse compression is further improved, and the main lobe energy is further reduced. Figure 9 As shown in (d), the main peak position is 0.0567 at this time, and the difference between the peak point of the main lobe and the peak point of the adjacent left side lobe is -2.795dB.

[0206] Figure 9 Figure (e) shows the result of synthesized bandwidth pulse pressure when Δθ=π. When Δθ increases from 3π / 4 to π, (1+e jΔθ ) decreases from 0.77 to 0, r1(t q ) has an effect ratio of 0; (1-e jΔθ ) increases from 1.85 to 2, r2(t q ) has the largest effect ratio. In this process, the left side lobe in the result of the synthesized bandwidth pulse compression is increased to the maximum value and is equal to the amplitude of the original main lobe. At this point, it can be considered that the original main lobe is split into two strong pseudo peaks. Figure 9As shown in (e), the main peak position (i.e., the position of the two strong pseudo peaks) is ±0.075, and the difference between the peak point of the main lobe and the peak point of the adjacent side lobe (i.e., the difference between the two strong pseudo peaks) is 0.

[0207] Figure 9 Figure (f) shows the result of synthesized bandwidth pulse pressure when Δθ=5π / 4. When Δθ increases from π to 5π / 4, (1+e jΔθ ) increases from 0 to 0.77, r1(t q ) gradually increases; (1-e jΔθ ) is reduced from 2 to 1.85, r2(t q ) gradually decreases. In this process, the left side lobe in the result of the synthetic bandwidth pulse compression has been transformed into the main lobe, its amplitude gradually increases, and the energy of the right side lobe gradually decreases. Figure 9 As shown in (f), the main peak position is -0.0567 at this time, and the difference between the peak point of the main lobe and the peak point of the adjacent right side lobe is -2.795dB.

[0208] Figure 9 Figure (g) shows the result of synthesized bandwidth pulse pressure when Δθ=3π / 2. When Δθ increases from 5π / 4 to 3π / 2, (1+e jΔθ ) increased from 0.77 to 1.41, r1(t q ) gradually increases; (1-e jΔθ ) decreased from 1.85 to 1.41, r2(t q ) gradually decreases. In this process, the amplitude of the main lobe in the result of the synthesized bandwidth pulse compression gradually increases, and the energy of the right side lobe gradually decreases. Figure 9 As shown in (g), the main peak position is -0.0367 at this time, and the difference between the peak point of the main lobe and the peak point of the adjacent right side lobe is -5.781dB.

[0209] Figure 9 The figure (h) in the middle describes the result of synthesized bandwidth pulse pressure when Δθ=7π / 4. When Δθ increases from 3π / 2 to 7π / 4, (1+e jΔθ ) increased from 1.41 to 1.85, r1(t q ) gradually increases; (1-e jΔθ ) decreased from 1.41 to 0.77, r2(t q ) gradually decreases. In this process, the amplitude of the main lobe in the result of the synthesized bandwidth pulse compression is further increased, and the energy of the right side lobe is gradually reduced. Figure 9 As shown in (h), the main peak position is -0.0183 at this time, and the difference between the peak point of the main lobe and the peak point of the adjacent right side lobe is -9.144dB.

[0210] Figure 9 Figure (i) shows the result of synthesized bandwidth pulse pressure when Δθ = 2π. When Δθ increases from 7π / 4 to 2π, (1+e jΔθ ) is raised from 1.85 to 2, r1(t q ) has the largest effect ratio; (1-e jΔθ ) decreases from 0.77 to 0, r2(t q ) is zero. In this process, the amplitude of the main lobe in the result of the synthetic bandwidth pulse compression is further increased to the maximum value, and the energy of the right side lobe gradually decreases to the minimum value. At this time, the peak position of the main lobe and the difference between the peak point of the main lobe and the peak point of the adjacent side lobe are all the same as Figure 9 (a) The response values ​​in the figure are the same.

[0211] according to Figure 9 From the simulation results in , we can get the mapping relationship between the residual constant phase error Δθ and the performance of the synthetic bandwidth pulse compression result, as shown in Table 1 below:

[0212] Table 1

[0213]

[0214] Further, according to Figure 9 From the simulation results in , we can get a mapping diagram of the residual constant phase error Δθ and the difference between the main lobe and the first side lobe of the synthetic bandwidth pulse compression result. Figure 10 An example of this map is shown in FIG. Figure 10 , when the residual constant phase error Δθ is π, the difference between the main lobe and the first side lobe is the smallest, that is, 0. As the residual constant phase error Δθ increases from 0 to π, the difference between the main lobe and the first side lobe decreases linearly until it reaches 0. At this time, the peak point corresponding to the first side lobe is to the left of the peak point corresponding to the main lobe, that is, the first side lobe is the side lobe adjacent to the left side of the main peak. As the residual constant phase error Δθ increases from π to 2π, the difference between the main lobe and the first side lobe increases linearly. At this time, the peak point corresponding to the first side lobe is to the right of the peak point corresponding to the main lobe, that is, the first side lobe is the side lobe adjacent to the right side of the main peak.

[0215] It should be noted that, through the above theoretical derivation process, the main lobe splitting inverse operation model can be obtained, which includes the mapping relationship between the difference between the main lobe and the first side lobe in the bandwidth synthesis signal and the residual constant phase error Δθ. In addition, the main lobe splitting inverse operation model can also include the residual constant phase error Δθ and the constant phase error θ err The mapping relationship is Δθ=θ-θ errTherefore, by substituting the difference between the main lobe and the first side lobe in the bandwidth synthesis signal into the main lobe splitting inverse operation model, the mapping relationship between the difference between the main lobe and the first side lobe in the bandwidth synthesis signal and the residual constant phase error Δθ, as well as the mapping relationship between the residual constant phase error Δθ and the constant phase error θ err The mapping relationship between sub-bands is obtained by obtaining the constant phase error θ err .

[0216] In one possible implementation, based on Figure 10 The mapping relationship in can be used to measure the amplitude value of the strongest peak, the position of the strongest peak, and the amplitude value of the second strongest peak in the synthetic bandwidth pulse pressure result for a given θ value. Then, the amplitude difference between the strongest peak and the second strongest peak can be calculated, as well as Figure 10 The mapping relationship in is used to determine the value of the residual constant phase error Δθ. Then, the residual constant phase error Δθ and the compensation value θ are substituted into Δθ=θ-θ err In the equation, we get θ err value.

[0217] Therefore, the embodiment of the present application obtains the difference between the main lobe and the first side lobe in the bandwidth synthesis signal, and is based on the main lobe splitting inverse operation model, that is, according to the mapping relationship between the difference between the main lobe and the first side lobe in the bandwidth synthesis signal and the residual constant phase error Δθ, and the residual constant phase error Δθ and the constant phase error θ err The mapping relationship between sub-bands is obtained by obtaining the constant phase error θ err .

[0218] In another possible implementation, based on Figure 10 Based on the mapping graph in , we can construct an objective function T(Δθ) as shown in the following formula (6):

[0219] T(Δθ)=min[F1(Δθ)-F2(Δθ)] (6)

[0220] Where F1(Δθ) represents the peak value of the main lobe in the synthesized bandwidth pulse compression result, and F2(Δθ) represents the peak value of the first side lobe in the synthesized bandwidth pulse compression result. That is, F1(Δθ) corresponds to the amplitude of the strongest peak in the synthesized bandwidth pulse compression result, and F2(Δθ) corresponds to the amplitude of the second strongest peak in the synthesized bandwidth pulse compression result.

[0221] In the above formula (6), when min[F1(Δθ)-F2(Δθ)] takes the minimum value (i.e., close to 0), the main lobe in the resultant composite bandwidth pulse compression is split into two lobes with substantially the same peak intensity. In this case, Δθ = π. Substituting Δθ = π into Δθ = θ - θ err , we can get the constant phase error θ between sub-bands err , that is, θ err=θ-π. It can be known that when the result of the synthesized bandwidth pulse pressure is determined, the constant phase error θ is err The compensation value θ is also determined, so we can get θ err value.

[0222] Therefore, the embodiment of the present application obtains the minimum value of the difference between the main lobe and the first side lobe in the bandwidth synthesis signal, and is based on the main lobe splitting inverse operation model, that is, when the difference between the main lobe and the first side lobe in the bandwidth synthesis signal is the minimum value, the remaining constant phase error Δθ is π, at this time θ err =θ-π, and the compensation value θ corresponding to the bandwidth synthesis signal is substituted into it to obtain the constant phase error θ between sub-bands. err .

[0223] (2) Inter-subband constant phase estimation based on left and right sidelobe equalization model

[0224] The left and right sidelobe equalization model characterizes the mapping between the difference between the peak values ​​of the left and right sidelobes in a bandwidth composite signal and the constant phase error between adjacent subbands. As shown in the simulation diagram of the bandwidth composite signal above, when the peak values ​​of the left and right sidelobes are equal and balanced, that is, when the difference between the peak values ​​of the left and right sidelobes is 0, there is no constant phase error between adjacent subbands of the composite signal, and the constant phase error of the bandwidth composite signal is 0.

[0225] In the inter-subband constant phase estimation scheme based on the left and right sidelobe equalization model, the difference between the peak points of the left and right sidelobes in the bandwidth-composite signal after the target synthesis and superposition is substituted into the left and right sidelobe equalization model. The constant phase error between the subbands can be obtained through the mapping relationship between the difference between the peak points of the left and right sidelobes and the inter-subband constant phase error.

[0226] Next, the process of obtaining a constant phase error based on the left and right sidelobe equalization models will be described.

[0227] Here, the process of obtaining a constant phase error based on the left and right sidelobe equalization models is described by using the first range-direction time domain signal of the first subband as formula (1) above and the second range-direction time domain signal of the second subband as formula (2) above.

[0228] By directly performing bandwidth synthesis and superposition (e.g., coherent synthesis) on the first range-direction time-domain signal and the second range-direction time-domain signal containing a constant phase error, i.e., formula (1) and formula (2), a preliminary synthesis result can be obtained, as shown in the following formula (7):

[0229]

[0230] Among them, Q(t q) represents the bandwidth composite signal containing phase error, which can also be called the third distance time domain signal.

[0231] Furthermore, by reorganizing formula (7), we can obtain the following formula (8):

[0232]

[0233] Here, the left adjacent side lobe and the right adjacent side lobe (i.e., the left and right side lobes) of the main lobe of the bandwidth synthesized signal and the constant phase error θ can be used. err The mapping relationship is constructed to construct the goal of equal balance of left and right sidelobes to solve the constant phase error θ err According to the sin c function characteristics, the left adjacent side lobe Q l (θ err ) is centered on the main lobe, and the left side is t q =-3 / (4γT), the right adjacent side lobe Q r (θ err ) is centered on the main lobe, on the right side t q =3 / (4γT). Combining the above formula (8), we can get Q l (θ err ) satisfies the following formula (9), Q r (θ err ) satisfies the following formula (10):

[0234]

[0235]

[0236] Among them, Q l (θ err ) means that the phase error between the adjacent side lobe on the left side of the main lobe and the constant phase error θ err The mapping relationship, Q r (θ err ) represents the phase error θ between the adjacent side lobe to the right of the main lobe and the constant err The mapping relationship.

[0237] Further sorting out formulas (9) and (10) can yield the following formulas (11) and (12):

[0238]

[0239] At this time, the left and right sidelobe equalization model P(θ err ), as shown in the following formula (13):

[0240]

[0241] P(θerr ) means the difference between the adjacent sidelobe on the left side of the main peak and the adjacent sidelobe on the right side of the main peak and the constant phase error θ err At this time, different θ err Substituting the value of into formula (13), we will get different P(θ err ) results.

[0242] Furthermore, by sorting out formula (13), we can get θ err About P(θ err ) is shown in the following formula (14):

[0243]

[0244] By directly measuring the bandwidth composite signal of adjacent sub-bands, we can obtain P(θ err ), and then substitute into formula (14) to directly solve θ err , that is, the constant phase error estimation between adjacent sub-bands is completed.

[0245] Therefore, the embodiment of the present application obtains the difference between the left adjacent side lobe and the right adjacent side lobe of the main lobe in the bandwidth synthesis signal, and is based on the left and right side lobe equalization model, that is, according to the difference between the left adjacent side lobe and the right adjacent side lobe of the main lobe in the bandwidth synthesis signal and the constant phase error θ between adjacent sub-bands. err The mapping relationship between sub-bands is obtained by obtaining the constant phase error θ err .

[0246] In some optional embodiments, the signal processing unit may store the above-mentioned mainlobe splitting inverse calculation model or the left and right sidelobe equalization model. In this way, the measured data of the mainlobe or left and right sidelobes of the measured bandwidth synthetic signal can be substituted into the corresponding calculation model to solve the corresponding constant phase error θ between adjacent subbands. err When obtaining the constant phase error between adjacent sub-bands, the embodiment of the present application can be solved in a single pass without iteration, which can help save computing resources and reduce system complexity.

[0247] To obtain the constant phase error θ between subbands err Then, the constant phase error θ can be calculated based on err , determine the constant phase error compensation function P comp (θ err ), as shown in the following formula (15):

[0248] P comp (θ err )=exp(jθ err ) (15)

[0249] Then, constant phase error compensation can be performed on adjacent subbands using formula (15), for example, compensation can be performed on the range time domain signal of the second subband. After constant phase error compensation is performed on adjacent subbands, the following steps 407 and 408 can be performed.

[0250] 407: Perform bandwidth synthesis. Specifically, bandwidth synthesis may be performed on echo signals of multiple sub-bands.

[0251] As an example, after obtaining the constant phase error between the first range-direction time domain signal of the first sub-band and the second range-direction time domain signal of the second sub-band, and compensating the first range-direction time domain signal or the second range-direction time domain signal according to the constant phase error, the compensated first range-direction time domain signal and the second range-direction time domain signal can be synthesized and superimposed to obtain a fourth range-direction time domain signal, which does not contain the constant phase error.

[0252] Figure 11 The figure shows a comparison of the range pulse envelope before and after bandwidth synthesis of the two sub-bands. (a) is a schematic diagram of the range pulse envelope of a single sub-band before bandwidth synthesis, and (b) is a schematic diagram of the range pulse envelope after bandwidth synthesis of the two sub-bands. Figure 11 As can be seen, the envelopes of the two targets in Figure (a) are indistinguishable, while in Figure (b) they can be clearly distinguished. Therefore, after bandwidth synthesis, the target resolution capability can be greatly improved.

[0253] 408. Obtain a radar image. Specifically, imaging processing can be performed based on the bandwidth synthesis result to obtain the radar image, such as a SAR image. Since constant phase errors between adjacent sub-bands are compensated, a high-resolution radar image in the range direction can be obtained.

[0254] Therefore, the embodiments of the present application obtain the peak points of the main lobe, or left and right side lobes, in the bandwidth-synthesized signal, and determine the constant phase error between adjacent subbands based on the mapping relationship between the peak points corresponding to the main lobe or left and right side lobes and the constant phase error between adjacent subbands. The constant phase error is then compensated for in the adjacent subbands based on the constant phase error, and bandwidth synthesis is performed on the compensated adjacent subbands to obtain a high-resolution radar imaging image in the range direction. Because the embodiments of the present application do not involve the overlapping subband common spectrum portion of adjacent subbands during the constant phase error determination process, the embodiments of the present application can improve spectrum utilization and are applicable to three modes: subband overlapping mode, subband adjacent mode, and subband spacing mode.

[0255] Figure 121 shows a schematic flow chart of a signal processing method 1200 provided in an embodiment of the present application. As an example, the method 1200 may be executed by a radar device, for example Figure 1 The radar device 100 or Figure 3 Furthermore, the method 1200 may be executed by a signal processing unit in the radar device. Alternatively, in other embodiments, the method 400 may also be executed by a processing unit disposed outside the radar device, such as an onboard computing system or a cloud server, etc., which is not limited in this application. Figure 12 As shown, method 1200 includes steps 1210 to 1240 .

[0256] 1210. Acquire a first range-direction time-domain signal of a first sub-band and a second range-direction time-domain signal of a second sub-band adjacent to the first sub-band.

[0257] For example, it can be achieved by Figure 4 Steps 401 to 405 in the above embodiment are used to obtain a first range-direction time domain signal of the first sub-band and a second range-direction time domain signal of the second sub-band. The first range-direction time domain signal of the first sub-band is, for example, the signal represented by formula (1) above, and the second range-direction time domain signal of the second sub-band is, for example, the signal represented by formula (2) above, which is not limited in this embodiment of the present application.

[0258] 1220. Synthesize and superimpose the first range-direction time domain signal and the second range-direction time domain signal to obtain a third range-direction time domain signal.

[0259] Exemplarily, the first range-direction time-domain signal and the second range-direction time-domain signal may be directly correlated and synthesized to obtain a synthesis result, which is the third range-direction time-domain signal.

[0260] In one example, the third range-time domain signal may be a signal represented by formula (3), (4), or (5) above. In this case, the constant phase error between the first range-time domain signal and the second range-time domain signal is compensated, and the compensation value is the independent variable θ.

[0261] In another example, the third range-direction time-domain signal may be the above formula (7) or (8).

[0262] 1230. Obtain a first peak point and a second peak point of the third range-direction time domain signal.

[0263] In one example, the first peak point may be a peak point corresponding to a main lobe, and the second peak point may be a peak point corresponding to a second peak in the third range-direction time domain signal. The second peak may be a side lobe adjacent to the left side of the main peak, or adjacent to the right side of the main peak, without limitation.

[0264] In another example, the first peak point may be a peak point corresponding to a side lobe adjacent to the left side of the main peak, and the second peak point may be a peak point corresponding to a side lobe adjacent to the right side of the main peak.

[0265] 1240. Determine a constant phase error θ between the first range-direction time-domain signal and the second range-direction time-domain signal based on the first peak point and the second peak point. err ,θ err ∈[0,2π].

[0266] Therefore, the embodiment of the present application obtains a spliced ​​bandwidth composite signal by synthesizing and superimposing the distance time domain signals of adjacent sub-bands, and obtains two peak points in the spliced ​​bandwidth composite signal, such as a first peak point and a second peak point. Since the two peak points in the bandwidth composite signal are related to the constant phase between the two adjacent sub-bands before splicing, the embodiment of the present application can obtain the constant phase error between the adjacent sub-bands based on the two peak points. Since the embodiment of the present application does not involve the overlapping sub-band common spectrum portion of adjacent sub-bands in the process of determining the constant phase error, the embodiment of the present application can improve spectrum utilization and can be applied to three modes: sub-band overlapping mode, sub-band adjacent mode, and sub-band spacing mode.

[0267] In some possible implementations, the first peak point is a peak point corresponding to a main lobe of the third range-direction time-domain signal, and the second peak point is a peak point corresponding to a first sidelobe adjacent to the main peak of the third range-direction time-domain signal, wherein the peak point corresponding to the first sidelobe is higher than the peak point corresponding to the second sidelobe adjacent to the main peak of the third range-direction time-domain signal. In other words, the second peak point is a peak point corresponding to the second highest peak in the third range-direction time-domain signal.

[0268] Wherein, the constant phase error θ between the first range-direction time domain signal and the second range-direction time domain signal is determined based on the first peak point and the second peak point. err A specific implementation method can be:

[0269] According to the difference between the first peak point and the second peak point, the residual constant phase error Δθ of the third range-direction time domain signal is determined, wherein the third range-direction time domain signal is the constant phase error θ err Compensation of the first compensation value θ is performed, Δθ=θ-θ err , Δθ∈[0,2π], θ∈[0,2π];

[0270] Determine the constant phase error θ according to the residual constant phase error Δθ and the first compensation value θ err .

[0271] In this implementation, the third range-time domain signal can be a signal represented by formula (3), (4) or (5) above. The remaining constant phase error Δθ has a mapping relationship with the difference between the first peak point and the second peak point. err There is also a mapping relationship between them, so the constant phase error θ err The difference between the first peak point and the second peak point also has a mapping relationship. Here, the constant phase error θ can be err The mapping relationship between the difference between the first peak point and the second peak point is called the main lobe splitting operation model. Figure 4 The description of step 406 is omitted here.

[0272] Therefore, the embodiment of the present application obtains the difference between the main lobe and the first side lobe (i.e., the side lobe corresponding to the second peak) in the bandwidth synthesis signal, and according to the mapping relationship between the difference between the main lobe and the first side lobe in the bandwidth synthesis signal and the residual constant phase error Δθ, and the residual constant phase error Δθ and the constant phase error θ err The mapping relationship is based on the main lobe splitting inverse operation model to obtain the constant phase error θ between sub-bands. err .

[0273] In some possible implementations, when the peak point corresponding to the first sidelobe is to the left of the peak point corresponding to the main lobe, the value range of the residual constant phase error Δθ is [0, π]. That is, in this case, the first sidelobe is a sidelobe adjacent to the left of the main lobe.

[0274] When the peak point corresponding to the first side lobe is to the right of the peak point corresponding to the main lobe, the value range of the residual constant phase error Δθ is [π, 2π]. In other words, at this time, the first side lobe is the side lobe adjacent to the right of the main lobe.

[0275] In this way, the value range of the residual constant phase error Δθ can be further obtained based on the position of the first side lobe relative to the main lobe, that is, whether the first side lobe is the left adjacent side lobe or the right adjacent side lobe, which can help to more accurately determine the corresponding residual constant phase error Δθ based on the difference between the peak point of the main lobe and the peak point of the first side lobe.

[0276] In some possible implementations, determining the residual constant phase error Δθ of the third range-direction time-domain signal according to the difference between the first peak point and the second peak point includes:

[0277] When the difference between the peak point corresponding to the main lobe of the third range-direction time-domain signal and the peak point corresponding to the first side lobe is a minimum value, the residual constant phase error Δθ is determined to be π. Here, the minimum value of the difference includes the difference being 0 and the difference being approximately 0, which is not limited in this embodiment of the present application.

[0278] Therefore, the embodiment of the present application obtains the minimum value of the difference between the main lobe and the first side lobe in the bandwidth synthesis signal (i.e., 0 or approximately 0), and when the difference between the main lobe and the first side lobe in the bandwidth synthesis signal is the minimum value, the remaining constant phase error Δθ is π, at this time θ err =θ-π, and the compensation value θ corresponding to the bandwidth synthesis signal is substituted into it to obtain the constant phase error θ between sub-bands. err .

[0279] In some possible implementations, the first peak point is a peak point corresponding to a sidelobe adjacent to the left side of the main lobe of the third range-direction time domain signal, and the second peak point is a peak point corresponding to a sidelobe adjacent to the right side of the main lobe of the third range-direction time domain signal.

[0280] Wherein, the constant phase error θ between the first range-direction time domain signal and the second range-direction time domain signal is determined based on the first peak point and the second peak point. err A specific implementation method can be:

[0281] According to the difference between the first peak point and the second peak point, and the difference and the constant phase error θ err The first mapping relationship determines the constant phase error θ err .

[0282] In this implementation, the third range-direction time domain signal may be a signal represented by formula (7) or (8) above. Here, the first mapping relationship between the constant phase error and the difference between the first peak point and the second peak point may be referred to as a left-right sidelobe equalization model. Specifically, the left-right sidelobe equalization model may be referred to above. Figure 4 The description of step 406 is omitted here.

[0283] Therefore, the embodiment of the present application obtains the difference between the left adjacent side lobe and the right adjacent side lobe of the main lobe in the bandwidth synthesis signal, and calculates the difference between the left adjacent side lobe and the right adjacent side lobe of the main lobe in the bandwidth synthesis signal and the constant phase error θ between adjacent sub-bands. err The mapping relationship is that based on the left and right sidelobe equalization models, the constant phase error θ between subbands is obtained. err .

[0284] In some possible implementations, the constant phase error θ between the first range-direction time-domain signal and the second range-direction time-domain signal is determined based on the first peak point and the second peak point. err Previously, it also included:

[0285] Get the phase error θ between the adjacent side lobe on the left side of the main lobe and the constant err A second mapping relationship, such as the mapping relationship represented by formula (9) or (11) above;

[0286] Get the phase error θ between the adjacent side lobe on the right side of the main lobe and the constant err A third mapping relationship, such as the mapping relationship represented by formula (10) or (12) above;

[0287] The first mapping relationship is determined based on the first mapping relationship and the second mapping relationship. For example, the difference between the left adjacent side lobe and the right adjacent side lobe can be calculated, and the difference is equal to the constant phase error θ err The mapping relationship is the first mapping relationship, such as the mapping relationship shown in formula (13) above.

[0288] Therefore, after obtaining the third range-direction time domain signal, the embodiment of the present application can respectively obtain the left adjacent side lobe of the main lobe of the third range-direction time domain signal and the constant phase error θ err The second mapping relationship, and the adjacent side lobe on the right side of the main lobe and the constant phase error θ err The third mapping relationship is determined, and then the first mapping relationship is determined according to the second mapping relationship and the third mapping relationship.

[0289] Some possible implementations also include:

[0290] According to the constant phase error θ err , determine the constant phase error compensation function;

[0291] Compensating the first range-direction time-domain signal or the second range-direction time-domain signal according to the constant phase error compensation function;

[0292] The compensated first range time domain signal and the second range time domain signal are synthesized and superimposed to obtain a fourth range time domain signal.

[0293] Therefore, after obtaining constant phase error compensation between adjacent subbands, the embodiments of the present application can compensate for the adjacent subbands based on the constant phase error and perform bandwidth synthesis on the compensated adjacent subbands to obtain a high-resolution radar image in the range direction. Because the common spectrum of the overlapping subbands of adjacent subbands is not involved in the constant phase error determination process, the bandwidth synthesis scheme of the embodiments of the present application can improve spectrum utilization and is applicable to three modes: overlapping subband mode, adjacent subband mode, and spaced subband mode.

[0294] In some possible implementations, before synthesizing and superimposing the first range-direction time domain signal and the second range-direction time domain signal to obtain a third range-direction time domain signal, the method further includes:

[0295] Channel amplitude calibration is performed on the first range time domain signal and the second range time domain signal respectively, for example, Figure 4 Step 402;

[0296] The first distance time domain signal and the second distance time domain signal are sequentially rearranged according to the carrier frequency point sequence, for example, Figure 4 Step 403 in

[0297] Compensate for the high-order phase errors in the sub-bands of the first range-direction time-domain signal and the second range-direction time-domain signal, respectively. Figure 4 Step 404;

[0298] Compensate for the first-order phase error between the first range-direction time-domain signal and the second range-direction time-domain signal, for example, Figure 4 Step 405 in .

[0299] Since the example of the present application calibrates the amplitude and phase characteristics within the adjacent sub-band, rearranges the signals of the adjacent sub-bands in sequence according to the carrier frequency points, and compensates for the high-order phase errors within the adjacent sub-bands and the first-order phase errors between adjacent sub-bands, before obtaining the constant phase error between adjacent sub-bands, the error between adjacent sub-bands only remains the constant phase error. Therefore, the embodiment of the present application can accurately obtain the constant phase error between adjacent sub-bands based on the mapping relationship between the correlation peak point of the synthesized bandwidth synthesis result and the remaining constant phase error.

[0300] The present application also provides a signal processing device. Figure 13Exemplarily, the signal processing device 1300 may be a radar device, or a signal processing unit disposed in the radar device, or a processing unit external to the radar device, such as an onboard computing system or a cloud server. In the embodiment of the present application, the device 1300 may include an acquisition unit 1310, a synthesis unit 1320, and a determination unit 1330.

[0301] An acquiring unit 1310 is configured to acquire a first range-direction time domain signal of a first sub-band and a second range-direction time domain signal of a second sub-band adjacent to the first sub-band;

[0302] A synthesis unit 1320 is configured to synthesize and superimpose the first range time domain signal and the second range time domain signal to obtain a third range time domain signal;

[0303] The acquiring unit 1320 is further configured to acquire a first peak point and a second peak point of the third range time domain signal;

[0304] The determining unit 1330 is configured to determine a constant phase error θ between the first range-direction time domain signal and the second range-direction time domain signal according to the first peak point and the second peak point. err ,θ err ∈[0,2π].

[0305] In some optional embodiments, the first peak point is a peak point corresponding to a main lobe of the third range-direction time domain signal, and the second peak point is a peak point corresponding to a first side lobe adjacent to the main peak of the third range-direction time domain signal, wherein the peak point corresponding to the first side lobe is higher than the peak point corresponding to the second side lobe adjacent to the main peak of the third range-direction time domain signal;

[0306] The determining unit 1330 is specifically configured to:

[0307] According to the difference between the first peak point and the second peak point, the residual constant phase error Δθ of the third range-direction time domain signal is determined, wherein the third range-direction time domain signal is a constant phase error θ err Compensation of the first compensation value θ is performed, Δθ=θ-θ err , Δθ∈[0,2π], θ∈[0,2π];

[0308] Determine the constant phase error θ according to the residual constant phase error Δθ and the first compensation value θ err .

[0309] In some optional embodiments, when the peak point corresponding to the first side lobe is to the left of the peak point corresponding to the main lobe, the value range of the residual constant phase error Δθ is [0, π];

[0310] When the peak point corresponding to the first side lobe is on the right side of the peak point corresponding to the main lobe, the value range of the residual constant phase error Δθ is [π, 2π].

[0311] In some optional embodiments, the determining unit 1330 is specifically configured to:

[0312] When the difference between the peak point corresponding to the main lobe of the third range time domain signal and the peak point corresponding to the first side lobe is a minimum value, the residual constant phase error Δθ is determined to be π.

[0313] In some optional embodiments, the first range time domain signal is expressed as the following formula:

[0314]

[0315] The second range time domain signal is expressed as follows:

[0316]

[0317] The third range time domain signal is expressed as follows:

[0318]

[0319] Among them, r1(t q )=sin c(2γTt q ), r2(t q )=sin c(γTt q )sin(πγTt q ), R d (t q ; θ) represents the third distance to the time domain signal, R i (t q ) indicates that the i-th subband is at distance t q The distance to the time domain signal at the moment, where i∈[1,I], I represents the number of sub-bands that need to be synthesized, q∈[1,Q] represents the time of distance discrete sampling, Q represents the total time of distance discrete sampling, sin c(γTt q ) represents the signal range envelope signal, γ represents the range linear frequency modulation slope, T represents the radar transmission time period, Indicates the signal distance phase information, Represents the phase error information of the inter-subband signal distance.

[0320] In some optional embodiments, the first peak point is a peak point corresponding to a side lobe adjacent to the left side of the main lobe of the third range-direction time domain signal, and the second peak point is a peak point corresponding to a side lobe adjacent to the right side of the main lobe of the third range-direction time domain signal;

[0321] The determining unit 1330 is specifically configured to:

[0322] According to the difference between the first peak point and the second peak point, and the difference and the constant phase error θ err The first mapping relationship determines the constant phase error θ err .

[0323] In some optional embodiments, the acquiring unit 1310 is further configured to:

[0324] Get the phase error θ between the adjacent side lobe on the left side of the main lobe and the constant err The second mapping relationship;

[0325] Get the phase error θ between the adjacent side lobe on the right side of the main lobe and the constant err The third mapping relationship;

[0326] The first mapping relationship is determined according to the first mapping relationship and the second mapping relationship.

[0327] In some optional embodiments, the first mapping relationship is shown in the following formula:

[0328]

[0329] Among them, P(θ err ) represents the difference between the first peak point and the second peak point.

[0330] In some optional embodiments, the first range time domain signal is expressed as the following formula:

[0331]

[0332] The second range time domain signal is expressed as follows:

[0333]

[0334] The third range time domain signal is expressed as follows:

[0335]

[0336] The side lobe Q adjacent to the left side of the main lobe of the third range time domain signal l (θ err ) satisfies the following formula:

[0337]

[0338] The side lobe Q adjacent to the right side of the main lobe of the third range time domain signal r (θerr ) satisfies the following formula:

[0339]

[0340] Among them, Q(t q ) represents the third range time domain signal, R i (t q ) indicates that the i-th subband is at distance t q The distance to the time domain signal at the moment, where i∈[1,I], I represents the number of sub-bands that need to be synthesized, q∈[1,Q] represents the time of distance discrete sampling, Q represents the total time of distance discrete sampling, sin c(γTt q ) represents the signal range envelope signal, γ represents the range linear frequency modulation slope, T represents the radar transmission time period, Indicates the signal distance phase information, Represents the phase error information of the inter-subband signal distance.

[0341] In some optional embodiments, the determining unit 1330 is further configured to determine the constant phase error θ according to the constant phase error θ. err , determine the constant phase error compensation function;

[0342] The apparatus 1300 further includes a compensation unit configured to compensate the first range time domain signal or the second range time domain signal according to the constant phase error compensation function;

[0343] The synthesis unit 1320 is further configured to synthesize and superimpose the compensated first range time domain signal and the second range time domain signal to obtain a fourth range time domain signal.

[0344] In some optional embodiments, the method further includes:

[0345] a channel amplitude calibration unit, configured to perform channel amplitude calibration on the first range-direction time domain signal and the second range-direction time domain signal respectively;

[0346] a spectrum shifting unit, configured to rearrange the first range-direction time domain signal and the second range-direction time domain signal in sequence according to the order of carrier frequency points;

[0347] a high-order phase error compensation unit, configured to compensate for high-order phase errors in sub-bands of the first range-direction time domain signal and the second range-direction time domain signal respectively;

[0348] The first-order phase error compensation unit is used to compensate for the first-order phase error between the first range-direction time domain signal and the second range-direction time domain signal.

[0349] Figure 1414 is a schematic diagram of the hardware structure of the signal processing device 1400 according to an embodiment of the present application. Figure 14 The device 1400 shown can be regarded as a computer device. The device 1400 can be used as an implementation of the signal processing device of the embodiment of the present application, and can also be used as an implementation of the signal processing method of the embodiment of the present application. The device 1400 includes a processor 1401, and optionally may also include a memory 1402, an input / output interface 1403, a communication interface 1404, and a bus 1405. Among them, the processor 1401, the memory 1402, the input / output interface 1403, and the communication interface 1404 can realize communication connection with each other through the bus 1405.

[0350] The processor 1401 can be a general-purpose central processing unit (CPU), a microprocessor, an application-specific integrated circuit (ASIC), or one or more integrated circuits for executing relevant programs to implement the functions required to be performed by the modules in the signal processing device of the embodiment of the present application, or to perform the signal processing method of the method embodiment of the present application. The processor 1401 may be an integrated circuit chip with signal processing capabilities. During implementation, the steps of the above method can be completed by hardware integrated logic circuits or software instructions in the processor 1401. The above-mentioned processor 1401 can be a general-purpose processor, a digital signal processor (DSP), an application-specific integrated circuit (ASIC), a field programmable gate array (FPGA) or other programmable logic device, a discrete gate or transistor logic device, or a discrete hardware component. The various methods, steps, and logic block diagrams disclosed in the embodiments of the present application can be implemented or executed. The general-purpose processor can be a microprocessor or the processor can also be any conventional processor. The steps of the method disclosed in conjunction with the embodiments of the present application can be directly embodied as being executed by a hardware decoding processor, or can be executed by a combination of hardware and software modules in the decoding processor. The software module can be located in a storage medium mature in the art, such as a random access memory, a flash memory, a read-only memory, a programmable read-only memory, or an electrically erasable programmable memory, a register, etc. The storage medium is located in the memory 1402, and the processor 1401 reads the information in the memory 1402 and, in combination with its hardware, completes the functions required to be executed by the modules included in the signal processing device of the embodiment of the present application, or executes the signal processing method of the method embodiment of the present application.

[0351] The memory 1402 may be a read-only memory (ROM), a static storage device, a dynamic storage device, or a random access memory (RAM). The memory 1402 may store an operating system and other application programs. When the functions required to be performed by the modules included in the signal processing apparatus of the embodiment of the present application are implemented by software or firmware, or when the signal processing method of the embodiment of the present application is executed, the program code for implementing the technical solution provided by the embodiment of the present application is stored in the memory 1402, and the processor 1401 executes the operations required to be performed by the modules included in the signal processing apparatus, or executes the signal processing method provided by the embodiment of the present application.

[0352] The input / output interface 1403 is used to receive input data and information, and output data such as operation results.

[0353] The communication interface 1404 uses a transceiver device such as, but not limited to, a transceiver to implement communication between the device 1400 and other devices or communication networks, and can serve as an acquisition module or a sending module in a processing device.

[0354] The bus 1405 may include a path for transmitting information between the various components of the device 1400 (eg, the processor 1401 , the memory 1402 , the input / output interface 1403 , and the communication interface 1404 ).

[0355] It should be noted that although Figure 14 The device 1400 shown only shows a processor 1401, a memory 1402, an input / output interface 1403, a communication interface 1404, and a bus 1405. However, in the specific implementation process, those skilled in the art should understand that the device 1400 also includes other devices necessary for normal operation, for example, it may also include a display for displaying radar imaging images. At the same time, according to specific needs, those skilled in the art should understand that the device 1400 may also include hardware devices for implementing other additional functions. In addition, those skilled in the art should understand that the device 1400 may also include only the devices necessary to implement the embodiments of the present application, and does not necessarily include Figure 14 All devices shown in .

[0356] The above-mentioned signal processing device 1300 or device 1400 can be a vehicle with radar signal processing capabilities, or other components with radar signal processing capabilities. The signal processing device 1300 or device 1400 includes, but is not limited to: an on-board terminal, an on-board controller, an on-board module, an on-board module, an on-board component, an on-board chip, an on-board unit, an on-board radar, or an on-board camera, and other sensors. The vehicle can implement the method provided in this application through the on-board terminal, on-board controller, on-board module, on-board module, on-board component, on-board chip, on-board unit, on-board radar, or camera.

[0357] The signal processing device 1300 or 1400 may also be a smart terminal other than a vehicle that has radar signal processing capabilities, or may be installed in a smart terminal other than a vehicle that has radar signal processing capabilities, or may be installed in a component of such a smart terminal. The smart terminal may be other terminal devices such as smart transportation equipment, smart home appliances, robots, etc. The signal processing device 1300 or 1400 includes, but is not limited to, the smart terminal or the controller, chip, other sensors such as radar or cameras, and other components within the smart terminal.

[0358] The signal processing device 1300 or 1400 can be a general-purpose device or a dedicated device. In a specific implementation, the device can also be a desktop computer, a portable computer, a network server, a personal digital assistant (PDA), a mobile phone, a tablet computer, a wireless terminal device, an embedded device, or other device with processing capabilities. The embodiments of the present application do not limit the type of the device.

[0359] The signal processing device 1300 or 1400 may also be a chip or processor with processing capabilities, and the signal processing device 1300 or 1400 may include multiple processors. The processor may be a single-core (single-CPU) processor or a multi-core (multi-CPU) processor. The chip or processor with processing capabilities may be disposed in the sensor, or may not be disposed in the sensor but be disposed at the receiving end of the sensor output signal.

[0360] The present application also provides a radar system for providing radar signal processing functionality for a vehicle. The system includes at least one signal processing device as described in the above embodiments of the present application. The at least one signal processing device within the system can be integrated into a complete device or apparatus, or can be independently configured as a component or apparatus.

[0361] The present application also provides a sensor system for providing radar signal processing capabilities for a vehicle. The sensor system includes at least one signal processing device as described in the above embodiments of the present application, as well as at least one other sensor, such as a camera or radar. The at least one sensor device within the system can be integrated into a complete device or apparatus, or independently configured as a component or apparatus.

[0362] An embodiment of the present application also provides a system for use in unmanned driving or intelligent driving, which includes at least one signal processing device mentioned in the above embodiments of the present application, at least one of the sensors such as cameras, radars, and other sensors. At least one device in the system can be integrated into a complete machine or equipment, or at least one device in the system can also be independently set as a component or device.

[0363] Furthermore, any of the above systems may interact with a central controller of a vehicle to provide detection and / or fusion information for decision-making or control of the vehicle's driving.

[0364] An embodiment of the present application also provides a vehicle, which includes at least one signal processing device or any of the above systems mentioned in the above embodiments of the present application.

[0365] An embodiment of the present application further provides a computer-readable storage medium, which includes a computer program. When the computer-readable storage medium is run on a computer, the computer is caused to execute the method provided in the above method embodiment.

[0366] An embodiment of the present application further provides a computer program product comprising instructions, which, when executed on a computer, enables the computer to execute the method provided in the above method embodiment.

[0367] It should be understood that in the various embodiments of the present application, the size of the serial numbers of the above-mentioned processes does not mean the order of execution. The execution order of each process should be determined by its function and internal logic, and should not constitute any limitation on the implementation process of the embodiments of the present application.

[0368] It should be understood that the first, second, etc. descriptions appearing in the embodiments of the present application are only for illustration and distinction of the description objects, and there is no order. They do not represent any special limitation on the number of devices in the embodiments of the present application, and cannot constitute any limitation on the embodiments of the present application.

[0369] Those skilled in the art will appreciate that the units and algorithm steps of each example described in conjunction with the embodiments disclosed herein can be implemented in electronic hardware, or a combination of computer software and electronic hardware. Whether these functions are performed in hardware or software depends on the specific application and design constraints of the technical solution. Professional and technical personnel can use different methods to implement the described functions for each specific application, but such implementation should not be considered beyond the scope of this application.

[0370] Those skilled in the art will clearly understand that, for the convenience and brevity of description, the specific working processes of the systems, devices and units described above can refer to the corresponding processes in the aforementioned method embodiments and will not be repeated here.

[0371] In the several embodiments provided in this application, it should be understood that the disclosed systems, devices and methods can be implemented in other ways. For example, the device embodiments described above are merely schematic. For example, the division of the units is merely a logical function division. In actual implementation, there may be other division methods, such as multiple units or components can be combined or integrated into another system, or some features can be ignored or not executed. Another point is that the mutual coupling or direct coupling or communication connection shown or discussed can be through some interfaces, indirect coupling or communication connection of devices or units, which can be electrical, mechanical or other forms.

[0372] The units described as separate components may or may not be physically separate, and the components shown as units may or may not be physical units, that is, they may be located in one place or distributed across multiple network units. Some or all of these units may be selected to achieve the purpose of this embodiment according to actual needs.

[0373] In addition, each functional unit in each embodiment of the present application may be integrated into one processing unit, or each unit may exist physically separately, or two or more units may be integrated into one unit.

[0374] If the functions are implemented in the form of software functional units and sold or used as independent products, they can be stored in a computer-readable storage medium. Based on this understanding, the technical solution of the present application, or the part that contributes to the prior art, or the part of the technical solution, can be embodied in the form of a software product. The computer software product is stored in a storage medium and includes several instructions for enabling a computer device (which can be a personal computer, a server, or a network device, etc.) to execute all or part of the steps of the method described in each embodiment of the present application. The aforementioned storage medium includes various media that can store program codes, such as a USB flash drive, a mobile hard disk, a read-only memory (ROM), a random access memory (RAM), a magnetic disk or an optical disk.

[0375] The above description is merely a specific embodiment of the present application, but the scope of protection of the present application is not limited thereto. Any changes or substitutions that can be easily conceived by a person skilled in the art within the technical scope disclosed in this application should be included in the scope of protection of this application. Therefore, the scope of protection of this application should be based on the scope of protection of the claims.

Claims

1. A signal processing method, characterized in that: include: Acquire a first range-direction time domain signal of a first sub-band and a second range-direction time domain signal of a second sub-band adjacent to the first sub-band; synthesizing and superimposing the first range-direction time domain signal and the second range-direction time domain signal to obtain a third range-direction time domain signal; Acquire a first peak point and a second peak point of the third range time domain signal; Determine the constant phase error θ between the first range time domain signal and the second range time domain signal based on the first peak point and the second peak point. err ,θ err ∈[0,2π]; Among them, the first peak point is the peak point corresponding to the main lobe of the third distance time domain signal, and the second peak point is the peak point corresponding to the first side lobe adjacent to the main peak of the third distance time domain signal, wherein the peak point corresponding to the first side lobe is higher than the peak point corresponding to the second side lobe adjacent to the main peak of the third distance time domain signal.

2. The method according to claim 1, characterized in that The constant phase error θ between the first range time domain signal and the second range time domain signal is determined based on the first peak point and the second peak point. err ,include: According to the difference between the first peak point and the second peak point, the residual constant phase error Δθ of the third range-direction time domain signal is determined, wherein the third range-direction time domain signal is a constant phase error θ err Compensation of the first compensation value θ is performed, Δθ=θ-θ err , Δθ∈[0,2π], θ∈[0,2π]; Determine the constant phase error θ according to the residual constant phase error Δθ and the first compensation value θ err .

3. The method according to claim 2, characterized in that When the peak point corresponding to the first side lobe is on the left side of the peak point corresponding to the main lobe, the value range of the residual constant phase error Δθ is [0, π]; When the peak point corresponding to the first side lobe is on the right side of the peak point corresponding to the main lobe, the value range of the residual constant phase error Δθ is [π, 2π].

4. The method according to claim 2, characterized in that The determining, based on the difference between the first peak point and the second peak point, a residual constant phase error Δθ of the third range-direction time-domain signal includes: When the difference between the peak point corresponding to the main lobe of the third range time domain signal and the peak point corresponding to the first side lobe is a minimum value, the residual constant phase error Δθ is determined to be π.

5. The method according to any one of claims 1 to 4, characterized in that The first range time domain signal is expressed as follows: The second range time domain signal is expressed as follows: The third range time domain signal is expressed as follows: Among them, r1(t q )=sinc(2γTt q ), r2(t q )=sinc(γTt q )sin(πγTt q ), R d (t q ; θ) represents the third distance to the time domain signal, R i (t q ) indicates that the i-th subband is at distance t q The distance to the time domain signal at the moment, where i∈[1,I], I represents the number of sub-bands that need to be synthesized, q∈[1,Q] represents the time of distance discrete sampling, Q represents the total time of distance discrete sampling, sinc(γTt q ) represents the signal range envelope signal, γ represents the range linear frequency modulation slope, T represents the radar transmission time period, Indicates the signal distance phase information, Represents the phase error information of the inter-subband signal distance.

6. The method according to any one of claims 1 to 4, characterized in that Also includes: According to the constant phase error θ err , determine the constant phase error compensation function; Compensating the first range-direction time-domain signal or the second range-direction time-domain signal according to the constant phase error compensation function; The compensated first range time domain signal and the second range time domain signal are synthesized and superimposed to obtain a fourth range time domain signal.

7. The method according to any one of claims 1 to 4, characterized in that Before synthesizing and superimposing the first range-direction time domain signal and the second range-direction time domain signal to obtain a third range-direction time domain signal, the method further includes: performing channel amplitude calibration on the first range-direction time domain signal and the second range-direction time domain signal respectively; Rearranging the first range-direction time domain signal and the second range-direction time domain signal in sequence according to the order of carrier frequencies; respectively compensating for high-order phase errors within sub-bands of the first range-direction time-domain signal and the second range-direction time-domain signal; A first-order phase error between the first range-direction time-domain signal and the second range-direction time-domain signal is compensated.

8. A signal processing device, characterized in that: include: An acquiring unit, configured to acquire a first range-direction time domain signal of a first sub-band and a second range-direction time domain signal of a second sub-band adjacent to the first sub-band; a synthesis unit, configured to synthesize and superimpose the first range-direction time domain signal and the second range-direction time domain signal to obtain a third range-direction time domain signal; The acquisition unit is further configured to acquire a first peak point and a second peak point of the third range time domain signal; A determining unit, configured to determine a constant phase error θ between the first range-direction time domain signal and the second range-direction time domain signal according to the first peak point and the second peak point. err ,θ err ∈[0,2π]; Among them, the first peak point is the peak point corresponding to the main lobe of the third distance time domain signal, the second peak point is the peak point corresponding to the first side lobe adjacent to the main peak of the third distance time domain signal, and the peak point corresponding to the first side lobe is higher than the peak point corresponding to the second side lobe adjacent to the main peak of the third distance time domain signal.

9. The device according to claim 8, characterized in that The determining unit is specifically configured to: According to the difference between the first peak point and the second peak point, the residual constant phase error Δθ of the third range-direction time domain signal is determined, wherein the third range-direction time domain signal is a constant phase error θ err Compensation of the first compensation value θ is performed, Δθ=θ-θ err , Δθ∈[0,2π], θ∈[0,2π]; Determine the constant phase error θ according to the residual constant phase error Δθ and the first compensation value θ err .

10. The device according to claim 9, characterized in that When the peak point corresponding to the first side lobe is on the left side of the peak point corresponding to the main lobe, the value range of the residual constant phase error Δθ is [0, π]; When the peak point corresponding to the first side lobe is on the right side of the peak point corresponding to the main lobe, the value range of the residual constant phase error Δθ is [π, 2π].

11. The device according to claim 9, characterized in that The determining unit is specifically configured to: When the difference between the peak point corresponding to the main lobe of the third range time domain signal and the peak point corresponding to the first side lobe is a minimum value, the residual constant phase error Δθ is determined to be π.

12. The device according to any one of claims 8 to 11, characterized in that The first range time domain signal is expressed as follows: The second range time domain signal is expressed as follows: The third range time domain signal is expressed as follows: Among them, r1(t q )=sinc(2γTt q ), r2(t q )=sinc(γTt q )sin(πγTt q ), R d (t q ; θ) represents the third distance to the time domain signal, R i (t q ) indicates that the i-th subband is at distance t q The distance to the time domain signal at the moment, where i∈[1,I], I represents the number of sub-bands that need to be synthesized, q∈[1,Q] represents the time of distance discrete sampling, Q represents the total time of distance discrete sampling, sinc(γTt q ) represents the signal range envelope signal, γ represents the range linear frequency modulation slope, T represents the radar transmission time period, Indicates the signal distance phase information, Represents the phase error information of the inter-subband signal distance.

13. The device according to any one of claims 8 to 11, characterized in that The determining unit is further configured to determine the constant phase error θ according to the err , determine the constant phase error compensation function; a compensation unit, configured to compensate the first range time domain signal or the second range time domain signal according to the constant phase error compensation function; The synthesis unit is further configured to synthesize and superimpose the compensated first range time domain signal and the second range time domain signal to obtain a fourth range time domain signal.

14. The device according to any one of claims 8 to 11, characterized in that Also includes: a channel amplitude calibration unit, configured to perform channel amplitude calibration on the first range-direction time domain signal and the second range-direction time domain signal respectively; a spectrum shifting unit, configured to rearrange the first range-direction time domain signal and the second range-direction time domain signal in sequence according to the order of carrier frequency points; a high-order phase error compensation unit, configured to compensate for high-order phase errors in sub-bands of the first range-direction time domain signal and the second range-direction time domain signal respectively; The first-order phase error compensation unit is used to compensate for the first-order phase error between the first range-direction time domain signal and the second range-direction time domain signal.

15. A signal processing device, characterized in that: include: A processor, configured to call and run a computer program from a memory to execute the method according to any one of claims 1 to 7.

16. The device according to claim 15, characterized in that Also includes: The memory.

17. A signal processing chip, characterized in that: The chip includes: A processor and a communication interface, wherein the processor is configured to call and run instructions from the communication interface, and when the processor executes the instructions, the method according to any one of claims 1 to 7 is implemented.

18. A computer-readable storage medium, characterized in that The computer-readable storage medium comprises a computer program, and when the computer program is run on a computer, the computer is caused to perform the method according to any one of claims 1 to 7.

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