Radar device

Abstract

The radar device includes: a signal generation circuit that generates a baseband signal; a code generation circuit that generates a plurality of code sequences; a phase rotation circuit that adds a phase rotation based on a part of the plurality of code sequences to the baseband signal to generate a plurality of transmitted signals after code multiplexing; and a plurality of transmitting antennas that respectively transmit the plurality of transmitted signals. The code length of the plurality of code sequences is greater than the code multiplexing number for the plurality of transmitted signals.

Description

Technical Field

[0001] The present disclosure relates to a radar device. Background Art

[0002] In recent years, radar devices that transmit signals using short-wavelength radars including microwaves or millimeter waves capable of obtaining high resolution have been promoted. In addition, in order to improve safety when outdoors, there is a demand for the development of radar devices that can detect small objects such as pedestrians not only in a wide angle range but also vehicles (for example, a "wide-angle radar device").

[0003] As a structure of a radar device having a wide-angle detection range, for example, there is a structure that uses the following method (direction of arrival (DOA) estimation method). In this method, a reflected wave from a target (or "object") is received by an array antenna composed of a plurality of antennas (or also referred to as "antenna elements"), and the direction of arrival of the reflected wave (or "angle of arrival") is estimated based on the reception phase difference with respect to the element interval (antenna interval).

[0004] For example, as the direction of arrival estimation method, the Fourier method (FFT (Fast Fourier Transform) method) can be cited. Or, as a method capable of obtaining high resolution, the Capon method, MUSIC (Multiple Signal Classification), and ESPRIT (Estimation of Signal Parameters via Rotational Invariance Techniques) can be cited.

[0005] In addition, as a radar device, for example, there has been proposed a structure (sometimes also referred to as "MIMO (Multiple Input Multiple Output) radar") in which a plurality of antennas (array antennas) are provided not only on the receiving side but also on the transmitting side, and beam scanning is performed by signal processing using the transceiver array antennas (for example, refer to Non-Patent Document 1).

[0006] Prior Art Documents

[0007] Patent Documents

[0008] Patent Document 1: U.S. Patent No. 9,541,638 Specification

[0009] Non-Patent Documents

[0010] Non-Patent Document 1: J. Li, and P. Stoica, "MIMO Radar with Colocated Antennas", Signal Processing Magazine, IEEE Vol. 24, Issue: 5, pp. 106 - 114, 2007

[0011] Non-Patent Document 2: M. Kronauge, H. Rohling, "Fast two-dimensional CFAR procedure", IEEE Trans. Aerosp. Electron. Syst., 2013, 49, (3), pp. 1817 - 1823

[0012] Non-Patent Document 3: Direction-of-arrival estimation using signal subspace modeling Cadzow, J. A.; Aerospace and Electronic Systems, IEEE Transactions on Volume: 28, Issue: 1 Publication Year: 1992, Page(s): 64 - 79

[0013] Non-Patent Document 4: V. Winkler, “Novel Waveform Generation Principle for Short-Range FMCW-Radars,” in Proc. German Microw. Conf., 2009, pp. 1 - 4.

[0014] Non-Patent Document 5: Y. Kozawa and H. Habuchi, “Theoretical Analysis of Atmospheric Optical DS / SS with On-Off Orthogonal M-sequence Pairs”, Sixth International Conference on Information, Communications and Signal Processing (ICICS 2007), P0686 (Dec. 2007) SUMMARY OF THE INVENTION

[0015] However, methods for detecting a target in a radar device (e.g., MIMO radar) have not been sufficiently studied.

[0016] Non-limiting embodiments of the present disclosure help to provide a radar device that improves the accuracy of target detection.

[0017] A radar device according to an embodiment of the present disclosure includes: a signal generation circuit that generates a baseband signal; a coding generation circuit that generates a plurality of coding sequences; a phase rotation circuit that adds a phase rotation based on a part of the plurality of coding sequences to the baseband signal to generate a plurality of transmitted signals after coding multiplexing; and a plurality of transmitting antennas that respectively transmit the plurality of transmitted signals; the plurality of coding sequences have a code length greater than the coding multiplexing number for the plurality of transmitted signals.

[0018] In addition, these broad or specific embodiments can be implemented by a system, a device, a method, an integrated circuit, a computer program, or a recording medium, or can be implemented by any combination of a system, a device, a method, an integrated circuit, a computer program, and a recording medium.

[0019] According to an embodiment of the present disclosure, the accuracy of target detection of a radar device can be improved.

[0020] Further advantages and effects in an embodiment of the present disclosure will be clearly presented by the description and the drawings. The above advantages and / or effects are provided by several embodiments and the features described in the description and the drawings, but it is not necessary to provide all of them in order to obtain one or more of the same features. BRIEF DESCRIPTION OF THE DRAWINGS

[0021] Figure 1 It is a block diagram showing a structural example of the radar device of Embodiment 1.

[0022] Figure 2 It is a diagram showing an example of a transmitted signal and a reflected wave signal in the case of using a chirp pulse.

[0023] Figure 3 It is a diagram showing an example of the Doppler shift amount of Embodiment 1.

[0024] Figure 4 It is a block diagram showing a structural example of the radar device of Embodiment 1.

[0025] Figure 5 It is a diagram showing an example of the computer simulation result of the aliasing determination process.

[0026] Figure 6 It is a diagram showing an example of the computer simulation result of the aliasing determination process.

[0027] Figure 7 It is a diagram showing an example of the calculation result of the extended Doppler frequency index.

[0028] Figure 8It is a block diagram showing a structural example of the radar device according to Embodiment 2.

[0029] Figure 9 It is a diagram showing an example of a transmission signal and a reflected wave signal in the case of using a chirp pulse.

[0030] Figure 10 It is a block diagram showing a structural example of the radar device according to Modification 1 of Embodiment 2.

[0031] Figure 11 It is a diagram showing an example of a transmission signal and a reflected wave signal in the case of using a chirp pulse.

[0032] Figure 12 It is a block diagram showing a structural example of the radar device according to Modification 2 of Embodiment 2.

[0033] Figure 13 It is a diagram showing an example of a transmission signal and a reflected wave signal in the case of using a chirp pulse.

[0034] Figure 14 It is a diagram showing a structural example of a radar transmission unit according to Another Modification 1.

[0035] Figure 15 It is a diagram showing a structural example of a radar transmission unit according to Another Modification 1. Detailed Embodiment

[0036] A MIMO radar transmits, for example, a signal (radar transmission wave) multiplexed using time division, frequency division, or code division from a plurality of transmission antennas (or referred to as "transmission array antennas"). Then, the MIMO radar receives, for example, a signal (radar reflected wave) reflected by surrounding objects using a plurality of receiving antennas (or referred to as "receiving array antennas"), and demultiplexes and receives the multiplexed transmission signals from each of the received signals. Through such processing, the MIMO radar can obtain the propagation path responses represented by the product of the number of transmission antennas and the number of receiving antennas, and perform array signal processing on these received signals as a virtual receiving array.

[0037] In addition, in a MIMO radar, by appropriately arranging the element intervals in the transceiver array antennas, it is possible to virtually enlarge the antenna aperture and improve the angular resolution.

[0038] Hereinafter, as an example, a MIMO radar that uses coded multiplexing transmission, which is one of the methods for simultaneously multiplexing and transmitting signals from a plurality of transmission antennas (for example, refer to Patent Document 1), is focused on.

[0039] For example, in a MIMO radar that uses coded multiplexed transmission, for each repetition of the transmitted signal (e.g., a chirp signal), phase modulation based on different coding strings (hereinafter also referred to as "codes" or "coding sequences") is repeatedly assigned to each transmitting antenna, and coded multiplexed transmission is performed by multiple (e.g., M) transmitting antennas. In addition, the MIMO radar extracts the range information of the received signal that has been coded multiplexed, for example, by performing a detection process on the signals received using multiple (e.g., N) receiving antennas.

[0040] In addition, the MIMO radar performs a Fourier transform process in M velocity directions on the range information obtained for each repetition of the transmitted signal, for example. The MIMO radar separates the received signal that has been coded multiplexed by adding a phase correction based on the detected velocity component to the result of the Fourier transform process in M velocity directions and multiplying by the inverse coding string that separates the coding strings assigned to each transmitting antenna. With such a structure of the MIMO radar, for example, even when the relative velocity between the target and the MIMO radar is not zero, the MIMO radar can suppress the mutual interference between the received signals that have been coded multiplexed and separate the received signals that have been coded multiplexed.

[0041] However, in the above structure of the MIMO radar, a Fourier transform process in multiple (e.g., M) velocity directions is performed. Therefore, in the MIMO radar, since the Fourier transform process in the velocity direction is performed at intervals of M transmission periods, the maximum Doppler frequency at which no Doppler aliasing occurs as specified by the sampling theorem is one-Mth (=1 / M) of the number of transmitting antennas M used in the coded multiplexed transmission. In the case where there is a Doppler frequency component that exceeds the maximum Doppler frequency at which no Doppler aliasing occurs as specified by the sampling theorem, the Doppler frequency cannot be determined and ambiguity occurs. Thus, compared with the transmission time when coded multiplexed transmission is not used (single-antenna transmission, M = 1), the Doppler range in which the Doppler component can be detected unambiguously is reduced to 1 / M. In addition, in other words, the "Doppler range" corresponds to the "relative velocity range of the target". In addition, in other words, the "Doppler range in which the Doppler component can be detected unambiguously" is the Doppler range in which the Doppler frequency can be determined unambiguously within this Doppler range, hereinafter referred to as the "unambiguously detectable Doppler range".

[0042] To this end, in an embodiment of the present disclosure, a method for expanding the range of Doppler frequencies that does not cause ambiguity during coded multiplexing transmission will be described. The radar device according to an embodiment of the present disclosure suppresses the generation of mutual interference between the signals multiplexed by coding, even when including Doppler variations caused by the movement of the target or the radar device, and expands the Doppler range that can be detected unambiguously to the same Doppler range as in the case of single-antenna transmission. Thus, the detection accuracy of the target can be improved within a wider Doppler frequency range.

[0043] Hereinafter, the implementation mode of an embodiment of the present disclosure will be described in detail with reference to the drawings. In addition, in the implementation mode, the same components are denoted by the same reference numerals, and the description thereof will be omitted due to repetition.

[0044] Hereinafter, a structure (in other words, a MIMO radar structure) will be described in which, in a radar device, different transmission signals multiplexed simultaneously are sent out from a plurality of transmission antennas in the transmission branch, and each transmission signal is separated and received in the reception branch for reception processing.

[0045] In addition, hereinafter, as an example, a structure of a radar method (for example, also referred to as "fast chirp modulation") using a pulse wave after frequency modulation such as a chirp pulse will be described. However, the modulation method is not limited to frequency modulation. For example, an embodiment of the present disclosure can also be applied to a radar method of a pulse compression radar that uses a pulse train phase-modulated or amplitude-modulated for transmission.

[0046] In addition, the radar device transmits the signals by coded multiplexing.

[0047] (Embodiment 1)

[0048] [Structure of Radar Device]

[0049] Figure 1 It is a block diagram showing a structural example of the radar device 10 of the present embodiment.

[0050] The radar device 10 includes a radar transmission unit (transmission branch) 100 and a radar reception unit (reception branch) 200.

[0051] The radar transmission unit 100 generates a radar signal (radar transmission signal) and uses a transmission array antenna composed of a plurality of transmission antennas 106 (for example, Nt antennas) to transmit the radar transmission signal at a prescribed transmission period.

[0052] The radar receiving unit 200 uses a receiving array antenna including a plurality of receiving antennas 202 (e.g., Na antennas) to receive the reflected wave signal, which is the radar transmission signal reflected by a target (target object, not shown). The radar receiving unit 200 performs signal processing on the reflected wave signals received by the respective receiving antennas 202, such as detecting the presence or absence of a target object, or estimating the arrival distance, Doppler frequency (in other words, relative speed), and arrival direction of the reflected wave signal, and outputs information related to the estimation result (in other words, positioning information).

[0053] In addition, the target is an object to be detected by the radar device 10, and includes, for example, vehicles (including four-wheeled and two-wheeled vehicles), people, boulders, or curbstones.

[0054] [Structure of the radar transmitting unit 100]

[0055] The radar transmitting unit 100 includes a radar transmission signal generation unit 101, a coding generation unit 104, a phase rotation unit 105, and a transmitting antenna 106.

[0056] The radar transmission signal generation unit 101 generates a radar transmission signal (in other words, a baseband signal). The radar transmission signal generation unit 101 includes, for example, a modulation signal generation unit 102 and a VCO (Voltage Controlled Oscillator) 103. Hereinafter, each structural unit in the radar transmission signal generation unit 101 will be described.

[0057] For example, as shown in the upper layer of Figure 2 , the modulation signal generation unit 102 generates a sawtooth-shaped modulation signal (in other words, a modulation signal for VCO control) at the radar transmission period Tr.

[0058] The VCO 103 outputs a frequency-modulated signal (hereinafter, for example, referred to as a "frequency chirp signal" or a "chirp signal") to the phase rotation unit 105 and the radar receiving unit 200 (a mixer unit described later) based on the radar transmission signal (modulation signal) output from the modulation signal generation unit 102.

[0059] The coding generation unit 104 generates different codes for the transmitting antenna 106 that performs coded multiplexing transmission. The coding generation unit 104 outputs a phase rotation amount corresponding to the generated code to the phase rotation unit 105. In addition, the coding generation unit 104 outputs information related to the generated code to the radar receiving unit 200 (an output switching unit 209 described later).

[0060] The phase rotation unit 105 adds the phase rotation amount input from the encoding generation unit 104 to the chirp signal input from the VCO 103, and outputs the phase-rotated signal to the transmission antenna 106. For example, the phase rotation unit 105 includes a phaser, a phase modulator, etc. (not shown). The output signal of the phase rotation unit 105 is amplified to a specified transmission power and radiated into space from each transmission antenna 106. In other words, the radar transmission signal is encoded and multiplexed for transmission from the plurality of transmission antennas 106 by being added with the phase rotation amount corresponding to the encoding.

[0061] Next, an example of the encoding (for example, orthogonal code) set in the radar device 10 will be described.

[0062] The encoding generation unit 104 generates different encodings for the transmission antennas 106 that perform encoded multiplexing transmission, for example.

[0063] For example, hereinafter, the number of transmission antennas 106 that perform encoded multiplexing transmission is set to "Nt", and the encoding multiplexing number is set to "N" CM ". In Figure 1 ", N CM = Nt.

[0064] The encoding generation unit 104 sets N allcode (hereinafter, sometimes also referred to as "N allcode (Loc)") of the N CM orthogonal codes in the encoding sequence of code length (in other words, the number of encoding elements) Loc (for example, an orthogonal code sequence in an orthogonal relationship with each other (or simply referred to as "encoding" or "orthogonal code")) as the encoding for encoded multiplexing transmission.

[0065] For example, the encoding multiplexing number N CM is less than the number of orthogonal codes N allcode , N CM < N allcode . In other words, the code length Loc of the orthogonal code is greater than the encoding multiplexing number N CM . For example, the N CM orthogonal codes of code length Loc are expressed as Code ncm = [OC ncm (1), OC ncm (2),..., OC ncm (Loc)]. Here, "OC ncm (noc)" represents the noc-th encoding element in the ncm-th orthogonal code Code ncm . In addition, "ncm" represents the index of the orthogonal code used for encoding multiplexing, ncm = 1,..., N CM . In addition, "noc" is the index of the encoding element, noc = 1,..., Loc.

[0066] Here, N of the code length Loc allcode among the allcode N CM orthogonal codes are not used by the encoding generation unit 104 (in other words, not used for encoding multiplexing transmission). Hereinafter, the (N allcode -N CM ) orthogonal codes not used by the encoding generation unit 104 are referred to as "unused orthogonal codes". At least one of the unused orthogonal codes is used, for example, for aliasing determination of the Doppler frequency in the aliasing determination unit 212 of the radar receiving unit 200 described later (an example will be described later).

[0067] By using the unused orthogonal codes, the radar device 10 can, for example, separately receive each signal that is encoded and multiplexed and transmitted from the plurality of transmission antennas 106 in a state where inter-symbol interference is suppressed, and can expand the range of detectable Doppler frequencies (an example will be described later).

[0068] As described above, the N CM orthogonal codes generated in the encoding generation unit 104 are, for example, mutually orthogonal codes (in other words, uncorrelated codes). For example, in the orthogonal code sequence, Walsh-Hadamard codes can be used. The code length of the Walsh-Hadamard code is a power of 2, and each Walsh-Hadamard code of the code length includes the same number of orthogonal codes as the code length. For example, the Walsh-Hadamard codes of code lengths 2, 4, 8, or 16 include 2, 4, 8, or 16 orthogonal codes, respectively.

[0069] Hereinafter, as an example, the code length Loc of the orthogonal code sequence with the number of encodings being N CM is set in a manner that satisfies the following formula (1).

[0070]

[0071] Here, ceil[x] is an operator (ceiling function) that outputs the smallest integer greater than or equal to the real number x. In the case of the Walsh-Hadamard code of code length Loc, the relationship N allcode (Loc)=Loc holds. For example, the Walsh-Hadamard codes of code lengths Loc = 2, 4, 8, or 16 include 2, 4, 8, or 16 orthogonal codes, respectively. Therefore, N allcode (2)=2, N allcode (4)=4, N allcode (8)=8, and N allcode (16)=16 hold. The encoding generation unit 104 uses, for example, N allcode (Loc) of the N CM orthogonal codes included in the Walsh-Hadamard code of code length Loc.

[0072] Here, a longer code length will be described. For example, when the moving speed of a target or a conventional radar device includes acceleration, the longer the code length, the more susceptible it is to inter-symbol interference. In addition, the longer the code length, the larger the candidate Doppler aliasing range in the subsequent Doppler aliasing determination. Therefore, when there are multiple targets with Doppler frequencies in different aliasing ranges within the same range index, the probability of repetition of the Doppler frequency indices detected in different aliasing ranges increases, and the probability that a conventional radar device has difficulty appropriately determining aliasing increases.

[0073] Therefore, from the perspective of the computational amount of aliasing determination in the aliasing determination unit 212 of the radar receiving unit 200 described later, the radar device 10 can also use a coding with a shorter code length. As an example, the radar device 10 can also use an orthogonal code sequence with the shortest code length among the code lengths Loc that satisfy Equation (1).

[0074] In addition, when the Walsh-Hadamard code of the code length Loc includes encodings [OC(1), OC(2),..., OC(Loc - 1), OC(Loc)] of the code length Loc, the Walsh-Hadamard code of the code length Loc also includes the encoding [OC(1), -OC(2),..., OC(Loc - 1), -OC(Loc)], where the odd-numbered encoding elements of this encoding are the same, and the signs of the even-numbered encoding elements are reversed.

[0075] Moreover, even for other encodings different from the Walsh-Hadamard code of the code length Loc, for example, when including encodings [OC(1), OC(2),..., OC(Loc - 1), OC(Loc)] of the code length Loc, the encoding of the code length Loc can be the encoding [OC(1), -OC(2),..., OC(Loc - 1), -OC(Loc)], where the odd-numbered encoding elements are the same, and the signs of the even-numbered encoding elements are reversed; or it can be the encoding [-OC(1), OC(2),..., -OC(Loc - 1), OC(Loc)], where the even-numbered encoding elements are the same, and the signs of the odd-numbered encoding elements are reversed.

[0076] When the number of unused orthogonal codes (N allcode - N CM)When it is 2 or more, the radar device 10 can also select codes in such a way that a group of codes not including the above relationship is not used in the orthogonal codes. For example, in the group of codes with the above relationship, it is also possible that the code of one is used for code multiplexing transmission, while the code of the other is included in the unused orthogonal codes. By selecting such unused orthogonal codes, it is possible to improve the aliasing determination accuracy of the Doppler frequency in the aliasing determination unit 212 of the radar receiving unit 200 described later (an example will be described later).

[0077] Hereinafter, an example of the orthogonal codes in each code multiplexing number N CM will be described.

[0078] <N CM = 2 or 3 case>

[0079] When N CM = 2 or 3, for example, Walsh-Hadamard codes with code lengths Loc = 4, 8, 16, 32,... can also be applied. In the case of these code lengths Loc, N CM < N allcode (Loc). In addition, when the code multiplexing number is N CM = 2 or 3, the Walsh-Hadamard code with the shortest code length among these code lengths Loc can be used (for example, Loc = 4).

[0080] For example, the Walsh-Hadamard code with code length Loc is described as "WH Loc (nwhc)". In addition, nwhc represents the code index included in the Walsh-Hadamard code with code length Loc, and nwhc = 1,..., Loc. For example, the Walsh-Hadamard code with code length Loc = 4 includes orthogonal codes WH4(1) = [1, 1, 1, 1], WH4(2) = [1, -1, 1, -1], WH4(3) = [1, 1, -1, -1], and WH4(4) = [1, -1, -1, 1].

[0081] Here, WH4(1) = [1, 1, 1, 1] and WH4(2) = [1, -1, 1, -1] in the Walsh-Hadamard code with code length Loc = 4 are a group of codes in which the odd-numbered code elements are the same for each other, while the even-numbered code elements are coded in reverse. In addition, WH4(3) = [1, 1, -1, -1] and WH4(4) = [1, -1, -1, 1] are also a group of codes with the same relationship as the group of WH4(1) and WH4(2).

[0082] For example, when the number of unused orthogonal codes (N allcode - N CM ) is 2 or more, the radar device 10 can also select codes in such a way that a group of codes not including such a relationship is not used in the unused orthogonal codes.

[0083] For example, in the coding multiplexing number N CM = 2, the code generation unit 104 determines two orthogonal codes in the Walsh-Hadamard code with code length Loc = 4 as the codes for code multiplexing transmission. In this case, the number of unused orthogonal codes (N allcode -N CM ) is 2.

[0084] For example, the code generation unit 104 may select the code for code-multiplexed transmission so that the unused orthogonal codes do not include the code group of WH4(1) and WH4(2) or the code group of WH4(3) and WH4(4). For example, the combination of codes (Code1 and Code2) for code-multiplexed transmission may be a combination of Code1=WH4(1) (=[1,1,1,1]) and Code2=WH4(3) (=[1,1,-1,-1]), a combination of Code1=WH4(1) and Code2=WH4(4), a combination of Code1=WH4(2) and Code2=WH4(3), or a combination of Code1=WH4(2) and Code2=WH4(4).

[0085] In addition, when the coding multiplexing number N CM =2, for example, the aliasing determination unit 212 in the radar receiving unit 200 may be N with a code length of Loc=4. allcode = 2 of the 4 Walsh-Hadamard codes not used by the code generation unit 104 (in other words, not used for code-multiplexed transmission) (= N allcode -N CM ) At least one of the unused orthogonal codes is used for aliasing determination (an example will be described later).

[0086] Next, the code length Loc N allcode The unused orthogonal codes in the orthogonal codes are recorded as "UnCode nuc =[UOC nuc (1),UOC nuc (2),…,UOC nuc (Loc)]". In addition, UnCode nuc Indicates the nucth unused orthogonal code. In addition, nuc represents the index of the unused orthogonal code, nuc = 1, ..., (N allcode -N CM ). In addition, UOC nuc (noc) represents the nucth unused orthogonal code UnCode nuc In addition, noc represents the index of the coding element, noc=1, ..., Loc.

[0087] For example, when the code multiplexing number is N CM = 2, and the codes for code multiplexing transmission determined by the code generation unit 104 are Code1 = WH4(1) (= [1, 1, 1, 1]) and Code2 = WH4(3) (= [1, 1, -1, -1]), the unused orthogonal codes are UnCode1 = WH4(2) (= [1, -1, 1, -1]) and UnCode2 = WH4(4) (= [1, -1, -1, 1]). In addition, the combination of the unused orthogonal codes (UnCode1 and UnCode2) is not limited to the combination of WH4(2) and WH4(4), and can also be a combination of other codes.

[0088] Similarly, when the code multiplexing number N CM = 3, the code generation unit 104 determines, for example, three orthogonal codes among the Walsh-Hadamard codes with a code length Loc = 4 as the codes for code multiplexing transmission. In this case, the number of unused orthogonal codes (N allcode - N CM ) is one.

[0089] For example, the code generation unit 104 can also select Code1 = WH4(3) = [1, 1, -1, -1], Code2 = WH4(4) = [1, -1, -1, 1], and Code3 = WH4(2) = [1, -1, 1, -1].

[0090] In addition, the aliasing determination unit 212 of the radar receiving unit 200 uses one of the N allcode = 4 Walsh-Hadamard codes with a code length Loc = 4 (= N allcode - N CM ) unused orthogonal code for aliasing determination (an example will be described later). For example, when the code multiplexing number is N CM = 3, and the codes for code multiplexing transmission determined by the code generation unit 104 are Code1 = WH4(3) = [1, 1, -1, -1], Code2 = WH4(4) = [1, -1, -1, 1], Code3 = WH4(2) = [1, -1, 1, -1], the unused orthogonal code is UnCode1 = WH4(1) = [1, 1, 1, 1]. In addition, the combination of the codes for code multiplexing transmission (Code1, Code2, and Code3) and the unused orthogonal code (UnCode1) is not limited to these combinations, and can also be a combination of other codes.

[0091] <N CM = 4, 5, 6, or 7 cases>

[0092] When N CMWhen = 4, 5, 6 or 7, for example, Walsh - Hadamard codes with code lengths Loc = 8, 16, 32, … can be applied. In the case of these code lengths Loc, N CM <N allcode (Loc). Additionally, when the coding multiplexing number is N CM = 4, 5, 6 or 7, the Walsh - Hadamard code with the shortest code length among these code lengths Loc can be used (for example, Loc = 8).

[0093] For example, the Walsh - Hadamard code with code length Loc = 8 includes the following 8 orthogonal codes.

[0094] WH8(1) = [–1 –1 –1 –1 –1 –1 –1 –1],

[0095] WH8(2) = [–1 1 –1 1 –1 1 –1 1],

[0096] WH8(3) = [–1 –1 1 1 –1 –1 1 1],

[0097] WH8(4) = [–1 1 1 –1 –1 1 1 –1],

[0098] WH8(5) = [–1 –1 –1 –1 1 1 1 1],

[0099] WH8(6) = [–1 1 –1 1 1 –1 1 –1],

[0100] WH8(7) = [–1 –1 1 1 1 1 –1 –1],

[0101] WH8(8) = [–1 1 1 –1 1 –1 –1 1]

[0102] Here, WH8(1) and WH8(2) in the Walsh - Hadamard code with code length Loc = 8 are a group of codes where the odd - numbered coding elements are the same for each other's encodings, and the even - numbered coding elements are encoded in reverse. Additionally, similarly, the groups of WH8(3) and WH8(4), the groups of WH8(5) and WH8(6), and the groups of WH8(7) and WH8(8) are also groups of codes with the same relationship as the group of WH8(1) and WH8(2) for each other.

[0103] For example, as the number of unused orthogonal codes (N allcode -N CM It should be noted that there seems to be an error in the original text as "WH8(1)=[1 1 1 1 1 1 1 1]" in the original is likely incorrect. I've translated it as "-1 -1 -1 -1 -1 -1 -1 -1" according to the context of the following description about the relationship between codes. If this is not what you intended, please check and correct the original text.) is greater than 2, an example of selecting a code in a manner that does not include a group of codes having such a relationship in unused orthogonal codes, and a code for code multiplexing transmission can be selected in a manner that does not include a group of codes of WH8(1) and WH8(2), a group of codes of WH8(3) and WH8(4), a group of codes of WH8(5) and WH8(6), or a group of codes of WH8(7) and WH8(8) in unused orthogonal codes.

[0104] For example, in the coding multiplexing number N CM = 4, the code generation unit 104 determines the four orthogonal codes in the Walsh-Hadamard code with code length Loc = 8 as the codes for code multiplexing transmission. In this case, the number of unused orthogonal codes (N allcode -N CM ) is 4.

[0105] For example, in the code generation unit 104, the combination of codes (Code 1, Code 2, Code 3, and Code 4) for code-multiplexed transmission may be a combination of Code 1 = WH8(1), Code 2 = WH8(3), Code 3 = WH8(5), and Code 4 = WH8(7), or a combination of Code 1 = WH8(1), Code 2 = WH8(4), Code 3 = WH8(5), and Code 4 = WH8(8). Furthermore, the combination of codes (Code 1, Code 2, Code 3, and Code 4) for code-multiplexed transmission is not limited to these combinations.

[0106] In addition, when the coding multiplexing number N CM = 4, for example, the aliasing determination unit 212 in the radar receiving unit 200 sets the N allcode = 4 of the 8 Walsh-Hadamard codes not used by the code generation unit 104 (= N allcode -N CM ) Part or all of the unused orthogonal codes are used for aliasing determination (an example will be described later).

[0107] For example, in the coding multiplexing number N CM =4, and the codes for code multiplexing transmission determined by the code generation unit 104 are Code1=WH8(1), Code2=WH8(3), Code3=WH8(5), and Code4=WH8(7), the unused orthogonal codes are UnCode1=WH8(2), UnCode2=WH8(4), UnCode3=WH8(6), and UnCode4=WH8(8). In addition, for example, when the code multiplexing number is N CM= 4, and when the codes determined by the code multiplexing generation unit 104 for multiplexed transmission are Code1 = WH8(1), Code2 = WH8(4), Code3 = WH8(5), and Code4 = WH8(8), the unused orthogonal codes are UnCode1 = WH8(2), UnCode2 = WH8(3), UnCode3 = WH8(6), and UnCode4 = WH8(7).

[0108] Similarly, for example, in the case where the code multiplexing number N CM = 5, the code generation unit 104 determines five orthogonal codes from Walsh-Hadamard codes with a code length Loc = 8 as the codes for multiplexed transmission. In this case, the number of unused orthogonal codes (N allcode - N CM ) is three.

[0109] For example, in the code generation unit 104, the combination of the codes for multiplexed transmission (Code1, Code2, Code3, Code4, and Code5) can also be the combination of Code1 = WH8(1), Code2 = WH8(3), Code3 = WH8(5), Code4 = WH8(7), and Code5 = WH8(8), or Code1 = WH8(1), Code2 = WH8(4), Code3 = WH8(5), Code4 = WH8(7), and Code5 = WH8(8). In addition, the combination of the codes for multiplexed transmission (Code1, Code2, Code3, Code4, and Code5) is not limited to these combinations.

[0110] In the case where the code multiplexing number N CM = 5, for example, the aliasing determination unit 212 in the radar receiving unit 200 uses some or all of the three (= N allcode - N allcode - N CM ) unused orthogonal codes among the eight Walsh-Hadamard codes with a code length Loc = 8 that are not used by the code generation unit 104 for aliasing determination (an example will be described later).

[0111] For example, in the case where the code multiplexing number is N CM = 5, and the codes determined by the code generation unit 104 for multiplexed transmission are Code1 = WH8(1), Code2 = WH8(3), Code3 = WH8(5), Code4 = WH8(7), and Code5 = WH8(8), the unused orthogonal codes are UnCode1 = WH8(2), UnCode2 = WH8(4), and UnCode3 = WH8(6). Additionally, for example, in the case where the code multiplexing number is N CM=5, and the codes used for coded multiplexing transmission determined by the code generation unit 104 are Code1=WH8(1), Code2=WH8(4), Code3=WH8(5), Code4=WH8(7) and Code5=WH8(8), the unused orthogonal codes are UnCode1=WH8(2), UnCode2=WH8(3) and UnCode3=WH8(6).

[0112] Similarly, for example, in the case of coding multiplexing number N CM = 6, the code generation unit 104 determines 6 orthogonal codes in the Walsh-Hadamard code with code length Loc = 8 as the codes for code multiplexing transmission. In this case, the number of unused orthogonal codes (N allcode -N CM ) is 2.

[0113] For example, in the code generation unit 104, the combination of codes for code-multiplexed transmission (Code 1, Code 2, Code 3, Code 4, Code 5, and Code 6) may be Code 1 = WH8(1), Code 2 = WH8(2), Code 3 = WH8(3), Code 4 = WH8(4), Code 5 = WH8(5), and Code 6 = WH8(8). Furthermore, the combination of codes for code-multiplexed transmission (Code 1, Code 2, Code 3, Code 4, Code 5, and Code 6) is not limited to these combinations.

[0114] In addition, when the coding multiplexing number N CM = 6, for example, the aliasing determination unit 212 in the radar receiving unit 200 sets the N allcode = 2 of the 8 Walsh-Hadamard codes not used by the code generation unit 104 (= N allcode -N CM ) Part or all of the unused orthogonal codes are used for aliasing determination (an example will be described later).

[0115] For example, when the coding multiplexing number is N CM =6, and the codes used for coded multiplexing transmission determined by the code generation unit 104 are Code1=WH8(1), Code2=WH8(2), Code3=WH8(3), Code4=WH8(4), Code5=WH8(5) and Code6=WH8(8), the unused orthogonal codes are UnCode1=WH8(6) and UnCode2=WH8(7).

[0116] Similarly, for example, in the case of coding multiplexing number N CMWhen = 7, the coding generation unit 104 determines seven orthogonal codes in the Walsh-Hadamard code with a code length Loc = 8 as the code for coded multiplexing transmission. In this case, the number of unused orthogonal codes (N allcode -N CM ) is one.

[0117] For example, the coding generation unit 104 can select Code1 = WH8(1), Code2 = WH8(2), Code3 = WH8(3), Code4 = WH8(4), Code5 = WH8(5), Code6 = WH8(6), and Code7 = WH8(7) as the code for coded multiplexing transmission. In addition, the combination of the code for coded multiplexing transmission is not limited to these.

[0118] In addition, the aliasing determination unit 212 in the radar receiving unit 200 can use one (= N allcode -N allcode ) of the eight Walsh-Hadamard codes with a code length Loc = 8 that are not used by the coding generation unit 104 for aliasing determination (an example will be described later). CM ) unused orthogonal code for aliasing determination (an example will be described later).

[0119] For example, when the coded multiplexing number N CM = 7, and the code for coded multiplexing transmission determined by the coding generation unit 104 is Code1 = WH8(1), Code2 = WH8(2), Code3 = WH8(3), Code4 = WH8(4), Code5 = WH8(5), Code6 = WH8(6), and Code7 = WH8(7), the unused orthogonal code is UnCode1 = WH(8).

[0120] The above has described the case where the coded multiplexing number N CM = 4, 5, 6, or 7.

[0121] In addition, even when the coded multiplexing number N CM ≥ 8, the radar device 10 can determine the code for coded multiplexing transmission and the unused orthogonal code in the same manner as in the case where the coded multiplexing number N CM = 2 to 7.

[0122] For example, the coding generation unit 104 can also select N CM orthogonal codes in the Walsh-Hadamard code with the code length Loc shown in Equation (2) as the code for coded multiplexing transmission. In this case, N CM < Loc = N allcode (Loc).

[0123]

[0124] In addition, the aliasing determination unit 212 in the radar reception unit 200 can use (N allcode - N allcode ) unused orthogonal codes among the Walsh-Hadamard codes of code length Loc for aliasing determination (an example will be described later). In addition, when the number of unused orthogonal codes (N CM - N allcode ) is two or more, the encoding generation unit 104 can also select the encoding for multiplexed transmission by encoding, for example, in such a manner that the encoding elements of either the odd-numbered encoding elements or the even-numbered encoding elements among the encodings of the Walsh-Hadamard codes of code length Loc are the same, and the group of encodings in which the encoding elements of the other of the odd-numbered encoding elements and the even-numbered encoding elements have their signs reversed is not included in the unused orthogonal codes. CM ) is two or more, the encoding generation unit 104 can also select the encoding for multiplexed transmission by encoding, for example, in such a manner that the encoding elements of either the odd-numbered encoding elements or the even-numbered encoding elements among the encodings of the Walsh-Hadamard codes of code length Loc are the same, and the group of encodings in which the encoding elements of the other of the odd-numbered encoding elements and the even-numbered encoding elements have their signs reversed is not included in the unused orthogonal codes.

[0125] In other words, it can also be that either the group of encodings in which the encoding elements of either the odd-numbered encoding elements or the even-numbered encoding elements among the encodings of the Walsh-Hadamard codes of code length Loc are the same, and the group of encodings in which the encoding elements of the other of the odd-numbered encoding elements and the even-numbered encoding elements have their signs reversed is included in the unused orthogonal codes, and the other encoding is not included in the unused orthogonal codes.

[0126] In addition, the elements constituting the orthogonal code sequence are not limited to real numbers and may also include complex values.

[0127] In addition, the encoding can also be other orthogonal codes different from the Walsh-Hadamard codes. For example, the encoding can also be an orthogonal M-sequence code or a pseudo-orthogonal code.

[0128] The above has described an example of the orthogonal codes for each encoding multiplexing number N CM .

[0129] Next, an example of the phase rotation amount based on the encoding for multiplexed transmission generated by the encoding generation unit 104 will be described.

[0130] The radar device 10, for example, performs multiplexed transmission by encoding using different orthogonal codes for the transmission antennas Tx#1 to Tx#Nt for which multiplexed transmission by encoding is performed. Therefore, the encoding generation unit 104 sets, for example, in the m-th transmission cycle Tr, the phase rotation amount ψ ncm based on the orthogonal code Code ncm to be given to the ncm-th transmission antenna Tx#ncm, and outputs it to the phase rotation unit 105. Here, ncm = 1,..., N CM .

[0131] For example, as the phase rotation amount ψ ncm(m), during the transmission period of Loc times of the code length, cyclically assign the Loc coded elements OC as shown in the following formula (3) to the orthogonal code Code ncm of the Loc coded elements OC ncm (1), …, OC ncm (Loc) respectively corresponding phase amounts.

[0132] ψ ncm ψ ncm (m) = angle[OC

[0133] Here, angle(x) is an operator that outputs the radian phase of the real number x, angle(1) = 0, angle(-1) = π, angle(j) = π / 2, and angle(-j) = -π / 2. j is the imaginary unit. In addition, OC_INDEX is an orthogonal code element index indicating the elements of the orthogonal code sequence Code ncm which can be cyclically changed within the range of 1 to Loc according to the transmission period (Tr) as shown in the following formula (4).

[0134] OC_INDEX = mod(m - 1, Loc) + 1 (4)

[0135] Here, mod(x, y) is a modulus operator and is a function that outputs the remainder after dividing x by y. In addition, m = 1, …, Nc. Nc is the specified number of transmission periods for the radar device 10 to perform radar positioning (hereinafter referred to as "the number of radar transmission signal transmissions"). In addition, the radar device 10, for example, performs transmissions of the number of radar transmission signals Nc that is an integer multiple of Loc (for example, Ncode times). For example, Nc = Loc × Ncode.

[0136] In addition, the coding generation unit 104 outputs the orthogonal code element index OC_INDEX to the output switching unit 209 of the radar receiving unit 200 according to the transmission period (Tr).

[0137] The phase rotation unit 105, for example, includes phase shifters or phase modulators corresponding to the Nt transmission antennas 106 respectively. The phase rotation unit 105, for example, assigns the phase rotation amount ψ ncm (m) input from the coding generation unit 104 to the chirp signals input from the radar transmission signal generation unit 101 according to the transmission period Tr.

[0138] For example, the phase rotation unit 105 assigns, according to the transmission period Tr, the phase rotation amount based on the orthogonal code Code ncm assigned to the ncm-th transmission antenna Tx#ncm to the chirp signal input from the radar transmission signal generation unit 101 ncm(m). Here, ncm = 1, …, N CM , m = 1, …, Nc.

[0139] For example, after the output from the phase rotation unit 105 for the Nt transmission antennas 106 is amplified to a specified transmission power, it is radiated into space from the Nt transmission antennas 106 (e.g., a transmission array antenna).

[0140] As an example, the case of coded multiplexing transmission when the number of transmission antennas Nt = 3 and the coded multiplexing number N CM = 3 is described. In addition, the number of transmission antennas Nt and the coded multiplexing number N CM are not limited to these values.

[0141] For example, in the m-th transmission period Tr, the phase rotation amounts ψ1(m), ψ2(m), and ψ3(m) are output from the coding generation unit 104 to the phase rotation unit 105.

[0142] The first (ncm = 1) phase rotation unit 105 (in other words, the phaser corresponding to the first (e.g., Tx#1) transmission antenna 106) gives the chirp signal generated in the radar transmission signal generation unit 101 at each transmission period Tr a phase rotation amount as shown in the following formula (5) at each transmission period Tr. The output of the first phase rotation unit 105 is transmitted by the transmission antenna Tx#1. Here, cp(t) represents the chirp signal for each m-th transmission period Tr.

[0143] exp[jψ1(1)]cp(t), exp[jψ1(2)]cp(t), exp[jψ1(3)]cp(t),..., exp[jψ1(Nc)]cp(t) (5)

[0144] Similarly, the second (ncm = 2) phase rotation unit 105 gives the chirp signal generated in the radar transmission signal generation unit 101 at each transmission period Tr a phase rotation amount as shown in the following formula (6) at each transmission period Tr. The output of the second phase rotation unit 105 is transmitted by the transmission antenna Tx#2.

[0145] exp[jψ2(1)]cp(t), exp[jψ2(2)]cp(t), exp[jψ2(3)]cp(t),..., exp[jψ2(Nc)]cp(t) (6)

[0146] Similarly, the third (ncm = 3) phase rotation unit 105 gives the chirp signal generated in the radar transmission signal generation unit 101 at each transmission period Tr a phase rotation amount as shown in the following formula (7) at each transmission period Tr. The output of the third phase rotation unit 105 is transmitted by the transmission antenna Tx#3.

[0147] exp[jψ3(1)]cp(t), exp[jψ3(2)]cp(t), exp[jψ3(3)]cp(t),..., exp[jψ3(Nc)]cp(t) (7)

[0148] In addition, when the radar device 10 continuously performs radar positioning, the orthogonal code Code used can also be variably set for each radar positioning performed (for example, for the transmission period of Nc times (Nc × Tr)). ncm encoding.

[0149] In addition, the radar device 10 can also variably set, for example, Nt transmission antennas 106 that transmit the outputs of the phase rotation unit 105 (in other words, the transmission antennas 106 corresponding to the respective outputs of the phase rotation unit 105). For example, the correspondence between the multiple transmission antennas 106 and the coding sequence for coded multiplex transmission can also be different for each radar positioning performed by the radar device 10. For example, when the radar device 10 receives a signal affected by interference from other radars that varies depending on the transmission antenna 106, the coded multiplex signal output from the transmission antenna 106 changes for each radar positioning performed, thereby enabling the effect of randomizing the interference influence to be obtained.

[0150] The structural example of the radar transmission unit 100 has been described above.

[0151] [Structure of Radar Reception Unit 200]

[0152] In Figure 1 it, the radar reception unit 200 includes Na reception antennas 202 (for example, also described as "Rx#1 to Rx#Na"), thereby forming an array antenna. In addition, the radar reception unit 200 includes Na antenna system processing units 201-1 to antenna system processing unit 201-Na, a CFAR (Constant False Alarm Rate) unit 211, an aliasing determination unit 212, a coded multiplex separation unit 213, and a direction estimation unit 214.

[0153] Each reception antenna 202 receives the radar transmission signal after being reflected by the target, that is, the reflected wave signal, and outputs the received reflected wave signal as a reception signal to the corresponding antenna system processing unit 201.

[0154] Each antenna system processing unit 201 includes a wireless reception unit 203 and a signal processing unit 206.

[0155] The wireless receiving unit 203 includes a mixer unit 204 and an LPF (low pass filter) 205. The mixer unit 204 mixes the received reflected wave signal with the transmission signal (i.e., the chirp signal) input from the radar transmission signal generation unit 101. The LPF 205 performs LPF processing on the output signal of the mixer unit 204, thereby outputting a beat signal, and the frequency of this beat signal corresponds to the delay time of the reflected wave signal. For example, as shown in the lower layer of Figure 2 obtains the differential frequency between the frequency of the transmitted chirp signal (transmitted frequency modulation wave) and the frequency of the received chirp signal (received frequency modulation wave) as the beat frequency.

[0156] The signal processing unit 206 of each antenna system processing unit 201-z (where z is a certain value from 1 to Na) includes an AD (Analog Digital) conversion unit 207, a beat frequency analysis unit 208, an output switching unit 209, and a Doppler analysis unit 210.

[0157] The signal output from the LPF 205 (e.g., the beat signal) is converted into discrete sampling data that has been discretely sampled by the AD conversion unit 207 in the signal processing unit 206.

[0158] The beat frequency analysis unit 208 performs FFT processing on N data discrete sampling data obtained within a specified time range (range gate) according to the transmission period Tr. Thus, the signal processing unit 206 outputs a spectrum in which the peak of the beat frequency corresponding to the delay time of the reflected wave signal (radar reflected wave) appears. In addition, as the FFT processing, the beat frequency analysis unit 208 can also multiply, for example, a window function coefficient such as a Hann window or a Hamming window. In addition, the radar device 10 can suppress the side lobes generated around the beat frequency peak by using the window function coefficient. In addition, when the number of N data discrete sampling data is not a power of 2, the beat frequency analysis unit 208 can also perform FFT processing by including zero-padded data as the FFT size of a power of 2.

[0159] Here, the beat frequency response output from the beat frequency analysis unit 208 in the z-th signal processing unit 206 obtained by transmitting the m-th chirp pulse is represented by RFT z (f b , m). Here, f b represents the beat frequency index, which corresponds to the index (binary number) of the FFT. For example, f b = 0,..., N data / 2, z = 0,..., Na, m = 1,..., N C . The beat frequency index f bThe smaller it is, the smaller the delay time of the reflected wave signal (in other words, the closer the distance to the target) of the beat frequency.

[0160] In addition, the beat frequency index f b can be converted into distance information R(f b ) using the following formula (8). Therefore, hereinafter, the beat frequency index f b will also be referred to as "distance index f b ".

[0161]

[0162] Here, B w represents the frequency modulation bandwidth of the chirp signal within the range gate, and C0 represents the speed of light.

[0163] The output switching unit 209 selectively switches and outputs the output of the beat frequency analysis unit 208 for each transmission cycle to the OC_INDEX-th Doppler analysis unit 210 among the Loc Doppler analysis units 210 based on the orthogonal code element index OC_INDEX output from the coding generation unit 104. In other words, in the m-th transmission cycle Tr, the output switching unit 209 selects the OC_INDEX-th Doppler analysis unit 210.

[0164] The signal processing unit 206 has Loc Doppler analysis units 210-1 to Doppler analysis unit 210-Loc. For example, through the output switching unit 209, data is input to the noc-th Doppler analysis unit 210 for Loc transmission cycles (Loc×Tr). Therefore, the noc-th Doppler analysis unit 210 uses the data of Ncode transmission cycles among Nc transmission cycles (for example, the beat frequency response RFT z (f b , m)) to perform Doppler analysis according to the distance index f b . Here, noc is the index of the coding element, and noc = 1,..., Loc.

[0165] For example, when Ncode is a power of 2, FFT processing can also be applied in Doppler analysis. In this case, the FFT size is Ncode, and the maximum Doppler frequency that does not cause aliasing derived from the sampling theorem is ±1 / (2Loc×Tr). In addition, the Doppler frequency interval of the Doppler frequency index f s is 1 / (Ncode×Loc×Tr), and the range of the Doppler frequency index f s is f s = -Ncode / 2,..., 0,..., Ncode / 2 - 1.

[0166] For example, the output VFT of the Doppler analysis unit 210 of the z-th signal processing unit 206 z noc (f b , f s ) is represented by the following equation (9). Further, j is the imaginary unit, and z = 1 to Na.

[0167]

[0168] In addition, when Ncode is not a power of 2, for example, it is also possible to perform FFT processing by including zero-padded data as a data size (FFT size) of a power of 2. For example, when the FFT size in the Doppler analysis unit 210 in the case of including zero-padded data is set to N codewzero the output VFT of the Doppler analysis unit 210 in the z-th signal processing unit 206 z noc (f b , f s ) is represented by the following equation (10).

[0169]

[0170] Here, noc is the index of the coding element, noc = 1,..., Loc. In addition, the FFT size is N codewzero , and the maximum Doppler frequency that does not cause aliasing derived from the sampling theorem is ±1 / (2Loc×Tr). In addition, the Doppler frequency interval of the Doppler frequency index f s is 1 / (N codewzero ×Loc×Tr), and the range of the Doppler frequency index f s is f s = -N codewzero / 2,..., 0,..., N codewzero / 2 - 1.

[0171] Hereinafter, as an example, the case where Ncode is a power of 2 will be described. In addition, when zero-padding is used in the Doppler analysis unit 210, in the following description, by replacing Ncode with N codewzero , it can be applied in the same manner and the same effect can be obtained.

[0172] In addition, the Doppler analysis unit 210 may also multiply by a window function coefficient such as a Hann window or a Hamming window during FFT processing. By applying the window function, the radar device 10 can suppress the side lobes generated around the beat frequency peak.

[0173] The processing in each component of the signal processing unit 206 has been described above.

[0174] In Figure 1 the CFAR unit 211 performs CFAR processing (in other words, performs adaptive threshold determination) using the outputs of the Loc Doppler analysis units 210 of the first to the Na-th signal processing units 206, and extracts the range index f that gives the peak signal b_cfar and the Doppler frequency index f s_cfar .

[0175] The CFAR unit 211 adds the powers of the outputs VFT z noc (f b , f s ) of the Doppler analysis units 210 of the first to the Na-th signal processing units 206 in the manner of the following equation (11), and performs two-dimensional CFAR processing including the range axis and the Doppler frequency axis (equivalent to the relative velocity), or CFAR processing combining one-dimensional CFAR processing. For the two-dimensional CFAR processing or the CFAR processing combining one-dimensional CFAR processing, for example, the processing disclosed in Non-Patent Document 2 can be applied.

[0176]

[0177] The CFAR unit 211 adaptively sets a threshold, and outputs the range index f b_cfar , the Doppler frequency index f s_cfar and the received power information PowerFT(f b_cfar , f s_cfar ) greater than the threshold to the aliasing determination unit 212.

[0178] Next, an operation example of the aliasing determination unit 212 shown Figure 1 will be described.

[0179] The aliasing determination unit 212 performs aliasing determination on the Doppler component VFT b_cfar s_cfar which is the output of the Doppler analysis unit 210, based on the range index f z noc extracted in the CFAR unit 211 and the Doppler frequency index f(f b_cfar , f s_cfar ). Here, z = 1,..., Na, noc = 1,..., Loc.

[0180] The aliasing determination unit 212 performs Doppler aliasing determination processing by setting the Doppler range of the assumed target to ±1 / (2×Tr), for example.

[0181] Here, for example, when Ncode is a power of 2, the Doppler analysis unit 210 applies FFT processing to each coding element. Therefore, FFT processing is performed at a period of (Loc × Tr), using the output from the beat frequency analysis unit 208. Thus, the Doppler range without aliasing according to the sampling theorem in the Doppler analysis unit 210 is ±1 / (2Loc × Tr).

[0182] Accordingly, the target Doppler range assumed by the aliasing determination unit 212 is larger than the Doppler range without aliasing in the Doppler analysis unit 210. For example, the aliasing determination unit 212 performs aliasing determination processing assuming a Doppler range of up to Loc times the Doppler range without aliasing of the Doppler analysis unit 210, i.e., ±1 / (2 × Tr), which is ±1 / (2Loc × Tr).

[0183] Hereinafter, an example of the aliasing determination processing in the aliasing determination unit 212 will be described.

[0184] Here, as an example, the case where the coding multiplexing number N CM = 3, and the coding generation unit 104 uses three orthogonal codes Code1 = WH4(3) = [1, 1, -1, -1], Code2 = WH4(4) = [1, -1, -1, 1], and Code3 = WH4(2) = [1, -1, 1, -1] in the Walsh-Hadamard code with a code length Loc = 4 will be described.

[0185] The aliasing determination unit 212, for example, uses one ( = N allcode - N allcode ) of the N CM = 4 Walsh-Hadamard codes with a code length Loc = 4 as the unused orthogonal code for aliasing determination. For example, when the coding multiplexing number is N CM = 3, and the codes determined by the coding generation unit 104 for coding multiplexing transmission are Code1 = WH4(3) = [1, 1, -1, -1], Code2 = WH4(4) = [1, -1, -1, 1], and Code3 = WH4(2) = [1, -1, 1, -1], the unused orthogonal code is UnCode1 = WH4(1) = [1, 1, 1, 1].

[0186] For example, when the radar device 10 performs coding multiplexing transmission using orthogonal codes with a code length Loc = 4, as described above, the Doppler analysis unit 210 applies FFT processing to each coding element. Therefore, FFT processing is performed at a period of (Loc × Tr) = (4 × Tr), using the output from the beat frequency analysis unit 208. Accordingly, the Doppler range without aliasing according to the sampling theorem in the Doppler analysis unit 210 is ±1 / (2Loc × Tr) = ±1 / (8 × Tr).

[0187] Compared with the Doppler analysis range (Doppler range) in the Doppler analysis unit 210, the aliasing determination unit 212 performs aliasing determination within a range that is Loc times the code length of the orthogonal code sequence. For example, the aliasing determination unit 212 assumes a Doppler range of ±1 / (2×Tr), which is 4 (=Loc) times the Doppler range of ±1 / (8×Tr) where no aliasing occurs in the Doppler analysis unit 210, and performs aliasing determination processing.

[0188] Here, the Doppler component VFT b_cfar which is the output of the Doppler analysis unit 210 corresponding to the distance index f s_cfar extracted in the CFAR unit 211 and the Doppler frequency index f z noc (f b_cfar , f s_cfar ) may, for example, include Doppler components with aliasing as shown in (a) of Figure 3 and (b) of Figure 3 within the Doppler range of ±1 / (2×Tr).

[0189] For example, as shown in (a) of Figure 3 , when f s_cfar < 0, within the Doppler range of ±1 / (2×Tr), there may be 4 (=Loc) types of Doppler components: f s_cfar - Ncode, f s_cfar , f s_cfar + Ncode, and f s_cfar + 2Ncode.

[0190] In addition, for example, as shown in (b) of Figure 3 , when f s_cfar > 0, within the Doppler range of ±1 / (2×Tr), there may be 4 (=Loc) types of Doppler components: f s_cfar - 2Ncode, f s_cfar - Ncode, f s_cfar , and f s_cfar + Ncode.

[0191] The aliasing determination unit 212, for example, uses an unused orthogonal code to perform code separation processing within the Doppler range of ±1 / (2×Tr) as shown in Figure 3 . For example, the aliasing determination unit 212 can also correct the phase changes of the 4 (=Loc) types of Doppler components with aliasing as shown in Figure 3 for the unused orthogonal code.

[0192] Next, the aliasing determination unit 212 determines whether each Doppler component includes aliasing based on the received power of the Doppler components that have been encoded and separated according to the unused orthogonal code. For example, the aliasing determination unit 212 detects the Doppler component with the minimum received power among the Doppler components including aliasing, and determines the detected Doppler component as the true Doppler component. In other words, the aliasing determination unit 212 determines the Doppler components with received powers different from the minimum received power among the Doppler components including aliasing as false Doppler components.

[0193] Through this aliasing determination process, the ambiguity of the Doppler range including aliasing can be reduced. In addition, through this aliasing determination process, compared with the Doppler range of the Doppler analysis unit 210, the range in which the Doppler frequency can be detected unambiguously can be expanded to a range greater than -1 / (2Tr) and less than 1 / (2Tr).

[0194] Based on the encoding separation using the unused orthogonal code, for example, for the true Doppler component, the phase change of this Doppler component is correctly corrected, and the orthogonality between the orthogonal code for coded multiplexing transmission and the unused orthogonal code is maintained. As a result, the unused orthogonal code is not correlated with the coded multiplexing transmission signal, and the received power is at the noise level.

[0195] On the other hand, for example, for the false Doppler component, the phase change of this Doppler component is incorrectly corrected, and the orthogonality between the orthogonal code for coded multiplexing transmission and the unused orthogonal code cannot be maintained. As a result, a correlation component (interference component) between the unused orthogonal code and the coded multiplexing transmission signal is generated. For example, a received power greater than the noise level can be detected.

[0196] Therefore, as described above, the aliasing determination unit 212 can determine the Doppler component with the minimum received power among the Doppler components that have been encoded and separated based on the unused orthogonal code as the true Doppler component, and determine the other Doppler components with received powers different from the minimum received power as false Doppler components.

[0197] For example, the aliasing determination unit 212 corrects the phase change of the Doppler components including aliasing based on the output of the Doppler analysis unit 210 in each antenna system processing unit 201, and calculates the received power DeMulUnCode after encoding separation using the unused orthogonal code UnCode according to the following formula (12). nuc of nuc (f b_cfar ,f s_cfar ,DR).

[0198]

[0199] In Equation (12), for the outputs of the Doppler analysis unit 210 in all the antenna system processing units 201, the sum of the received powers after code separation using the unused orthogonal code UnCode is calculated. Thus, even when the received signal level is low, the aliasing determination accuracy can be improved. However, it is also possible that, instead of Equation (12), for the outputs of the Doppler analysis unit 210 in a part of the antenna system processing units 201, the received powers after code separation using the unused orthogonal code are calculated. Even in this case, for example, the aliasing determination accuracy can be maintained within a range where the received signal level is sufficiently high, and the amount of arithmetic processing can be reduced. nuc In addition, in Equation (12), nuc = 1, …, N

[0200] -N allcode . Further, DR is an index indicating the Doppler aliasing range. For example, it takes integer values in the range of DR = ceil[-Loc / 2], ceil[-Loc / 2]+1, …, 0, …, ceil[Loc / 2]-1. CM In addition, in Equation (12), the operator

[0201] represents the product of each element of vectors with equal number of elements. For example, for the n-dimensional vectors A = [a1,..,a and B = [b1,..,b n , the product of each element is represented by the following Equation (13). n In addition, in Equation (12), the operator “·” represents the vector inner product operator. In addition, in Equation (12), the superscript T represents vector transpose, and the superscript * (asterisk) represents the complex conjugate operator.

[0202]

[0203] In Equation (12), α(f

[0204] ) represents the “Doppler phase correction vector”. For the Doppler phase correction vector α(f s_cfar ), for example, in the case of setting it to the output range (in other words, the Doppler range) of the Doppler analysis unit 210 that does not include Doppler aliasing, the Doppler frequency index f s_cfar extracted in the CFAR unit 211 corrects the Doppler phase rotation amount caused by the time difference of Doppler analysis between Loc Doppler analysis units 210. s_cfar For example, the Doppler phase correction vector α(f

[0205] ) is represented in the following Equation (14). The Doppler phase correction vector α(f s_cfar ) shown in Equation (14) s_cfar)For example, it is a vector with Doppler phase correction coefficients as elements, and the Doppler phase correction coefficients correct the following phase rotation amount, which is the output VFT of the first Doppler analysis unit 210 z 1 (f b_cfar ,f s_cfar ) is based on the Doppler analysis time, and is the output VFT of the second Doppler analysis unit 210 z 2 (f b_cfar ,f s_cfar ) to the output of each of the Loc-th Doppler analysis unit VFT z Loc (f b_cfar ,f s_cfar ) The phase rotation amount in the Doppler component of the Doppler frequency index f generated by the time delays of Tr, 2Tr,..., (Loc - 1)Tr in each output s_cfar .

[0206]

[0207] In addition, in Equation (12), β(DR) represents the "aliasing phase correction vector". Considering the case of Doppler aliasing, the aliasing phase correction vector β(DR) corrects, for example, the Doppler phase rotation amount that is an integer multiple of 2π in the Doppler phase rotation caused by the time difference of Doppler analysis between the Loc Doppler analysis units 210

[0208] For example, the aliasing phase correction vector β(DR) is expressed in the following Equation (15).

[0209]

[0210] For example, when Loc = 4, taking integer values of DR = -2, -1, 0, 1, the aliasing phase correction vector β(DR) is expressed in Equations (16), (17), (18), and (19).

[0211] β(-2) = [1, -1, 1, -1] (16)

[0212]

[0213] β(0) = [1, 1, 1, 1] (18)

[0214]

[0215] For example, when Loc = 4 Figure 3 of (a) or Figure 3 in (b), the output of the detected Doppler analysis unit 210, that is, the Doppler frequency index fs_cfar The Doppler range of the Doppler component of DR = 0 (eg, -1 / 8 Tr to +1 / 8 Tr) corresponds to DR = 0. In addition, according to the Doppler frequency index f for DR = 0 s_cfar The Doppler phase rotation of an integer multiple of 2π (e.g., β(1), β(-1), and β(-2)) is performed to calculate the Doppler component corresponding to the Doppler range of DR=1 (e.g., 1 / 8Tr to 3 / 8Tr), the Doppler component corresponding to the Doppler range of DR=-1 (e.g., -3 / 8Tr to -1 / 8Tr), and the Doppler component corresponding to the Doppler range of DR=-2 (e.g., -1 / 2Tr to -3 / 8Tr and 3 / 8Tr to 1 / 2Tr).

[0216] In addition, in formula (12), VFTALL z (f b_cfar ,f s_cfar ) For example, the output VFT of the Loc Doppler analysis unit 210 in the z-th antenna system processing unit 201 is expressed in vector form as follows (20): z noc (f b ,f s ) and the distance index f extracted in the CFAR unit 211 b_cfar and the Doppler frequency index f s_cfar Corresponding component VFT z noc (f b_cfar ,f s_cfar )(where noc=1,…,Loc).

[0217] VFTALL z (f b_cfar ,f s_cfar )=[VFT z 1 (f b_cfar ,f s_cfar ),VFT z 2 (f b_cfar ,f s_cfar )...,VFT z Loc (f b_cfar ,f s_cfar )] (20)

[0219] For example, the aliasing determination unit 212 calculates, according to Equation (12), the received power DeMulUnCode after code separation using the unused orthogonal code UnCode that corrects the phase change of the Doppler component including aliasing within the range of DR = ceil[-Loc / 2], ceil[-Loc / 2]+1, …, 0, …, ceil[Loc / 2]-1 nuc where the received power DeMulUnCode nuc (f b_cfar ,f s_cfar ,DR).

[0220] Next, the aliasing determination unit 212 detects the DR with the minimum received power DeMulUnCode nuc (f b_cfar ,f s_cfar ,DR) in each DR range. Hereinafter, as shown in the following Equation (21), the DR with the minimum received power DeMulUnCode nuc (f b_cfar ,f s_cfar ,DR) in each DR range is described as "DR min ".

[0221]

[0222] Hereinafter, the reason for determining Doppler aliasing through the above-described aliasing determination process will be described.

[0223] For example, if the noise component is ignored, the radar transmission signal component transmitted from the ncm-th transmitting antenna 106 (e.g., Tx#ncm) included in VFTALL z (f b_cfar ,f s_cfar ) shown in Equation (20) is represented in the following Equation (22).

[0224]

[0225] Here, γz,ncm represents the complex reflection coefficient when the signal of the radar transmission signal transmitted from the ncm-th transmitting antenna 106 is received by the z-th antenna system processing unit 201 after being reflected by the target. In addition, DR true represents the following index, which represents the true Doppler aliasing range. DR true is set to the index value in the range of ceil[-Loc / 2], ceil[-Loc / 2]+1, …, 0, …, ceil[Loc / 2]-1. Hereinafter, it is shown that the determination can be made in such a way that DR min = DR ture .

[0226] For the first to NCM The radar transmission signal component transmitted by one transmission antenna 106 uses the unused orthogonal code UnCode nuc The sum PowDeMul(nuc, DR, DR true ) of the received powers after encoding separation using

[0227]

[0228] In addition, PowDeMul(nuc, DR, DR shown in Equation (23) true ) corresponds to the evaluation value of the term in Equation (12) .

[0229] In Equation (23), when DR = DRtrue, the unused orthogonal code UnCode nuc and the orthogonal code Code for coded multiplex transmission ncm The correlation value between them is zero (for example, UnCode nuc * ·{Code ncm} T = 0), so PowDeMul(nuc, DR, DR true ) = 0.

[0230] On the other hand, in Equation (23), when DR ≠ DRtrue, the output depends on The correlation value between the orthogonal code Code for coded multiplex transmission ncm PowDeMul(nuc, DR, DR true ). Here, if PowDeMul(nuc, DR, DR nuc ) is not zero in all UnCodes true , for example, as long as the following Equation (24) is satisfied, when DR = DR true , the power of PowDeMul(nuc, DR true , DR true ) is the smallest, and the aliasing determination unit 212 can detect DR true (= DR min ). In other words, the aliasing determination unit 212 can perform Doppler aliasing determination according to Equation (12).

[0231]

[0232] For example, in order to satisfy Equation (24), as long as The term does not match other unused orthogonal codes UnCode nuc2 . Here, nuc2 ≠ nuc.

[0233] Therefore, when the number of unused orthogonal codes is 1, Equation (24) is satisfied. Additionally, when there are multiple unused orthogonal codes, for example, the encoding generation unit 104 can also select the encoding for multiplexed transmission in such a way that the terms of do not match other unused orthogonal codes.

[0234] Here, when using encodings such as Walsh-Hadamard codes or orthogonal M-sequence codes, sometimes in the orthogonal codes of code length Loc, there are groups of codes where the odd-numbered coding elements between the encodings are the same, and the signs of the even-numbered coding elements are reversed.

[0235] On the other hand, since β(0) = [1, 1,..., 1], β(-Loc / 2) = [1, -1, 1, -1,..., 1, -1], so the terms of are converted into codes where the odd-numbered coding elements are the same, and the signs of the even-numbered coding elements are reversed. nuc

[0236] Therefore, when the number of unused orthogonal codes (N allcode -N CM ) is 2 or more, for example, the encoding generation unit 104 can also select the encoding for multiplexed transmission or the unused orthogonal codes in the following way, that is, the group of codes where one of the odd-numbered and even-numbered coding elements among the orthogonal codes of code length Loc is the same, and the signs of the other odd-numbered and even-numbered coding elements are reversed is not included in the unused orthogonal codes.

[0237] For example, the Walsh-Hadamard code with code length Loc = 4 includes WH4(1) = [1, 1, 1, 1] and WH4(2) = [1, -1, 1, -1], or, Therefore, for example, the encoding generation unit 104 can also select the encoding for multiplexed transmission or the unused orthogonal codes in such a way that the group of WH4(1) and WH4(2) is not included in the multiple unused orthogonal codes. Additionally, for WH4(3) = [1, 1, -1, -1] and WH4(4) = [1, -1, -1, 1], the relationship is the same. Therefore, for example, the encoding generation unit 104 can also select the encoding for multiplexed transmission or the unused orthogonal codes in such a way that the group of WH4(3) and WH4(4) is not included in the multiple unused orthogonal codes.

[0238] Furthermore, when there are multiple unused orthogonal codes UnCode nuc , instead of the received power DeMulUnCode nuc (fb_cfar , f s_cfar , DR), the received power DeMulUnCodeAll(f after coding separation using all unused orthogonal codes is calculated in the following manner of Equation (25) b_cfar , f s_cfar , DR).

[0239]

[0240] By calculating the received power after coding separation using all unused orthogonal codes, the aliasing determination unit 212 can improve the aliasing determination accuracy even when the received signal level is low.

[0241] For example, the aliasing determination unit 212 calculates DeMulUnCodeAll(f in each range of DR = ceil[-Loc / 2], ceil[-Loc / 2]+1, …, 0, …, ceil[Loc / 2]-1 b_cfar , f s_cfar , DR), detects the received power DeMulUnCodeAll(f b_cfar , f s_cfar , DR min ). In the case of using Equation (25), hereinafter, as shown in the following Equation (26), the DR that gives the minimum received power within the DR range is described as "DR min ".

[0242]

[0243] In addition, the aliasing determination unit 212 can also perform the following processing, for example, that is, when comparing the minimum received power DeMulUnCode nuc after coding separation using the unused orthogonal code UnCode nuc (f b_cfar , f s_cfar , DR min ) with the received power to determine (in other words, measure) the certainty of the aliasing determination. In this case, the aliasing determination unit 212 can also determine the certainty of the aliasing determination according to the following Equation (27) and Equation (28).

[0244] DeMulUnCode nuc (f b_cfar , f s_cfar , DR min ) < Threshold DR × PowerFT(f b_cfar , f s_cfar )(27)

[0245] DeMulUnCode nuc (f b_cfar ,f s_cfar ,DR min )≥Threshod DR ×PowerFT(f b_cfar ,f s_cfar )(28)

[0246] For example, when using the unused orthogonal code UnCode nuc The minimum receiving power after coding separation DeMulUnCode nuc (f b_cfar ,f s_cfar ,DR min ) is less than the distance index f extracted in the CFAR unit 211 b_cfar and the Doppler frequency index f s_cfar The received power value PowerFT(f b_cfar ,f s_cfar ) multiplied by the specified value Threshold DR When the obtained value is (for example, equation (27)), the aliasing determination unit 212 determines that the aliasing determination is sufficiently certain. In this case, the radar device 10 performs the subsequent processing (for example, code separation processing), for example.

[0247] On the other hand, for example, when using the unused orthogonal code UnCode nuc The minimum receiving power after coding separation DeMulUnCode nuc (f b_cfar ,f s_cfar ,DR min ) is equal to or greater than the received power value PowerFT(f b_cfar ,f s_cfar ) multiplied by Threshold DR If the obtained value is (e.g., equation (28)), the aliasing determination unit 212 determines that the accuracy of aliasing determination is insufficient (e.g., a noise component). In this case, the radar device 10 may not perform subsequent processing (e.g., code separation processing).

[0248] By this process, the aliasing determination error in the aliasing determination unit 212 can be reduced, and the noise component can be removed. DR For example, it can be set in the range of greater than 0 and less than 1. As an example, if noise components are included, the Threshold can also be set in the range of about 0.1 to 0.5. DR .

[0249] In addition, when there are multiple unused orthogonal codes UnCodenuc In the case of, the aliasing determination unit 212 may also perform the following processing, that is, instead of the received power DeMulUnCode nuc (f b_cfar , f s_cfar , DR), use DeMulUnCodeAll(f b_cfar , f s_cfar , DR) to compare with the received power, thereby determining (in other words, measuring) the certainty of the aliasing determination. In this case, the aliasing determination unit 212 may also use, for example, DeMulUnCodeAll(f b_cfar , f s_cfar , DR) instead of DeMulUnCode nuc (f b_cfar , f s_cfar , DR) in equations (27) and (28) to determine the certainty of the aliasing determination. By obtaining the received power after coding separation using all unused orthogonal codes, even in the case of a low received signal level, the aliasing determination unit 212 can improve the accuracy of the certainty of the aliasing determination.

[0250] In addition, instead of equation (12), the calculation formula for the received power DeMulUnCode nuc after coding separation using the unused orthogonal code UnCode nuc (f b_cfar , f s_cfar , DR) may also be, for example, the following equation (29).

[0251]

[0252] In equation (29), the term does not depend on the index of the Doppler component (Doppler frequency index) f s , so, for example, by tabulating it in advance, the amount of computation in the aliasing determination unit 212 can be reduced.

[0253] The above has described the operation example of the aliasing determination unit 212. [[ID=…]] [[ID=…]]

[0254] Next, the operation example of the code multiplexing and separation unit 213 will be described.

[0255] The code multiplexing and separation unit 213 performs the separation process of the code multiplexed signal based on the aliasing determination result in the aliasing determination unit 212 and the code for code multiplexing transmission.

[0256] For example, the code multiplexing and separation unit 213, in the following manner of equation (30), based on the aliasing phase correction vector β(DR min using the aliasing determination result in the aliasing determination unit 212, that is, DR min), for the distance index f extracted in the CFAR unit 211 b_cfar and the Doppler frequency index f s_cfar the output of the Doppler analysis unit 210 corresponding thereto, i.e., the Doppler component VFTALL z (f b_cfar , f s_cfar ) is subjected to coded separation processing. In the aliasing determination unit 212, it is possible to determine, within the Doppler range of -1 / (2Tr) or more and less than 1 / (2Tr), the index as the true Doppler aliasing range (in other words, it is possible to make a determination in such a way that DR min = DR true ), and thus, in the coded multiplexing separation unit 213, within the Doppler range of -1 / (2Tr) or more and less than 1 / (2Tr), the correlation value between the orthogonal codes for coded multiplexing is set to zero, thereby enabling separation processing that suppresses interference between the coded multiplexing signals.

[0257]

[0258] Here, DeMul z ncm (f b_cfar , f s_cfar ) is the output obtained by performing coded separation on the coded multiplexing signal using the orthogonal code Code b_cfar and the Doppler frequency index f s_cfar of the output of the Doppler analysis unit 210 in the z-th antenna system processing unit 201 (for example, the coded separation result). In addition, z = 1,..., Na, ncm = 1,..., N ncm CM .

[0259] In addition, the coded multiplexing separation unit 213 may use the following formula (31) instead of formula (30).

[0260]

[0261] In formula (31), the term (however, in formula (31), DR = DR min ) does not depend on the index of the Doppler component (for example, the Doppler frequency index) f s , and thus, for example, by tabulating it in advance, the amount of computation in the coded multiplexing separation unit 213 can be reduced.

[0262] ​Through the code separation process described above, the radar device 10 can obtain the orthogonal code Code assigned to the ncm-th transmitting antenna Tx#ncm based on the result of the aliasing determination performed by the aliasing determination unit 212 based on the Doppler range ±1 / (2Loc×Tr) of the Loc times the Doppler range where aliasing does not occur in the Doppler analysis unit 210. ncm The signal obtained by separating the signal sent by code multiplexing.

[0263] Furthermore, the radar device 10 performs Doppler phase correction including Doppler aliasing (for example, based on the aliasing phase correction vector β (DR )) on the output of the Doppler analysis unit 210 for each code element during the code separation process. min ) processing). Therefore, mutual interference between code-multiplexed signals can be reduced to, for example, the noise level. In other words, in the radar device 10, inter-symbol interference can be reduced, thereby suppressing the impact of degradation on the detection performance of the radar device 10.

[0264] Figure 4 Another configuration example of the radar device 10 is shown. Figure 1 In the configuration of the radar device 10 shown in FIG. , as shown in equations (12), (29), (30), and (31), The items are used in common by the aliasing determination unit 212 and the coding multiplexing separation unit 213. For example, Figure 4 The radar device 10a shown in the figure includes a phase correction unit 215, which can convert the Doppler component VFTALL z (f b_cfar ,f s_cfar ) multiplied by the Doppler phase correction vector α(f s_cfar ) The aliasing determination unit 212a and the code multiplexing separation unit 213a may output the aliasing determination unit 212a and the code multiplexing separation unit 213a. By performing calculations on the terms, repeated calculations of the above terms can be reduced in the radar device 10a.

[0265] The above describes an example of the operation of the code multiplexing separation unit 213.

[0266] exist Figure 1 In the example, the direction estimation unit 214 calculates the distance index f based on the distance index f input from the code multiplexing separation unit 213. b_cfar , Doppler frequency index f s_cfar The code separation result DeMul outputted from the corresponding Doppler analysis unit 210 z ncm (f b_cfar ,f s_cfar), perform direction estimation processing of the target.

[0267] For example, the direction estimation unit 214 generates the virtual receiving array correlation vector h(f shown in Equation (32) b_cfar , f s_cfar ), and performs direction estimation processing.

[0268] The virtual receiving array correlation vector h(f b_cfar , f s_cfar ) contains Nt×Na elements, which is the product of the number of transmitting antennas Nt and the number of receiving antennas Na. The virtual receiving array correlation vector h(f b_cfar , f s_cfar ) is used to perform direction estimation processing on the reflected wave signal from the target based on the phase difference between the receiving antennas 202. Here, z = 1,..., Na.

[0269]

[0270] The direction estimation unit 214, for example, changes the azimuth direction θ in the direction estimation evaluation function value P H (θ, f b_cfar , f s_cfar ) within a specified angle range to calculate the spatial distribution. The direction estimation unit 214 extracts a specified number of maximum peaks of the calculated spatial distribution in descending order, and outputs the azimuth direction of the maximum peaks as the direction-of-arrival estimation value (for example, positioning output).

[0271] In addition, the direction estimation evaluation function value P H (θ, f b_cfar , f s_cfar ) has various methods according to the direction-of-arrival estimation algorithm. For example, the estimation method using an array antenna disclosed in Non-Patent Document 3 can also be used.

[0272] For example, when Nt×Na virtual receiving arrays are arranged in a straight line at equal intervals d H , the beamforming method can be expressed as shown in Equation (33) and Equation (34) below. In addition, methods such as Capon and MUSIC can also be applied in the same way.

[0273] P H (θ u , f b_cfar , f s_cfar ) = |a H (θ u ) D cal h(f b_cfar , f s_cfar )| 2 (33)

[0274]

[0275] Here, the superscript H is the Hermitian transpose operator. Additionally, a(θ u ) represents the direction vector of the virtual receiving array for the wave arriving with respect to the azimuth direction θ u .

[0276] Additionally, the azimuth direction θ u is a vector that changes with an azimuth interval DStep within the azimuth range θmin to θmax where direction-of-arrival estimation is performed. For example, θ u can be set as follows.

[0277] θ u = θmin + uDStep, u = 0, …, NU

[0278] NU = floor[(θmax - θmin) / DStep]

[0279] Here, floor(x) is a function that returns the largest integer value not exceeding the real number x.

[0280] Furthermore, in Equation (33), D cal is a square matrix of size (Nt × Na), containing the array correction coefficients for correcting the phase and amplitude deviations between the transmitting antennas and between the receiving array antennas, and the coefficients for reducing the influence of the element - to - element coupling between the antennas. In the case where the coupling between the antennas of the virtual receiving array can be ignored, D cal becomes a diagonal matrix, and the diagonal components contain the array correction coefficients for correcting the phase and amplitude deviations between the transmitting array antennas and between the receiving array antennas.

[0281] The direction estimation unit 214 can, for example, also output the direction estimation result, and further output the distance information based on the distance index f b_cfar , the Doppler velocity information of the target based on the Doppler frequency index f b_cfar of the target, and the determination result DR min of the aliasing determination unit 212 as the positioning result.

[0282] The direction estimation unit 214 can, for example, also calculate the Doppler frequency index f s_cfar based on the Doppler frequency index f min and the determination result DR es_cfar of the aliasing determination unit 212 according to Equation (35). The Doppler frequency index f es_cfar corresponds, for example, to the Doppler index when the FFT size of the Doppler analysis unit 210 is extended to Loc × Ncode. Hereinafter, f es_cfar is referred to as the "extended Doppler frequency index".

[0283] f es_cfar = f s_cfar + DR min × Ncode(35)

[0284] In addition, consider a Doppler range up to ±1 / (2×Tr), and the extended Doppler frequency index f es_cfar corresponding to this Doppler range has a range of -Loc×Ncode / 2 ≤ f es_cfar < Loc×Ncode / 2. Therefore, when the calculation result of Equation (35) is f es_cfar < -Loc×Ncode / 2, set f es_cfar + Loc×Ncode as f es_cfar . In addition, when f es_cfar ≥ Loc×Ncode / 2, set f es_cfar - Loc×Ncode as f es_cfar .

[0285] In addition, the Doppler frequency information can also be converted into a relative velocity component and output. When converting the Doppler frequency index f es_cfar into the relative velocity component v d (f es_cfar ), the conversion can also be performed using the following Equation (36). Here, λ is the wavelength of the carrier frequency of the RF (Radio Frequency) signal output from the wireless transmission unit (not shown). In addition, Δ f is the Doppler frequency interval in the FFT processing of the Doppler analysis unit 210. For example, in this embodiment, Δ f = 1 / {Loc×N code × T r}.

[0286]

[0287] Figure 5 (A), Figure 5 (B), Figure 5 (C), Figure 5 (D), and Figure 6 represent an example of the computer simulation result of the aliasing determination process.

[0288] Figure 5 (A), Figure 5 (B), Figure 5 (C), and Figure 5 As an example, the code multiplexing number is N CM= 3, the code length is Loc = 4, and the codes determined by the encoding generation unit 104 for encoding multiplexing transmission are Code1 = WH4(3) = [1, 1, -1, -1], Code2 = WH4(4) = [1, -1, -1, 1], and Code3 = WH4(2) = [1, -1, 1, -1], and the unused orthogonal code is UnCode1 = WH4(1) = [1, 1, 1, 1].

[0289] Figure 5 of (A), Figure 5 of (B), Figure 5 of (C) and Figure 5 of (D) indicate that, for example, in the aliasing determination unit 212, for the distance index f extracted in the CFAR211 unit b and the Doppler frequency index f s_cfar of the target, the result of calculating the received power DeMulUnCode1(f b , f s_cfar , DR) after encoding separation using the unused orthogonal code UnCode1 is calculated.

[0290] In addition, Figure 5 of (A) is an example of the calculation result of DeMulUnCode1(f b , f s_cfar , DR) when DR = -2 is plotted, Figure 5 of (B) is an example of the calculation result of DeMulUnCode1(f b , f s_cfar , DR) when DR = -1 is plotted, Figure 5 of (C) is an example of the calculation result of DeMulUnCode1(f b , f s_cfar , DR) when DR = 0 is plotted, Figure 5 of (D) is an example of the calculation result of DeMulUnCode1(f b , f s_cfar , DR) when DR = 1 is plotted.

[0291] In addition, in Figure 5 of (A), Figure 5 of (B), Figure 5 of (C) and Figure 5 of (D), the Doppler frequency ftarget of the target can be changed within the range of -fr / 2 or more and less than fr / 2, and each DeMulUnCode1(f b , f s , DR) is calculated.

[0292] Figure 5 of (A),Figure 5 of (B), Figure 5 of (C), and Figure 5 of (D), with the Doppler frequency f_target of the target on the horizontal axis, and DeMulUnCode1(f b , f s_cfar , DR) plotted on the vertical axis in the case where the Doppler frequency f_target of the target shown on the horizontal axis is plotted. In addition, for example, fr = 1 / Tr is set.

[0293] Figure 6 For example, it is as follows, that is, according to Figure 5 of (A), Figure 5 of (B), Figure 5 of (C), and Figure 5 of (D), the received power DeMulUnCode1(f b , f s_cfar , DR) when the Doppler frequency f_target of the target shown on the horizontal axis is changed within the range of -fr / 2 or more and less than fr / 2, and the DR with the minimum value, that is, DR min is plotted on the vertical axis as an example.

[0294] According to Figure 5 of (A), Figure 5 of (B), Figure 5 of (C), Figure 5 of (D), and Figure 6 , when the Doppler frequency range of the target is -fr / 8 ≤ f_target < fr / 8, DR min = 0, when the Doppler frequency range of the target is fr / 8 ≤ f_target < 3fr / 8, DR min = 1, when the Doppler frequency range of the target is -3fr / 8 ≤ f_target < -fr / 8, DR min = -1, when the Doppler frequency range of the target is -fr / 2 ≤ f_target < -3fr / 8 and 3r / 8 ≤ f_target < fr / 2, DR min = -2. This is consistent with, for example, the description using (a) of the above Figure 3 or (b) of Figure 3 . According to the result of DR Figure 6 shown in min , it can be seen that the Doppler range corresponding to the Doppler frequency range f_target of the target can be accurately detected.

[0295] In addition, Figure 7 is a diagram showing the following example, which is based on the result of Figure 6 , and plots, in the direction estimation unit 214, based on the Doppler frequency index fs_char and the determination result DR in the aliasing determination unit 212 min , calculate the extended Doppler frequency index f according to Equation (35) es_cfar and the obtained result.

[0296] Figure 7 Take the following example. When changing the Doppler frequency f_target of the target shown on the horizontal axis within the range greater than -fr / 2 and less than fr / 2, the calculated extended Doppler frequency index f es_cfar is converted into the Doppler frequency f_target_est and plotted on the vertical axis.

[0297] In addition, when converting the extended Doppler frequency index f es_cfar into the Doppler frequency f_target_est, the Doppler frequency interval Δ of the FFT processing in the Doppler analysis unit 210 can be used f for conversion. That is, f_target_est = f es_cfar ×Δ f can be used.

[0298] According to Figure 7 the result, it can be confirmed that by using the extended Doppler frequency index f es_cfar to detect the Doppler frequency f_target_est, within the range where the Doppler frequency f_target of the target is greater than -fr / 2 and less than fr / 2, that is, greater than -1 / (2Tr) and less than 1 / (2Tr), accurate detection can be performed.

[0299] As described above, in the present embodiment, the radar device 10 receives, in the receiving antenna 202, the reflected wave signal reflected by the target of the radar transmission signal that is encoded and multiplexed for transmission based on a part of the coding sequences in a plurality of orthogonal code sequences. In the aliasing determination unit 212, other orthogonal code sequences different from the part of the orthogonal code sequences used for encoding and multiplexing transmission among the plurality of orthogonal code sequences are used to determine the aliasing in the Doppler frequency domain of the reflected wave signal. For example, in the MIMO radar using encoded multiplexing transmission, the radar device 10 uses an orthogonal code sequence with a code length that can generate more orthogonal codes than the number of encoded multiplexing transmissions for encoding and multiplexing transmission.

[0300] With this configuration, for example, the radar device 10 determines Doppler aliasing for a received signal (e.g., the output of the Doppler analysis unit 210 for each coding element of the coded multiplexed signal) using an orthogonal code that is not used for coded multiplexed transmission. For example, compared to the Doppler analysis range in the Doppler analysis unit 210, the radar device 10 can determine aliasing within a Doppler range that is a multiple of the code length of the orthogonal code sequence. Thus, according to this embodiment, the radar device 10 can expand the Doppler range that can be detected unambiguously to the same Doppler range as in single-antenna transmission.

[0301] In addition, for example, when the radar device 10 performs code separation based on the determination result of Doppler aliasing, by performing Doppler phase correction including aliasing, the mutual interference between the coded multiplexed signals can be suppressed to the level of noise, and thus, the deterioration of radar detection performance can be suppressed, and coded multiplexed transmission of the MIMO radar can be performed.

[0302] Thus, according to this embodiment, for example, even when there is a Doppler variation caused by the movement of the target or the radar device 10, the generation of mutual interference between the signals multiplexed by coding is suppressed, and the Doppler range that can be detected unambiguously is set to the same Doppler range as in single-antenna transmission. Thus, in a wider Doppler frequency range, the detection accuracy of the target can be improved.

[0303] (Embodiment 2)

[0304] In Embodiment 1, the following case was described (e.g., refer to Figure 2 ), in the radar device 10, when transmitting a chirp pulse as a radar transmission signal Nc times, the center frequency of the chirp signal is kept constant. However, it is not limited to the case where the center frequency of the chirp signal is constant.

[0305] In this embodiment, the case where the center frequency of the chirp signal is variably set is described.

[0306] [Structure of the radar device]

[0307] Figure 8 It is a block diagram showing a structural example of the radar device 10b of this embodiment. In addition, in Figure 8 , the same reference numerals are attached to the same structures as in Embodiment 1 ( Figure 1 ) and their description is omitted.

[0308] Hereinafter, for example, the radar device 10b transmits a radar transmission signal in which the center frequency fc of the chirp signal changes by Δf in each transmission period Tr (e.g., increases when Δf > 0 and decreases when Δf <).

[0309] The radar transmission signal generation unit 101b in the radar transmission unit 100b includes a modulation signal generation unit 102, a VCO 103b, and a transmission frequency control unit 107.

[0310] For example, the modulation signal generation unit 102 periodically generates a modulation signal for controlling the VCO in a sawtooth shape. Here, the transmission period is set to Tr.

[0311] The transmission frequency control unit 107 controls the center frequency fc of the frequency modulation signal (chirp signal) output from the VCO 103b at the transmission period Tr. For example, the transmission frequency control unit 107 can cause the center frequency fc of the frequency modulation signal to change by Δf at the transmission period Tr.

[0312] The VCO 103b outputs a frequency modulation signal to the phase rotation unit 105 and the radar reception unit 200 (for example, the mixer unit 204) based on the outputs of the transmission frequency control unit 107 and the modulation signal generation unit 102.

[0313] Figure 9 Shows an example of the frequency modulation signal (hereinafter, chirp signal).

[0314] In Figure 9 , for example, the VCO 103b outputs a chirp signal with a center frequency fc(1) of f0 in the first transmission period Tr#1. In addition, as Figure 9 shows, the VCO 103b outputs a chirp signal with a center frequency fc(2) of f0 + Δf in the second transmission period Tr#2. Similarly, in Figure 9 , the VCO 103b outputs a chirp signal with a center frequency fc(m) of f0 + (m - 1)Δf in the m-th transmission period Tr#m. In this way, the VCO 103b changes the center frequency of the chirp signal by Δf at the transmission period Tr.

[0315] That is, in Figure 9 , the center frequency fc(N c ) of the chirp signal in the N-th c transmission period Tr#N c is f0 + Δf × (N c - 1).

[0316] In addition, each chirp signal can use, for example, a chirp signal with the same frequency modulation bandwidth Bw within the time range T A of the range gate. In addition, in the example shown in Figure 9 , the case where Δf > 0 (in other words, the case where the center frequency fc increases) is shown, and the case where Δf < 0 (in other words, the case where the center frequency fc decreases) is the same.

[0317] Figure 8 Other operations in the radar transmission unit 100b shown can also be the same as those in the first embodiment.

[0318] Next, an operation example in the radar reception unit 200b of the radar device 10b will be described.

[0319] In the radar reception unit 200b, the processing of the signal received by the reception antenna 202 by each antenna system processing unit 201 and the operations in the subsequent CFAR unit 211, aliasing determination unit 212, and coding multiplexing separation unit 213 are the same as those in the first embodiment. In addition, in the radar reception unit 200b, the direction estimation processing using the output of the coding multiplexing separation unit 213 in the direction estimation unit 214b is also the same as the operation in the first embodiment.

[0320] In the radar reception unit 200b, for example, the conversion processing of the Doppler velocity information regarding the target in the direction estimation unit 214b is different from that in the first embodiment.

[0321] In addition, the distance information R(f b ) is the same as that in the first embodiment. The radar reception unit 200b can, for example, based on Equation (8), use the beat frequency index (or distance index) f b to output the distance information R(f b ).

[0322] The direction estimation unit 214b can also, for example, based on the Doppler frequency index f s_cfar and the determination result of the aliasing determination unit 212, i.e., DR min , calculate the Doppler frequency index f es_cfar according to Equation (37). The Doppler frequency index f es_cfar corresponds to, for example, the Doppler index when the FFT size of the Doppler analysis unit 210 is extended to Loc × Ncode. Hereinafter, f es_cfar will be referred to as the "extended Doppler frequency index".

[0323] f es_cfar = f s_cfar + DR min × Ncode (37)

[0324] In addition, assuming a Doppler range up to ±1 / (2 × Tr), the range of the extended Doppler frequency index f es_cfar corresponding to this Doppler range is -Loc × Ncode / 2 ≤ f es_cfar < Loc × Ncode / 2. Therefore, when the calculation result of Equation (37) is f es_cfar < -Loc × Ncode / 2, f es_cfar + Loc × Ncode is set as f es_cfarIn addition, when f es_cfar ≧ Loc × Ncode / 2, set f es_cfar - Loc × Ncode as f es_cfar .

[0325] In addition, the direction estimation unit 214b can output the Doppler velocity information v es_cfar of the detected target using, for example, the extended Doppler frequency index f b_cfar and the distance index f d in the following manner.

[0326] For example, in the case of using a radar transmission signal in which the center frequency fc of the chirp signal changes by Δf in each transmission period Tr, even when the relative velocity of the target is zero, the center frequency fc of the chirp signal changes in each transmission period Tr. Therefore, the received signal of the radar device 10b includes a phase rotation caused by the change in the center frequency of the chirp signal in each transmission period Tr.

[0327] For the center frequency fc in the m-th transmission period Tr of the target distance R target , a change of (m - 1)Δf occurs with respect to the first center frequency. Considering the arrival time (2R target / Co) of the reflected wave from the target distance R target , the accompanying phase rotation amount Δη(m, R target ) is represented by Equation (38-1). In addition, the following Equation (38-1) represents the relative phase rotation amount with respect to the phase in the first transmission period Tr. C0 represents the speed of light. Therefore, the output of each of the Loc Doppler analysis units 210 of the radar device 10b includes a phase rotation caused by the change in the center frequency of the chirp signal in each transmission period Tr.

[0328]

[0329] Thus, as shown in Equation (38-2), the direction estimation unit 214b calculates the Doppler velocity information v d (f es_cfar , f b_cfar ) based on a conversion formula considering the change amount Δf of the center frequency fc of the chirp signal in each transmission period Tr.

[0330] The first term in Equation (38-2) corresponds to Equation (36) and is the extended Doppler frequency index f es_cfarThe relative Doppler velocity component represented. The second term in Equation (38-2) is the Doppler velocity component generated by changing the center frequency fc of the chirp signal by Δf according to the transmission period Tr. The direction estimation unit 214b can calculate the relative Doppler velocity v of the original target, for example, as shown in Equation (38-2), by removing the Doppler component of the second term from the first term. d (f es_cfar ,f b_cfar ). Here, R(f b_cfar ) is the distance information R(f b_cfar ) using the beat frequency index f b_cfar according to Equation (8).

[0331]

[0332] In addition, since the Doppler range up to ±1 / (2×Tr) is assumed, when v d is v d <-C0 / (4f0Tr), the direction estimation unit 214b can output the Doppler velocity information v d of the detected target according to the following Equation (39).

[0333]

[0334] Similarly, since the Doppler range up to ±1 / (2×Tr) is assumed, when v d is v d >C0 / (4f0Tr), the direction estimation unit 214b can output the Doppler velocity information v d of the detected target according to the following Equation (40).

[0335]

[0336] As described above, in the present embodiment, in the radar device 10b, the center frequency fc of the chirp signal changes based on the transmission period Tr of the radar transmission signal. For example, the radar device 10b changes the center frequency fc of the chirp signal by Δf according to the transmission period Tr (for example, increases when Δf>0 and decreases when Δf<), and transmits this chirp signal as the radar transmission signal. Even in this case, the radar device 10b (for example, a MIMO radar) can apply coded multiplexing transmission. In addition, the radar device 10b can determine Doppler aliasing using the output (in other words, the received signal) of the Doppler analysis unit 210 for each coding element of the coded multiplexing signal and without using orthogonal codes, in the same manner as in Embodiment 1.

[0337] In addition, similar to Embodiment 1, the radar device 10b can set the Doppler frequency range that can be detected unambiguously to ±1 / (Tr) by performing Doppler phase correction including aliasing during code separation, and suppress the mutual interference between the code multiplexed signals to approximately the noise level. Thus, according to this embodiment, the degradation of the radar detection performance can be suppressed, and code multiplexed transmission of the MIMO radar can be performed.

[0338] In addition, in this embodiment, the radar device 10b transmits, for example, a chirp signal in which the center frequency fc of the chirp signal changes by Δf in accordance with the transmission period Tr. Therefore, according to the change amplitude of the center frequency of the chirp signal, the range resolution can be improved (for example, refer to Non-Patent Document 4). According to this embodiment, since the range resolution can be improved according to the change amplitude of the center frequency of the chirp signal, the chirp sweep frequency band (for example, Bw) can be reduced as compared with the case where the center frequency of the chirp signal is transmitted constantly. By reducing the chirp sweep frequency band, for example, the range resolution can be improved and the transmission period Tr can be shortened, so that the Doppler range that can be detected unambiguously can be further expanded during code multiplexed transmission.

[0339] (Modification 1 of Embodiment 2)

[0340] The period for changing the center frequency of the chirp signal is not limited to the transmission period Tr. In Modification 1, the case where the center frequency of the chirp signal is set to be variable in each transmission period (Loc×Tr) (hereinafter, referred to as “code transmission period”) of the code length Loc of one orthogonal code used for code multiplexed transmission will be described.

[0341] [Structure of Radar Device]

[0342] Figure 10 is a block diagram showing a structural example of the radar device 10c of Modification 1. In addition, in Figure 10 , the same reference numerals are attached to the same structures as those in Embodiment 1 ( Figure 1 ) or Embodiment 2 ( Figure 8 ), and their descriptions are omitted.

[0343] In Modification 1, for example, the radar device 10c transmits a radar transmission signal in which the center frequency fc of the chirp signal changes by Δf in accordance with the code transmission period (Loc×Tr) (for example, increases when Δf>0 and decreases when Δf<).

[0344] The radar transmission signal generation unit 101c in the radar transmission unit 100c includes a modulation signal generation unit 102, a VCO 103c, and a transmission frequency control unit 107c.

[0345] For example, the modulation signal generation unit 102 periodically generates a sawtooth-shaped modulation signal for VCO control. Here, the transmission period is set to Tr.

[0346] The transmission frequency control unit 107c controls the center frequency fc of the frequency modulation signal (chirp signal) output from the VCO 103c at the coded transmission period (Loc × Tr) based on the orthogonal code element index OC_INDEX output from the coding generation unit 104c.

[0347] For example, in the transmission period Tr when OC_INDEX = 1, the transmission frequency control unit 107c can change the center frequency fc of the frequency modulation signal output from the VCO 103c by Δf at the transmission period Tr. In other words, in the transmission period Tr when OC_INDEX ≠ 1, the transmission frequency control unit 107c controls the center frequency fc of the frequency modulation signal output from the VCO 103c to be the same as the center frequency fc in the previous transmission period Tr. Through this control, the transmission frequency control unit 107c can control the center frequency fc to change by Δf at the coded transmission period (Loc × Tr).

[0348] The VCO 103c outputs a frequency modulation signal to the phase rotation unit 105 and the radar receiving unit 200 (for example, the mixer unit 204) based on the outputs of the transmission frequency control unit 107c and the modulation signal generation unit 102.

[0349] Figure 11 Shows an example of the frequency modulation signal (hereinafter, chirp signal) after frequency modulation.

[0350] In Figure 11 , for example, in the first transmission period Tr#1 (for example, OC_INDEX = 1), the VCO 103c outputs a chirp signal with a center frequency fc(1) of f0. In addition, as Figure 11 shown, in the second transmission period Tr#2 (for example, OC_INDEX = 2), the VCO 103c outputs a chirp signal with a center frequency fc(2) of f0. Similarly, in the third transmission period (for example, OC_INDEX = 3. Not shown) to the Loc-th transmission period Tr#Loc (for example, OC_INDEX = Loc), the VCO 103c outputs chirp signals with center frequencies fc(3) to fc(Loc) of f0, respectively.

[0351] In the (Loc + 1)-th transmission period Tr#(Loc + 1), VCO103c outputs a chirp signal with a center frequency fc(Loc + 1) of f0 + Δf. Additionally, in the (Loc + 2)-th transmission period Tr#(Loc + 2) to the (2Loc)-th transmission period Tr#(2Loc), VCO103c outputs chirp signals with center frequencies fc(Loc + 2) to fc(2Loc) of f0 + Δf respectively.

[0352] Similarly, in the m-th transmission period Tr#m, VCO103c outputs a chirp signal with a center frequency fc(m) of f0 + floor[(m - 1) / Loc]Δf.

[0353] That is, in Figure 11 the c N-th c transmission period Tr#N c the center frequency fc(N c ) of the chirp signal is f0 + (Ncode - 1)Δf. Here, Ncode = N

[0354] / Loc. A In addition, each chirp signal can use, for example, a chirp signal with the same frequency modulation bandwidth Bw within the time range T Figure 11 of the range gate. Also, in the example shown in

[0355] Figure 10 Δf > 0 (in other words, the case where the center frequency fc increases) is shown, and the same applies to the case where Δf < 0 (in other words, the case where the center frequency fc decreases).

[0356] Next, an example of the operation in the radar receiving unit 200c of the radar device 10c will be described.

[0357] In the radar receiving unit 200c, the processing of the signal received by the receiving antenna 202 by each antenna system processing unit 201 and the subsequent operations in the CFAR unit 211 are the same as those in the first embodiment. Additionally, in the radar receiving unit 200c, the direction estimation processing using the output of the code multiplexing separation unit 213 in the direction estimation unit 214c is also the same as the operation in the first embodiment.

[0358] In the radar receiving unit 200c, for example, the operations of the aliasing determination unit 212c, the code multiplexing separation unit 213c, and the conversion processing of the Doppler velocity information of the target in the direction estimation unit 214c are different from those in the first embodiment.

[0359] Hereinafter, an example of the operation different from that in the first embodiment in the aliasing determination unit 212c will be described.

[0360] For example, in the case of using a radar transmission signal in which the center frequency fc of a chirp signal changes by Δf according to a coded transmission period (Loc × Tr), even when the relative speed of the target is zero, the center frequency fc of the chirp signal changes according to the coded transmission period (Loc × Tr). Therefore, the output of each of the Loc Doppler analysis units 210 of the radar device 10c includes a phase rotation caused by the change in the center frequency of the chirp signal for each coded transmission period (Loc × Tr).

[0361] That is, for the target distance R target the center frequency fc in the m-th transmission period Tr changes by floor[(m - 1) / Loc]Δf with respect to the center frequency fc in the first transmission period Tr. Therefore, considering the arrival time of the reflected wave from the target distance R target (2R target / Co), the phase rotation amount Δη(m, R target ) caused by the change in the center frequency is represented by Equation (40-1). In addition, Equation (40-1) represents the relative phase rotation amount with respect to the phase of the first transmission period Tr. C0 represents the speed of light.

[0362]

[0363] Since the switching period of the Doppler analysis unit 210 for each coding element is made to coincide with the coded transmission period (Loc × Tr) in which the center frequency fc of the chirp signal changes by Δf, each of the Loc Doppler analysis units 210 performs Doppler analysis including the phase rotation shown in Equation (40-1).

[0364] Therefore, it is different in the following respect, that is, when the aliasing determination unit 212c corrects the Doppler phase rotation caused by the time difference of the Doppler analysis between the Loc Doppler analysis units 210, in addition to the Doppler phase correction vector α(f s_cfar ), the center frequency change correction vector ξ(f b_cfar ) shown in Equation (40-2) is also used to correct the phase. That is, the aliasing determination unit 212c uses instead of α(f s_cfar ). In addition, R(f b_cfar ) is the distance information R(f b_cfar ) using the beat frequency index f according to Equation (8) b_cfar .

[0365]

[0366] In Equation (40-2), according to the information from R(f​b_cfar ) arrival time of the reflected wave (2R(f b_cfar ), the change in Δf of ( / Co), the phase rotation amount becomes 2πΔf×(2R(f b_cfar ) / Co) within the coding transmission period (Loc×Tr). Therefore, for the phase rotation caused by the time difference of Doppler analysis between each of the Loc Doppler analysis units 210, with the first Doppler analysis unit 210 as the reference, it is derived as (noc - 1) / Loc times for each of the noc-th Doppler analysis units 210. In addition, noc = 1, …, Loc.

[0367] By setting the period in which the center frequency fc of the chirp signal changes by Δf as the coding transmission period (Loc×Tr), the switching period of the Doppler analysis unit 210 for each coding element becomes consistent. Therefore, the aliasing determination unit 212c can easily perform phase correction in the separation process of the coded multiplexed signal using unused codes (in addition to the Doppler phase correction vector α(f s_cfar ), the center frequency change correction vector of Equation (40 - 2) is also used).

[0368] For the above reasons, the aliasing determination unit 212c can use Equation (41) instead of Equation (12) to calculate the received power DeMulUnCode after coding separation using the unused orthogonal code UnCode nuc nuc (f b_cfar , f s_cfar , DR). The difference between Equation (41) and Equation (12) is that Equation (41) uses to replace α(f s_cfar ) in Equation (12). Here, nuc = 1, …, N allcode -N CM . In addition, DR is an index indicating the Doppler aliasing range, taking integer values in the range of DR = ceil[-Loc / 2], ceil[-Loc / 2]+1, …, 0, …, ceil[Loc / 2]-1.

[0369]

[0370] In addition, the aliasing determination unit 212c can also use Equation (42) instead of Equation (29).

[0371]

[0372] ​Next, an operation example different from that in the first embodiment in the encoding multiplexing separation unit 213c will be described. In the encoding multiplexing separation unit 213c, for the same reason as the description of the operation example of the above aliasing determination unit 212c, Equation (43) is used instead of Equation (30), and the aliasing determination result DR in the aliasing determination unit 212c is used min , for the distance index f extracted in the CFAR unit 211 b_cfar and the Doppler frequency index f s_cfar , the output of the Doppler analysis unit 210 corresponding thereto, i.e., the Doppler component VFTALL z (f b_cfar , f s_cfar ) is subjected to encoding separation processing. The difference between Equation (43) and Equation (30) is that is used instead of α(f s_cfar ) in Equation (30).

[0373]

[0374] In addition, the encoding multiplexing separation unit 213c can also use Equation (44) instead of Equation (31), and use the aliasing determination result DR of the aliasing determination unit 212c min , for the distance index f extracted in the CFAR unit 211 b_cfar and the Doppler frequency index f s_cfar , the output of the Doppler analysis unit 210 corresponding thereto, i.e., the Doppler component VFTALL z (f b_cfar , f s_cfar ) is subjected to encoding multiplexing signal separation processing. In Equation (44), does not depend on the index f of the Doppler component s , so by tabulating in advance, the amount of calculation can be reduced.

[0375]

[0376] In this way, by making the period of the change of the center frequency fc of the chirp signal by Δf coincide with the encoding transmission period (Loc × Tr), the switching period of the Doppler analysis unit 210 for each encoding element can be made consistent, and the phase correction in the encoding multiplexing separation processing can be easily performed.

[0377] Next, an operation example different from that in the first embodiment in the direction estimation unit 214c will be described.

[0378] The direction estimation unit 214c can also, for example, calculate the Doppler frequency index f according to Equation (45) based on the Doppler frequency index f s_cfar and the determination result DR of the aliasing determination unit 212c min , es_cfar。Doppler frequency index f es_cfar For example, it is the Doppler index when the FFT size of the Doppler analysis unit 210 is extended to Loc×Ncode. Hereinafter, f es_cfar is referred to as the "extended Doppler frequency index".

[0379] f es_cfar = f s_cfar + DR min × Ncode (45)

[0380] In addition, assuming a Doppler range up to ±1 / (2×Tr), the range of the extended Doppler frequency index f es_cfar corresponding to this Doppler range is -Loc×Ncode / 2 ≤ f es_cfar < Loc×Ncode / 2. Therefore, when the calculation result of Equation (45) is f es_cfar < -Loc×Ncode / 2, set f es_cfar + Loc×Ncode as f es_cfar . In addition, when f es_cfar ≥ Loc×Ncode / 2, set f es_cfar - Loc×Ncode as f es_cfar .

[0381] For a radar transmission signal in which the center frequency fc of the chirp signal changes by Δf according to the coding transmission period (Loc×Tr), even when the relative velocity of the target is zero, since the center frequency fc of the chirp signal changes according to the coding transmission period (Loc×Tr), each coding transmission period (Loc×Tr) also includes a phase rotation caused by the change in the center frequency of the chirp signal.

[0382] For the target distance R target the center frequency fc in the m-th transmission period Tr changes by floor[(m - 1) / Loc]Δf. Therefore, considering the arrival time of the reflected wave from the target distance R target (2R target / Co), the phase rotation amount Δη(m, R target ) caused by the change in the center frequency is represented by Equation (46). In addition, Equation (46) represents the relative phase rotation amount with respect to the phase of the first transmission period Tr. C0 represents the speed of light.

[0383]

[0384] Therefore, the direction estimation unit 214c can use, for example, the extended Doppler frequency index f es_cfar and the distance index f b_cfar, according to Equation (47), the Doppler velocity information v of the detected target is output d (f es_cfar , f b_cfar ). The first term in Equation (47) is equivalent to Equation (36) and represents the relative Doppler velocity component indicated by the Doppler frequency index f es_cfar . In addition, the second term in Equation (47) is the Doppler velocity component generated by changing the center frequency fc of the chirp signal by Δf according to the coded transmission period (Loc×Tr).

[0385] The direction estimation unit 214c can calculate the relative Doppler velocity v of the original target by removing the Doppler component of the second term from the first term in Equation (47) d (f es_cfar , f b_cfar ). Here, R(f b_cfar ) is the distance information R(f b_cfar ) using the beat frequency index f according to Equation (8) b_cfar ).

[0386]

[0387] As shown in Equation (47), the direction estimation unit 214c calculates the Doppler velocity information v based on a conversion formula considering the change amount Δf of the center frequency fc of the chirp signal for each coded transmission period (Loc×Tr) d .

[0388] In addition, since a Doppler range up to ±1 / (2×Tr) is assumed, when v d is v d < -C0 / (4f0Tr), the direction estimation unit 214c can output the detected Doppler velocity information v of the target according to the following Equation (48) d .

[0389]

[0390] Similarly, since a Doppler range up to ±1 / (2×Tr) is assumed, when v d is v d > C0 / (4f0Tr), the direction estimation unit 214c can output the detected Doppler velocity information v of the target according to the following Equation (49) d .

[0391]

[0392] As described above, in Modification 1, in the radar device 10c, the center frequency fc of the chirp signal varies based on the transmission period (Loc × Tr) of an orthogonal code sequence. For example, the radar device 10c changes the center frequency fc of the chirp signal by Δf according to the coded transmission period (Loc × Tr) (for example, increasing when Δf > 0 and decreasing when Δf < 0), and transmits this chirp signal as a radar transmission signal. Even in this case, the radar device 10c (e.g., MIMO radar) can apply coded multiplexing transmission. In addition, similar to Embodiment 1, the radar device 10c can determine Doppler aliasing using the output (in other words, the received signal) of the Doppler analysis unit 210 for each coding element of the coded multiplexing signal and the unused orthogonal code.

[0393] In addition, according to Modification 1, similar to Embodiment 1, the radar device 10c can set the Doppler frequency range that can be detected unambiguously to ±1 / (Tr) and suppress the mutual interference between the coded multiplexing signals to approximately the noise level by performing Doppler phase correction including aliasing during coded separation. Thus, according to Modification 1, it is possible to suppress the degradation of radar detection performance and perform coded multiplexing transmission of the MIMO radar.

[0394] In addition, according to Modification 1, when the period of changing the center frequency fc of the chirp signal by Δf is multiple transmission periods Tr, by making the coded transmission period (Loc × Tr) consistent and also consistent with the switching period of the Doppler analysis unit 210 for each coding element, it is possible to easily perform the separation process of the coded multiplexing signal using the unused code in the aliasing determination unit 212c and the phase correction in the coded multiplexing separation process of the coded multiplexing separation unit 213c.

[0395] In addition, in Modification 1, for example, the radar device 10c changes the center frequency fc of the chirp signal by Δf according to the coded transmission period (Loc × Tr) in the radar transmission signal and then transmits it. Therefore, the center frequency change amplitude of the chirp signal is Δf × Ncode, and the range resolution is 0.5C0 / (Δf × Ncode).

[0396] Thus, by increasing Δf × Ncode, according to Modification 1, since the range resolution can be improved based on the change amplitude of the center frequency of the chirp signal, the chirp sweep frequency band (e.g., Bw) can be reduced compared to the case of transmitting with a constant center frequency of the chirp signal. By reducing the chirp sweep frequency band, for example, the range resolution can be improved and the transmission period Tr can be shortened. Therefore, the Doppler range that can be detected unambiguously can be further expanded during coded multiplexing transmission.

[0397] In addition, in Modification 1 of Embodiment 2, the case where a radar transmission signal is used in which the center frequency fc of the chirp signal changes by Δf according to the coding transmission period (Loc×Tr) has been described. However, a radar transmission signal in which the center frequency fc of the chirp signal changes by Δf according to (a divisor of Loc×Tr) may also be used. In addition, in the case where 1 among the divisors of Loc is used, the center frequency fc changes by Δf according to Tr, similarly to Embodiment 2.

[0398] In the case where a radar transmission signal is used in which the center frequency fc of the chirp signal changes by Δf according to the transmission period ε which is a divisor of Loc, i.e., according to the ε-th transmission period (ε×Tr), the center frequency fc of the chirp signal changes by an amount equivalent to Δf×Loc / ε in each coding transmission period (Loc×Tr). Therefore, Δf in Expressions (47), (48), and (49) is replaced with Δf×Loc / ε. In addition, the center frequency change correction vector ξ(f b_cfar ) uses Expression (50). As described above, the same effect as in Modification 1 above can be obtained. Here, ε is a divisor of Loc.

[0399]

[0400] In addition, Modification 1 of Embodiment 2 may be implemented in combination with Embodiment 1, but the coding multiplexing method described in Embodiment 1 may not be applied.

[0401] For example, the coding generation unit 104c makes the number of coding multiplexes N allcode in the N CM orthogonal codes included in the coding sequence of code length Loc allcode equal to the number of orthogonal codes N. The phase rotation unit 105 performs coding multiplexing using all N allcode orthogonal codes included in the coding sequence of code length Loc, and outputs a radar transmission signal in which the center frequency fc of the chirp signal changes by Δf according to the coding transmission period (Loc×Tr).

[0402] As a result, the radar device 10c can improve the range resolution by the change amplitude of the center frequency of the chirp signal by increasing Δf×Ncode, and therefore can reduce the chirp sweep frequency band (for example, Bw) compared to the case where the center frequency of the chirp signal is constant during transmission. Therefore, the radar device 10c can improve the range resolution and shorten the transmission period Tr, for example, by reducing the chirp sweep frequency band.

[0403] In addition, the radar device c of Modification 1 of Embodiment 2, when not combined with Embodiment 1, has a structure in which the aliasing determination unit 212 of the radar device 10 is not applied. Therefore, the Doppler frequency range is ±1 / (2Loc×Tr). However, since the chirp sweep band is reduced to less than 1 / Loc by increasing Δf×Ncode, the range resolution can be improved, and the transmission period can be shortened to less than Tr / Loc.

[0404] Therefore, by using a radar transmission signal in which the center frequency fc of the chirp signal changes by Δf over the coded transmission period (Loc×Tr), the radar device 10c of Modification 1 of Embodiment 2 can obtain the effect of expanding the detectable Doppler range without ambiguity even in the case of conventional coded multiplex transmission.

[0405] (Modification 2 of Embodiment 2)

[0406] The method of controlling the center frequency fc of the frequency modulation signal (chirp signal) is not limited to the methods of Embodiment 2 ( Figure 9 ) and Modification 1 ( Figure 11 ). In Modification 2, another method of controlling the center frequency fc of the frequency modulation signal (chirp signal) will be described.

[0407] [Structure of Radar Device]

[0408] Figure 12 is a block diagram showing a structural example of the radar device 10d of Modification 2. In addition, in Figure 12 , the same reference numerals are attached to the same structures as those in Embodiment 1 ( Figure 1 ) or Embodiment 2 ( Figure 8 ), and their descriptions are omitted.

[0409] In Modification 2, for example, the radar device 10d makes the center frequency fc of the chirp signal periodically variable within the coded transmission period (Loc×Tr) over a plurality of transmission periods. In this case, the radar device 10d can perform the aliasing determination process in the aliasing determination unit 212d or the coded multiplex separation process in the coded multiplex separation unit 213d (details will be described later) by making the timing of one cycle of the change in the center frequency fc of the chirp signal coincide with the coded transmission period (Loc×Tr).

[0410] The radar transmission signal generation unit 101d in the radar transmission unit 100d includes a modulation signal generation unit 102, a VCO 103d, and a transmission frequency control unit 107d.

[0411] For example, the modulation signal generation unit 102 periodically generates a sawtooth-shaped modulation signal for controlling the VCO. Here, the transmission period is set to Tr.

[0412] The transmission frequency control unit 107d controls the center frequency fc of the frequency modulation signal (chirp signal) output from the VCO 103d at the transmission period Tr based on the orthogonal code element index OC_INDEX output from the encoding generation unit 104d.

[0413] For example, in the transmission period Tr when OC_INDEX = 1, the transmission frequency control unit 107d sets the center frequency fc of the frequency modulation signal output from the VCO 103d to f0. Additionally, in the transmission period Tr when OC_INDEX = 2, the transmission frequency control unit 107d sets the center frequency fc of the frequency modulation signal output from the VCO 103d to f0 + Δf. Similarly, in each transmission period Tr when OC_INDEX = 3 to Loc, the transmission frequency control unit 107d sets the center frequency fc of the frequency modulation signal output from the VCO 103d to (f0 + 2Δf) to (f0 + (Loc - 1)Δf), respectively.

[0414] Through this control, the transmission frequency control unit 107d can control, for example, the center frequency fc of the chirp signal to periodically change in each encoding transmission period (Loc × Tr).

[0415] Based on the outputs of the transmission frequency control unit 107d and the modulation signal generation unit 102, the VCO 103d outputs the frequency modulation signal to the phase rotation unit 105 and the radar receiving unit 200 (e.g., the mixer unit 204).

[0416] Figure 13 This represents an example of the frequency modulation signal (hereinafter, chirp signal).

[0417] In Figure 13 for example, in the first transmission period Tr#1, the VCO 103d outputs a chirp signal with the center frequency fc(1) being f0. Additionally, for example, in the second transmission period Tr#2, the VCO 103d outputs a chirp signal with the center frequency fc(2) being f0 + Δf. Similarly, in the 3rd transmission period Tr#3 to the Loc transmission period Tr#Loc, the VCO 103d outputs chirp signals with the center frequencies fc(3) to fc(Loc) being (f0 + 2Δf) to (f0 + (Loc - 1)Δf), respectively.

[0418] In addition, in the (Loc + 1)-th transmission period Tr#(Loc + 1), VCO103d outputs a chirp signal with a center frequency fc(Loc + 1) of f0. Similarly, in the (Loc + 2)-th transmission period Tr#(Loc + 2) to the (2Loc)-th transmission period Tr#(2Loc), VCO103d outputs chirp signals with center frequencies fc(Loc + 2) to fc(2Loc) of (f0 + Δf) to (f0 + (Loc - 1)Δf), respectively.

[0419] Similarly, in the m-th transmission period Tr#m, VCO103d outputs a chirp signal with a center frequency fc(m) of f0 + mod(m - 1, Loc)Δf.

[0420] That is, in Figure 13 , the N-th c transmission period Tr#N c the center frequency fc(N c ) of the chirp signal is f0 + (Loc - 1)Δf.

[0421] In this way, in Figure 13 , the change in the center frequency fc of the chirp signal cycles once with the transmission period of the radar transmission signal that is an approximate multiple of the code length of multiple coding sequences (e.g., orthogonal codes).

[0422] In addition, each chirp signal can use, for example, a chirp signal with the same frequency modulation bandwidth Bw within the time range T A of the range gate. Additionally, in the example shown in Figure 13 , the case where Δf > 0 (in other words, the case where the center frequency fc increases) is shown, and the same applies to the case where Δf < 0 (in other words, the case where the center frequency fc decreases).

[0423] Figure 12 Other operations in the radar transmission unit 100d shown in can be the same as those in Embodiment 1.

[0424] Next, an example of the operation in the radar reception unit 200d of the radar device 10d will be described.

[0425] In the radar reception unit 200d, the processing of the signal received by the reception antenna 202 by each antenna system processing unit 201 and the subsequent operations in the CFAR unit 211 are the same as those in Embodiment 1. Additionally, in the radar reception unit 200d, the direction estimation processing performed by the direction estimation unit 214d using the output of the code multiplexing separation unit 213d is also the same as the operation in Embodiment 1.

[0426] In the radar receiving unit 200d, for example, the operations of the aliasing determination unit 212d and the encoding multiplexing / demultiplexing unit 213d are different from those in the first embodiment.

[0427] Hereinafter, an example of the operation different from that in the first embodiment in the aliasing determination unit 212d will be described.

[0428] For example, in the case of using a radar transmission signal in which the center frequency fc of the chirp signal is changed to f0, f0 + Δf,..., f0 + (Loc - 1)Δf at each transmission period Tr within each encoding transmission period (Loc × Tr), in the first, second,..., Loc-th Doppler analysis units 210, the radar reflected wave when the chirp signal with the transmission center frequencies fc being f0, f0 + Δf,..., f0 + (Loc - 1)Δf is input is used as the received signal.

[0429] Therefore, the Loc Doppler analysis units 210 are respectively the same as the center frequency of the input radar reflected wave.

[0430] On the other hand, since the center frequency of the chirp signal is different among the Loc Doppler analysis units 210, in order to correct the phase rotation caused by the time difference of the Doppler analysis among the Loc Doppler analysis units 210, the radar receiving unit 200d, in addition to using the Doppler phase correction vector α(f s_cfar ) used for the aliasing determination unit 212 in the first embodiment, also uses the center frequency change correction vector ζ(f b_cfar ) shown in Equation (51) to correct the phase. That is, Equation (51) uses instead of α(f s_cfar ).

[0431]

[0432] Here, R(f b_cfar ) is the distance estimation value R(f b_cfar ) obtained using the beat frequency index f b_cfar according to Equation (8). Equation (51) is derived based on the following situation. When the center frequency of the transmitted chirp signal of the first Doppler analysis unit 210 is used as a reference, the center frequencies of the transmitted chirp signals of the second to Loc-th Doppler analysis units 210 are different by Δf,..., (Loc - 1)Δf, respectively. As a result, the phase rotation amounts within the arrival time (2R(f b_cfar ) / Co) of the reflected wave from R(f b_cfar ) are different, respectively.

[0433] That is, when the output of the first Doppler analysis unit 210 is used as the phase reference, the phase rotation amount in the noc-th Doppler analysis unit 210 is 2π(noc - 1)Δf × (2R(f b_cfar) / Co). In order to eliminate such phase rotation, the center frequency change correction vector ζ(f shown in Equation (51) is derived b_cfar ). Here, noc = 1, …, Loc.

[0434] In this way, in the following case, that is, when the center frequency fc of the chirp signal is periodically changed within multiple chirp transmission periods during the coding transmission period (Loc × Tr), since the switching period of the Doppler analysis unit 210 for each coding element coincides with the timing of one cycle of the change in the center frequency fc of the chirp signal, the aliasing determination unit 212d can easily perform phase correction in the separation process of the coded multiplexed signal using the unused code.

[0435] As described above, the aliasing determination unit 212d can use Equation (52) instead of Equation (12) to calculate the received power DeMulUnCode after coding separation using the unused orthogonal code UnCode nuc nuc (f b_cfar , f s_cfar , DR). Equation (52) uses to replace α(f s_cfar ) in Equation (12). Here, nuc = 1, …, N allcode -N CM . In addition, DR is an index indicating the Doppler aliasing range, and takes integer values in the range of DR = ceil[-Loc / 2], ceil[-Loc / 2]+1, …, 0, …, ceil[Loc / 2]-1.

[0436]

[0437] In addition, Equation (53) can be used instead of Equation (29).

[0438]

[0439] Next, an operation example of the coded multiplexing separation unit 213d in Modification 2 of Embodiment 2 will be described. Based on the same reason as the description of the operation example of the aliasing determination unit 212d in Modification 2 of Embodiment 2 above, the coded multiplexing separation unit 213d uses Equation (54) instead of Equation (30), and uses the aliasing determination result DR in the aliasing determination unit 212d min , and performs coding separation processing on the Doppler component VFTALL b_cfar corresponding to the distance index f s_cfar extracted in the CFAR unit 211 and the Doppler frequency index f z (f b_cfar , f s_cfar ). Equation (54) is to use​ to replace α(f in Equation (30) s_cfar ) as an example.

[0440]

[0441] In addition, the coded multiplexing separation unit 213d can also use Equation (55) to replace Equation (31), and use the aliasing determination result DR in the aliasing determination unit 212d min , for the range index f extracted in the CFAR unit 211 b_cfar and the Doppler frequency index f s_cfar to perform the separation process of the coded multiplexing signal on the output of the Doppler analysis unit 210 corresponding to the Doppler component VFTALL z (f b_cfar , f s_cfar ). In addition, in Equation (55), the term does not depend on the index f of the Doppler component s , so by pre-tabulating, the computational load in the coded multiplexing separation unit 213d can be reduced.

[0442]

[0443] In this way, the coded multiplexing separation unit 213d makes the center frequency fc of the chirp signal change periodically within multiple chirp transmission periods during the coded transmission period (Loc × Tr), so that the switching period of the Doppler analysis unit 210 for each coding element coincides with the timing of one cycle of the change of the center frequency fc of the chirp signal, and the phase correction in the separation process of the coded multiplexing signal can be easily performed.

[0444] Next, an example of the operation of the direction estimation unit 214d in Modification 2 of Embodiment 2 will be described.

[0445] For example, the direction estimation unit 214d can also be based on the Doppler frequency index f s_cfar and the determination result of the aliasing determination unit 212d, that is, DR min , and calculate the Doppler frequency index f according to Equation (56) es_cfar .

[0446] f es_cfar = f s_cfar + DR min × Ncode (56)

[0447] In addition, the Doppler frequency index f es_cfar for example corresponds to the Doppler index when the FFT size of the Doppler analysis unit 210 is expanded to Loc × Ncode. Hereinafter, f es_cfaris referred to as the "extended Doppler frequency index". In addition, consider a Doppler range up to ±1 / (2×Tr), and the extended Doppler frequency index f corresponding to this Doppler range es_cfar has a range of -Loc×Ncode / 2 ≦ f es_cfar < Loc×Ncode / 2. Therefore, in the calculation result of Equation (56), when f es_cfar < -Loc×Ncode / 2, set f es_cfar +Loc×Ncode as f es_cfar . Additionally, when f es_cfar ≧Loc×Ncode / 2, set f es_cfar -Loc×Ncode as f es_cfar .

[0448] In addition, the direction estimation unit 214d can, for example, use the extended Doppler frequency index f es_cfar and output the Doppler velocity information of the detected target using Equation (57).

[0449]

[0450] As described above, in Variant 2, in the radar device 10d, the center frequency fc of the chirp signal is periodically variable within a plurality of transmission periods. In this case, the timing for one cycle of the change in the center frequency fc of the chirp signal coincides with the coded transmission period (Loc×Tr). In addition, the radar device 10d (e.g., MIMO radar) can also apply coded multiplexing transmission. Additionally, similar to Embodiment 1, the radar device 10d can determine Doppler aliasing using the output of the Doppler analysis unit 210 for each coding element of the coded multiplexing signal (in other words, the received signal) and unused orthogonal codes.

[0451] Furthermore, according to Variant 2, similar to Embodiment 1, the radar device 10d can set the Doppler frequency range that can be detected unambiguously to ±1 / (Tr) and suppress the mutual interference between the coded multiplexing signals to approximately the noise level by performing Doppler phase correction including aliasing during coded separation. Thus, according to Variant 2, it is possible to suppress the degradation of radar detection performance and perform coded multiplexing transmission of the MIMO radar.

[0452] In addition, according to Modification 2, by setting the timing for one cycle of the change in the center frequency fc of the chirp signal to the coding transmission period (Loc × Tr), which is thus made consistent with the switching period of the Doppler analysis unit 210 for each coding element (i.e., the transmission period of the coding for coding multiplexing), it becomes possible to easily perform the separation process of the coding multiplexed signal using the unused coding in the aliasing determination unit 212d and the phase correction in the coding multiplex separation process of the coding multiplex separation unit 213d.

[0453] Furthermore, according to Modification 2, the timing for one cycle of the change in the center frequency fc of the chirp signal is set to the coding transmission period (Loc × Tr), but it can also be set to (a divisor of Loc × Tr). When the timing for one cycle of the change in the center frequency fc of the chirp signal is set to the period of (a divisor ε of Loc × Tr), i.e., the ε-th transmission period (ε × Tr), Equation (58) can be used as the center frequency change correction vector ζ(f b_cfar ). As described above, the same effect as in the above Modification 1 can be obtained. Here, ε is a divisor of Loc and ε > 1.

[0454]

[0455] Moreover, according to Modification 2, a radar transmission signal is used in which the center frequency fc of the chirp signal is set to f0, f0 + Δf,..., f0 + (Loc - 1)Δf at each transmission period Tr within the coding transmission period (Loc × Tr) and changes in integer multiples of Δf, but it is not limited to this, and it can also be variable at any frequency.

[0456] For example, a radar transmission signal can also be used in which the center frequency fc of the chirp signal changes to f0, f0 + Δf1, f0 + Δf2,..., f0 + Δf Loc-1 at each transmission period Tr of the coding transmission period (Loc × Tr). Here, Δf1, Δf2,..., Δf Loc-1 are the frequency variable values of the center frequency fc of the chirp signal at each transmission period Tr of the coding transmission period (Loc × Tr). In this case, Equation (59) is used as the center frequency change correction vector ζ(f b_cfar ). As described above, the same effect as in the above Modification 1 can be obtained.

[0457]

[0458] The above has described one embodiment of the present disclosure.

[0459] (Other Modification 1)

[0460] The transmitting antenna of the above radar device can be a sub-array structure. For example, the radar device can perform Doppler multiplex transmission that combines sub-array beamforming (sub-array BF) and coded multiplex transmission.

[0461] By combining several of the transmitting antennas as sub-arrays, it is possible to narrow the beam width of the transmitting directivity beam pattern and increase the transmitting directivity gain. As a result, although the angular range that can be detected becomes narrower, the detectable distance range can be increased. In addition, by making the beam weight coefficient for generating the directive beam variable, the beam direction can be variably controlled.

[0462] Figure 14 It is a block diagram showing a structural example of the radar transmitting unit 100e of this modification. In addition, in Figure 14 the same reference numerals are attached to the structures that perform the same operations as the radar transmitting unit 100 shown in Figure 1 and their descriptions are omitted. In addition, the radar receiving unit of this modification, for example, shares the basic structure with the radar receiving unit 200 shown in Figure 1 and thus is described by referring to Figure 1 .

[0463] Figure 14 The radar transmitting unit 100e shown in

[0464] has a plurality of transmitting antennas 106e capable of controlling the transmission phase on the output side of the phase rotation unit 105. SA For example, for the outputs of each of the Nt phase rotation units 105, sub-arrays SA#1 to sub-array SA#N SA composed of N Figure 14 transmitting antennas 106e are formed (for example, Nt groups of sub-arrays). In addition, the sub-array structure of the transmitting antennas 106e is not limited to the example shown in SA . For example, the number of transmitting antennas included in the sub-array for the output of each phase rotation unit 105 (in other words, N SA ) can be different between the phase rotation units 105. Here, N SA is an integer of 1 or more. In addition, when N Figure 1 = 1, it is the same as

[0465] In Figure 14 , the beam weight generation unit 108 generates a beam weight that uses the sub-array to direct the main beam direction of the transmission beam to a specified direction. For example, when a sub-array using N SA transmitting antennas is linearly arranged with an element interval d SA , the transmission beam direction is represented as θ TxBF . In this case, the beam weight generation unit 108 generates a beam weight W such as the following equation (60), for example. Tx(Index_TxSubArray, θ TxBF )。

[0466]

[0467] Here, Index_TxSubArray represents the element index of the subarray, where Index_TxSubArray = 1, …, N SA . Additionally, λ represents the wavelength of the radar transmission signal, and d SA represents the subarray antenna spacing.

[0468] Each beam weight multiplication unit 109 multiplies the output from the corresponding phase rotation unit 105 by the beam weight coefficient W Tx (Index_TxSubArray, θ TxBF ). The transmission signal multiplied by the beam weight W Tx (Index_TxSubArray, θ TxBF ) is transmitted by N SA subarray antennas. Here, Index_TxSubArray = 1, …, N SA .

[0469] Regarding subsequent operations, for example, by replacing the transmission antenna 106 (e.g., Tx#1 to Nt) in Embodiment 1 with Figure 14 the transmission antenna 106e shown (e.g., transmission subarray antennas Tx#1 to Nt ) , and performing the same operations, the same effects as in Embodiment 1 or Embodiment 2 can be obtained.

[0470] According to the above operations, the radar transmission unit 100e can transmit a coded multiplexed radar transmission signal in such a way that the transmission directional beam is directed towards a specified direction using the subarray. Thereby, the transmission directional gain in the specified direction can be increased, and the detectable distance range can be expanded. Additionally, the radar transmission unit 100e can variably control the beam direction by variably setting the beam weight coefficient for generating the transmission directional beam.

[0471] Furthermore, in Figure 14 , a structure is shown in which the phase rotation of the phase rotation unit 105 and the beam weight multiplication of the beam weight multiplication unit 109 are performed separately, but it is not limited to this structure. For example, it can also be a structure in which the phase rotation unit 105 includes a phase rotation equivalent to the beam weight multiplication of the beam weight multiplication unit 109. Figure 15 is a block diagram showing a structural example of the radar transmission unit 100f having this structure.

[0472] In Figure 15In the radar transmission unit 100f shown, the phase adder 110 adds the phase rotation with respect to each subarray SA#1 to SA#N output from the beam weight generation unit 108f SA to the phase rotation amount output from the code generation unit 104, and outputs the result to the phase rotation unit 105f.

[0473] For example, the phase adder 110 adds the phase rotation amount ψ ncm (m) given to the ncm-th transmission antenna Tx#ncm output from the code generation unit 104 in each transmission period Tr SA to the phase rotation angle(W Tx (Index_TxSubArray, θ TxBF )) with respect to each subarray SA#1 to SA#N that make up the ncm-th transmission antenna Tx#ncm output from the beam weight generation unit 108f. Further, the phase adder 110 outputs the added value (ψ ncm (m) + angle(W Tx (Index_TxSubArray, θ TxBF ))) to the N SA phase rotation units 105f (e.g., phase shifters) connected to the transmission antenna Tx#ncm.

[0474] Each phase rotation unit 105f gives a phase rotation to the chirp signal output from the radar transmission signal generation unit 101 in each transmission period Tr based on the output from the phase adder 110. Here, ncm = 1,..., N CM , and m = 1,..., Nc.

[0475] In addition, the structure for performing subarray transmission described in this modification is not limited to the structure of Embodiment 1 (e.g., Figure 1 ), and can also be applied to other modifications or embodiments (e.g., Figure 4 , Figure 8 , Figure 10 or Figure 12 ). For example, Figure 14 and Figure 15 the structures of the radar transmission units 100e or 100f shown can also be applied to the radar devices 10a, 10b, 10c, and 10d.

[0476] (Other Modification 2)

[0477] In the above embodiment, the case where the Walsh - Hadamard code is used as the code generated in the code generation units 104, 104c, and 104d has been described, but the code is not limited to the Walsh - Hadamard code, and other codes can also be used.

[0478] For example, an orthogonal M-sequence code or a pseudo-orthogonal code may also be used as the orthogonal code generated by the encoding generation units 104, 104c, and 104d. Hereinafter, an operation example in the case of using an orthogonal M-sequence code and a pseudo-orthogonal code will be described.

[0479] <Orthogonal M-sequence code>

[0480] The orthogonal M-sequence code is, for example, a coding sequence in which a coding element is added to the sequence generated from the M-sequence so that the numbers of "1" and "-1" of the coding element are equal (in other words, balanced) (for example, refer to Non-Patent Document 5). In addition, since the code including the above coding element and the code in which all coding elements are "1" are in an orthogonal relationship, hereinafter, the case where the orthogonal M-sequence code also includes the code in which all coding elements are 1 will be described.

[0481] As an example, the orthogonal M-sequence code with a code length of 4 includes the following codes.

[0482] OMS4(1) = [1 1 -1 -1]

[0483] OMS4(2) = [-1 1 1 -1]

[0484] OMS4(3) = [1 -1 1 -1]

[0485] OMS4(4) = [1 1 1 1]

[0486] Here, the orthogonal M-sequence code with a code length of Loc is represented as OMS Loc (nomc). Among them, nomc represents the coding index included in the orthogonal M-sequence code with a code length of Loc. nomc = 1,..., Loc.

[0487] For example, OMS4(1) and OMS4(2) are a group of codes in which the even-numbered coding elements of the codes are the same, and the positive and negative signs of the odd-numbered coding elements are reversed. For example, when the number of unused orthogonal codes (N allcode -N CM ) is two or more, the encoding generation units 104, 104c, or 104d can select codes in such a way that the unused orthogonal codes do not include a group of codes in the relationship such as OMS4(1) and OMS4(2). Through this code selection, for example, aliasing determination of the Doppler frequency in the aliasing determination unit 212 of the radar receiving unit 200 can be performed.

[0488] Similarly, OMS4(3) and OMS4(4) are a group of codes in which the odd-numbered coding elements of the codes are the same, and the positive and negative signs of the even-numbered coding elements are reversed. For example, when the number of unused orthogonal codes (N allcode -N CM)When there are two or more, the coding generation units 104, 104c, or 104d can select codings in a group of codings where unused orthogonal codes do not include relationships such as OMS4(3) and OMS4(4). Through this coding selection, for example, it is possible to perform aliasing determination of the Doppler frequency in the aliasing determination unit 212 of the radar receiving unit 200.

[0489] In addition, it is not limited to the code length Loc = 4 of the orthogonal M - series coding, and other code lengths can also be used. For example, the following codings are included in the orthogonal M - series coding with the code length Loc = 8.

[0490] OMS4(1)=[-1 1 -1 -1 1 1 1 -1]

[0491] OMS4(2)=[1 -1 1 -1 -1 1 1 -1]

[0492] OMS4(3)=[1 1 -1 1 -1 -1 1 -1]

[0493] OMS4(4)=[1 1 1 -1 1 -1 -1 -1]

[0494] OMS4(5)=[-1 1 1 1 -1 1 -1 -1]

[0495] OMS4(6)=[-1 -1 1 1 1 -1 1 -1]

[0496] OMS4(7)=[1 -1 -1 1 1 1 -1 -1]

[0497] OMS4(8)=[1 1 1 1 1 1 1 1]

[0498] For example, the number of codings N CM The code length Loc of the orthogonal M - series codings can be expressed by the following formula (61).

[0499]

[0500] Here, ceil[x] is an operator (ceiling function) that outputs the smallest integer greater than or equal to the real number x.

[0501] The coding generation units 104, 104c, or 104d, for example, set N allcode (Loc) codings among the N codings included in the orthogonal M - series coding with the code length Loc as the codings for coding multiplexing transmission. For example, in the case of the orthogonal M - series coding, N CM orthogonal codes are set as the codings for coding multiplexing transmission. allcode(Loc) = the relationship of Loc holds. For example, since the orthogonal M-sequence codes with code lengths Loc = 4, 8, or 16 contain 4, 8, or 16 orthogonal codes respectively, thus N allcode (4) = 4, N allcode (8) = 8 and N allcode (16) = 16.

[0502] <Pseudo-orthogonal code>

[0503] For example, in Embodiment 1, in the coding multiplexing and demultiplexing unit 213 (e.g., Figure 1 ), when using an orthogonal code, the orthogonal code Code used in coding multiplexing transmission is used ncm to separate the signal subjected to coding multiplexing. In contrast, when using a pseudo-orthogonal code, the following aspect is different from the orthogonal code, that is, the coding multiplexing and demultiplexing unit 213, for example, uses the inverse code "InvCode ncm " for coding multiplexing separation for the pseudo-orthogonal code Code used in coding multiplexing transmission ncm to separate the signal subjected to coding multiplexing.

[0504] In addition, for example, in Embodiment 1, in the aliasing determination unit 212 (e.g., Figure 1 ), when using an orthogonal code, the unused orthogonal code UnCode nuc is used for aliasing determination. In contrast, when using a pseudo-orthogonal code, the following aspect is different from the orthogonal code, that is, the aliasing determination unit 212, for example, uses the inverse code "InvUnCode nuc " for separating the unused pseudo-orthogonal code UnCode nuc for aliasing determination.

[0505] Furthermore, the inverse code InvCode ncm is a code derived from the pseudo-orthogonal code.

[0506] Hereinafter, an example of the operation different from that in Embodiment 1 in the aliasing determination unit 212 and the coding multiplexing and demultiplexing unit 213 of this modification will be described.

[0507] The coding generation unit 104, for example, sets N allcode of the N CM pseudo-orthogonal codes contained in the pseudo-orthogonal code sequence of code length Loc (in other words, the code sequence orthogonal to the corresponding inverse code) as the codes for coding multiplexing transmission.

[0508] For example, the coding multiplexing number N CM is less than the number of pseudo-orthogonal codes N allcode , N CM < N allcodeThat is to say, the code length Loc of the pseudo-orthogonal code is greater than the coding multiplexing number N CM For example, N of the code length Loc CM pseudo-orthogonal codes are expressed as Code ncm =[OC ncm (1), OC ncm (2), …, OC ncm (Loc)].

[0509] In addition, the code length Loc of the pseudo-orthogonal code is not limited to the power of 2 and can be any code length. Thus, for example, the code length Loc of the pseudo-orthogonal code can be a value obtained by adding 1 to the coding multiplexing number N CM (Loc = N CM +1). In this case, for example, N allcode = N CM +1, and a pseudo-orthogonal code sequence with the code length Loc = N allcode can be generated.

[0510] Hereinafter, as an example, a case of generating N pseudo-orthogonal code sequences by using a part of the Walsh-Hadamard code with the code length Loc CM is described. However, the method for generating the pseudo-orthogonal code sequence is not limited thereto.

[0511] As an example, a case where the coding multiplexing number N CM = 4 and a part of the Walsh-Hadamard code with the code length 8 is used to generate a pseudo-orthogonal square matrix with the code length Loc = 5 where N allcode = N CM +1 = 5 is described.

[0512] For example, the coding generation unit 104 can use 5 (= Loc) coding elements (for example, the first to fifth of each coding element: WH8(1) to WH8(5)) in the Walsh-Hadamard code with the code length 8 to generate the following pseudo-orthogonal codes PWH5(1) to pseudo-orthogonal code PWH5(5).

[0513] PWH5(1) = [1 1 1 1 1]

[0514] PWH5(2) = [1 -1 1 -1 1]

[0515] PWH5(3) = [1 1 -1 -1 1]

[0516] PWH5(4) = [1 -1 -1 1 1]

[0517] PWH5(5) = [1 1 1 1 -1]

[0518] For example, the inverse coding square matrix "InvPOC" of the coding square matrix "POC" with the pseudo-orthogonal codes PWH5(1) to PWH5(5) as row elements can be derived according to the following formula (62). Here, the superscript H is the complex conjugate transpose operator.

[0519] InvPOC = POC H (POC H POC) -1 (62)

[0520] For example, the POC composed of the above pseudo-orthogonal codes PWH5(1) to PWH5(5) is the coding square matrix shown in the following formula (63).

[0521]

[0522] In addition, the inverse coding square matrix InvPOC of the coding square matrix POC shown in formula (63) is calculated, for example, in the manner of the following formula (64). In addition, each row of the inverse coding square matrix InvPOC is expressed as inverse coding InvPWH5(1) to inverse coding InvPWH5(5).

[0523]

[0524] Here, the pseudo-orthogonal code PWH5(npoc1) and the inverse coding InvPWH5(npoc2) have the following relationship in formula (65). Among them, npoc represents the coding index contained in the pseudo-orthogonal code or the inverse coding. For example, npoc1 = 1,..., Loc, npoc2 = 1,..., Loc. In addition, npoc1 represents the coding index npoc1 of the pseudo-orthogonal code PWH5, and npoc2 represents the coding index npoc2 of the inverse coding InvPWH5, for example.

[0525]

[0526] For example, for the pseudo-orthogonal code PWH5(npoc) generated by the coding generation unit 104, the coding for multiplexed transmission with the coding multiplexing number N CM = 4 is set as Code1 = PWH5(1), Code2 = PWH5(2), Code3 = PWH5(3), and Code4 = PWH5(4). In this case, the unused pseudo-orthogonal code is UnCode1 = PWH5(5), and the inverse coding for the unused pseudo-orthogonal code UnCode1 is InvUnCode1 = InvPWH5(5).

[0527] For example, the coding generation unit 104 when the number of unused pseudo-orthogonal codes (N allcode - N CM) When there are two or more, the coding can also be selected in the following manner, that is, the odd-numbered coding elements among the inverse codings of the pseudo-orthogonal codes with respect to the code length Loc are the same, and the groups of codings in which the signs of the even-numbered coding elements are reversed are not included in the unused pseudo-orthogonal codes. By this selection of the unused pseudo-orthogonal codes, the aliasing determination accuracy of the Doppler frequency in the aliasing determination unit 212 of the radar receiving unit 200 can be improved.

[0528] In addition, the groups of codings in which the odd-numbered coding elements among the inverse codings of the pseudo-orthogonal codes with respect to the code length Loc are the same, and the signs of the even-numbered coding elements are reversed, have the following relationship, that is, the odd-numbered coding elements among the codings are the same, and the signs of the even-numbered coding elements are reversed. Therefore, when the number of unused pseudo-orthogonal codes (N allcode -N CM ) is two or more, the coding can also be selected in the following manner, that is, the groups of codings in which the odd-numbered coding elements among the codings of the pseudo-orthogonal codes with the code length Loc are the same, and the signs of the even-numbered coding elements are reversed, are not included in the unused pseudo-orthogonal codes. By this selection of the unused pseudo-orthogonal codes, the aliasing determination accuracy of the Doppler frequency in the aliasing determination unit 212 of the radar receiving unit 200 can be improved.

[0529] As another example, the coding multiplexing number N CM = 4 and a part of the Walsh-Hadamard code with the code length 8 is used to generate N allcode = N CM +2 = 6 of the pseudo-orthogonal square matrix with the code length Loc = 6 will be described.

[0530] For example, the coding generation unit 104 can use six (= Loc) coding elements (for example, the first to sixth of each coding element: WH8(1) to WH8(6)) in the Walsh-Hadamard code with the code length 8 to generate the following pseudo-orthogonal codes PWH6(1) to pseudo-orthogonal code PWH6(6).

[0531] PWH6(1) = [1 1 1 1 1 1]

[0532] PWH6(2) = [1 -1 1 -1 1 -1]

[0533] PWH6(3) = [1 1 -1 -1 1 1]

[0534] PWH6(4) = [1 -1 -1 1 1 -1]

[0535] PWH6(5) = [1 1 1 1 -1 -1]

[0536] PWH6(6) = [1 -1 1 -1 -1 1]

[0537] For example, the inverse coding InvPWH6(1) to InvPWH6(6) with respect to the pseudo-orthogonal codes PWH6(1) to PWH6(6) can be calculated as follows. Here, the inverse coding with respect to the pseudo-orthogonal code PWH6(npoc) is InvPWH6(npoc). Among them, npoc = 1, …, Loc.

[0538] InvPWH6(1) = 0.25 × [0 0 1 1 1 1]

[0539] InvPWH6(2) = 0.25 × [0 0 1 -1 1 -1]

[0540] InvPWH6(3) = 0.25 × [1 1 -1 -1 0 0]

[0541] InvPWH6(4) = 0.25 × [1 -1 -1 1 0 0]

[0542] InvPWH6(5) = 0.25 × [1 1 0 0 -1 -1]

[0543] InvPWH6(6) = 0.25 × [1 -1 0 0 -1 1]

[0544] Here, InvPWH6(1) and InvPWH6(2) in the inverse coding are a group of inverse codings where the odd-numbered coding elements are the same between the codings, and the signs of the even-numbered coding elements are reversed (except for the coding elements that are 0). In addition, the corresponding codes PWH6(1) and PWH6(2) to InvPWH6(1) and InvPWH6(2) respectively are also a group of codes where the odd-numbered coding elements are the same between the codes, and the signs of the even-numbered coding elements are reversed.

[0545] Similarly, the group of InvPWH6(3) and InvPWH6(4), and the group of InvPWH6(5) and InvPWH6(6) are also groups of inverse codings with the same relationship between each other as the group of InvPWH6(1) and InvPWH6(2). In addition, the group of InvPWH6(3) and InvPWH6(4), the group of InvPWH6(5) and InvPWH6(6), and the groups of the corresponding codes to the group of InvPWH6(1) and InvPWH6(2), that is, the group of PWH6(3) and PWH6(4), and the group of PWH6(5) and PWH6(6), are all groups of codes with the same relationship between each other as the group of PWH6(1) and PWH6(2).

[0546] For example, when the number (N allcode -N CM ) of unused pseudo-orthogonal codes is two or more, the radar device 10 may also select codes in such a way that the pseudo-orthogonal codes corresponding to the inverse-coded groups of this relationship are not included in the unused pseudo-orthogonal codes.

[0547] For example, the combination of codes (Code1, Code2, Code3, and Code4) for multiplexed transmission by coding may be, for example, the combination of Code1 = PWH6(1), Code2 = PWH6(3), Code3 = PWH6(5), and Code4 = PWH6(6). In this case, the unused pseudo-orthogonal codes are, for example, UnCode1 = PWH6(2) and UnCode2 = PWH6(4). In addition, the combination of the codes for multiplexed transmission by coding and the unused pseudo-orthogonal codes is not limited to these.

[0548] Next, an operation example of the aliasing determination unit 212 in the case of using pseudo-orthogonal codes will be described.

[0549] The aliasing determination unit 212, for example, uses the inverse codes corresponding to the (N allcode -N CM ) unused pseudo-orthogonal codes in the pseudo-orthogonal codes of code length Loc to perform aliasing determination. In the aliasing determination unit 212, other operations are the same as those in the first embodiment except that pseudo-orthogonal codes are used instead of orthogonal codes.

[0550] For example, the aliasing determination unit 212 corrects the phase change of the Doppler component including aliasing based on the output of the Doppler analysis unit 210 in each antenna system processing unit 201, and calculates the received power DeMulUnCode nuc after code separation using the inverse code InvUnCode nuc corresponding to the unused pseudo-orthogonal code UnCode nuc (f b_cfar , f s_cfar ) according to the following formula (66) instead of according to formula (12).

[0551]

[0552] In formula (66), by calculating the output of the Doppler analysis unit 210 in all antenna system processing units 201, the inverse code InvUnCode nuc corresponding to the unused pseudo-orthogonal code UnCode nucThe sum of the received powers after coding separation, so that even when the received signal level is low, the aliasing determination accuracy can be improved. However, it can also be that, instead of Equation (66), for the output of the Doppler analysis unit 210 in a part of the antenna system processing unit 201, the received power after coding separation using the inverse coding corresponding to the unused pseudo-orthogonal code is calculated. Even in this case, for example, the aliasing determination accuracy can be maintained within a range where the received signal level is sufficiently high, and the amount of arithmetic processing can be reduced.

[0553] In addition, in Equation (53), nuc = 1,..., N allcode -N CM . Additionally, DR is an index representing the Doppler aliasing range. For example, it takes integer values in the range of DR = ceil[-Loc / 2], ceil[-Loc / 2]+1,..., 0,..., ceil[Loc / 2]-1.

[0554] Similarly, the received power DeMulUnCode nuc after coding separation using the inverse coding InvUnCode nuc corresponding to the unused pseudo-orthogonal code UnCode nuc (f b_cfar , f s_cfar , DR) can also be deformed in the following manner of Equation (67) instead of Equation (29).

[0555]

[0556] Next, an operation example of the code multiplexing and separation unit 213 in the case of using a pseudo-orthogonal code is described.

[0557] Based on the aliasing determination result in the aliasing determination unit 212, the code multiplexing and separation unit 213 performs separation processing of the code multiplexed signal. For example, in the separation processing of the code multiplexed signal, the code multiplexing and separation unit 213 separates the code multiplexed signal by the inverse coding InvCode ncm corresponding to the pseudo-orthogonal code Code ncm used for code multiplexing transmission.

[0558] For example, the code multiplexing and separation unit 213 replaces Equation (30) with the following Equation (68), and based on the aliasing phase correction vector β(DR min ) using the aliasing determination result DR min in the aliasing determination unit 212, for the output Doppler component VFTALL b_cfar of the Doppler analysis unit 210 corresponding to the distance index f s_cfar extracted in the CFAR unit 211 and the Doppler frequency index f z (fb_cfar , f s_cfar ) performs coding separation processing.

[0559]

[0560] Here, DeMul z ncm (f b_cfar , f s_cfar ) is the inverse code InvCode corresponding to the pseudo-orthogonal code Code b_cfar and the Doppler frequency index f s_cfar using the output of the range index f ncm for the Doppler analysis unit 210 in the z-th antenna system processing unit 201, ncm , which is the output (e.g., coding separation result) of coding separation of the coded multiplexed signal. In addition, z = 1,..., Na, ncm = 1,..., N CM .

[0561] In addition, the coding multiplexing separation unit 213 can use the following formula (69) instead of formula (68).

[0562]

[0563] In formula (69), the term of (however, in formula (69), DR = DR min ) does not depend on the index of the Doppler component (e.g., Doppler frequency index) f s , so, for example, by tabulating it in advance, the computational amount in the coding multiplexing separation unit 213 can be reduced.

[0564] Through the operations of the coding generation unit 104, the aliasing determination unit 212, and the coding multiplexing separation unit 213 using the inverse code corresponding to the pseudo-orthogonal code as described above, even when using a pseudo-orthogonal code, the same effect as in Embodiment 1 can be obtained.

[0565] In addition, the coding structure described in this modification is not limited to the structure of Embodiment 1 (e.g., Figure 1 ), and can also be applied to other modifications or embodiments (e.g., Figure 4 , Figure 8 , Figure 10 or Figure 12 ). For example, the coding in this modification (e.g., orthogonal M-series coding or pseudo-orthogonal code) can be used instead of the Walsh-Hadamard code in the radar devices 10a, 10b, 10c, and 10d.

[0566] (Other Modification 3)

[0567] For example, the operation of the aliasing determination unit 212 will be described in the following case where there are multiple targets with substantially equal received levels at the same distance index f b_cfar and the intervals between the Doppler peaks of the multiple targets are the same as the intervals of Doppler aliasing.

[0568] Here, since the Doppler frequencies between the multiple targets may be different, the relative moving speeds between the targets and the radar device 10 may be different. Therefore, in the radar device 10, by continuously performing radar observations, even if it is difficult to distinguish aliasing in the radar positioning output at a certain point in time, the possibility that the distances between multiple targets are measured as different can be increased in the radar positioning output at a subsequent point in time.

[0569] Furthermore, in order to more reliably separate the signals corresponding to multiple targets, for example, the radar device 10 can variably set at least one of the transmission period Tr and the code length of the orthogonal code (or pseudo-orthogonal code) generated by the code generation unit 104 each time radar positioning is performed (for example, for every Nc transmission periods (Nc × Tr)) while continuously performing radar positioning.

[0570] For example, the radar device 10 can switch the code length of the orthogonal code (or pseudo-orthogonal code) generated by the code generation unit 104 each time radar positioning is performed and perform coded multiplexing transmission.

[0571] For example, set Loc = N CM + δ. By variably setting δ each time radar positioning is performed, the radar device 10 can vary the interval of Doppler aliasing. At this time, the radar device 10 can use an orthogonal code for transmission when Loc is a power of 2, or can use a pseudo-orthogonal code for transmission when Loc is not a power of 2. For example, δ can be variably set periodically in a pattern such as 1, 2, 1, 2,... each time radar positioning is performed.

[0572] Alternatively, the radar device 10 can, for example, switch between the following (a) and (b) each time radar positioning is performed, thereby switching the code length and type of the code for coded multiplexing transmission and performing coded multiplexing transmission. The interval of Doppler aliasing can be varied.

[0573] (a): The code generation unit 104 uses N orthogonal codes in Walsh-Hadamard codes or orthogonal M-sequence codes with a code length of CM for coded multiplexing transmission.

[0574] (b): The code generation unit 104 uses N CM pseudo-orthogonal codes in pseudo-orthogonal codes with a code length of Loc = N CM + δ for coded multiplexing transmission. Among them, use as of δ.

[0575] Thus, for example, even when the reception levels of the Doppler peaks of multiple targets are approximately equal at the same distance index f b_cfar and the intervals of the Doppler peaks are consistent with the intervals of Doppler aliasing, the possibility that the intervals of Doppler aliasing are different in other radar localizations (e.g., the next radar localization) can be further increased. Thus, the radar device 10 can more reliably separate the signals corresponding to multiple targets.

[0576] In addition, the radar device 10 can obtain the same effect, for example, by variably setting the transmission period Tr for each radar localization performed, thereby changing the interval of Doppler aliasing.

[0577] Furthermore, the structure described in this modification is not limited to the structure of Embodiment 1 (e.g., Figure 1 ), and can also be applied to other modifications or embodiments (e.g., Figure 4 , Figure 8 , Figure 10 or Figure 12 ). For example, the setting of at least one of the transmission period Tr and the code length Loc in this modification can also be applied to the radar devices 10a, 10b, 10c, and 10d.

[0578] The above has described Other Modifications 1 to 3.

[0579] In the radar device according to an embodiment of the present disclosure, the radar transmission unit and the radar reception unit may also be independently arranged at physically separated locations. In addition, in the radar reception unit according to an embodiment of the present disclosure, the direction estimation unit and other constituent units may also be independently arranged at physically separated locations.

[0580] Although not shown, the radar device according to an embodiment of the present disclosure, for example, has a CPU (Central Processing Unit), a recording medium such as a ROM (Read Only Memory) storing a control program, and a working memory such as a RAM (Random Access Memory). At this time, the functions of the above respective parts are realized by the CPU executing the control program. However, the hardware structure of the radar device is not limited to this example. For example, each functional unit of the radar device may also be realized as an IC (Integrated Circuit) of an integrated circuit. Each functional unit may be independently monolithicized, or may be monolithicized in a manner including a part or all of them.

[0581] As described above with reference to the accompanying drawings, various embodiments have been illustrated, but the present disclosure is of course not limited to these examples. It will be obvious to those skilled in the art that various modification examples or correction examples can be conceived within the scope recited in the claims, and it should be understood that these modification examples or correction examples also naturally belong to the technical scope of the present disclosure. In addition, the constituent elements of the above-described embodiments can be arbitrarily combined without departing from the gist of the disclosure.

[0582] In addition, the expression “... part” in the above-described embodiments can also be replaced with other expressions such as “... circuitry”, “... assembly”, “device element”, “unit”, or “module”.

[0583] In each of the above-described embodiments, an example in which the present disclosure is configured by hardware has been described, but the present disclosure can also be implemented by software in cooperation with hardware.

[0584] In addition, each functional block used in the description of the above-described embodiments is typically implemented as an integrated circuit, i.e., LSI (Large Scale Integration). The integrated circuit can also control each functional block used in the description of the above-described embodiments and includes an input terminal and an output terminal. These functional blocks can be individually monolithized or can be monolithized in a manner including a part or all of them. Here, it is called “LSI”, but depending on the degree of integration, it can also be called “IC”, “system LSI”, “ultra large LSI”, or “extra large LSI”.

[0585] In addition, the method of integrating into an integrated circuit is not limited to LSI, and it can also be implemented using a dedicated circuit or a general-purpose processor. It is also possible to use an FPGA (Field Programmable Gate Array) that can be programmed after the LSI is manufactured, or a reconfigurable processor that can reconfigure the connection or setting of the circuit blocks inside the LSI.

[0586] Furthermore, if, with the progress of semiconductor technology or other derived technologies, an integrated circuit technology that replaces LSI appears, of course, that technology can be used to integrate the functional blocks. There is also a possibility of applying biotechnology or the like.

[0587] <Summary of the present disclosure>

[0588] A radar device according to an embodiment of the present disclosure includes: a signal generation circuit that generates a baseband signal; a coding generation circuit that generates a plurality of coding sequences; a phase rotation circuit that adds a phase rotation based on a part of the plurality of coding sequences to the baseband signal to generate a plurality of multiplexed transmission signals; and a plurality of transmission antennas that respectively transmit the plurality of transmission signals; the code lengths of the plurality of coding sequences are greater than the coding multiplexing number for the plurality of transmission signals.

[0589] In an embodiment of the present disclosure, the plurality of coding sequences are orthogonal code sequences, and the code length is a power of 2.

[0590] In an embodiment of the present disclosure, the plurality of coding sequences are pseudo-orthogonal code sequences, and the code length is a value obtained by adding 1 to the coding multiplexing number.

[0591] In an embodiment of the present disclosure, among the first coding sequence and the second coding sequence included in the plurality of coding sequences, the coding elements of one of the odd-numbered coding elements and the even-numbered coding elements are the same, while the signs of the coding elements of the other of the odd-numbered coding elements and the even-numbered coding elements are reversed, and one of the first coding sequence and the second coding sequence is included in other coding sequences different from a part of the plurality of coding sequences.

[0592] In an embodiment of the present disclosure, each of the plurality of transmission antennas transmits a transmission signal that is a transmission signal with different phase rotations added according to each positioning performed by the radar device.

[0593] In an embodiment of the present disclosure, the code lengths of the plurality of coding sequences are different according to each positioning performed by the radar device.

[0594] In an embodiment of the present disclosure, the transmission periods of the plurality of transmission signals are different according to each positioning performed by the radar device.

[0595] In an embodiment of the present disclosure, each of the plurality of transmission antennas has a sub-array structure.

[0596] A radar device according to an embodiment of the present disclosure includes: a signal generation circuit that generates a baseband signal; a coding generation circuit that generates a plurality of coding sequences; a phase rotation circuit that adds a phase rotation based on a part of the plurality of coding sequences to the baseband signal to generate a plurality of multiplexed transmission signals; and a plurality of transmission antennas that respectively transmit the plurality of transmission signals; the plurality of transmission signals are chirp signals, and the center frequency of the chirp signals changes according to the transmission period of the transmission signals or according to the transmission period of the coding sequences.

[0597] A radar device according to an embodiment of the present disclosure includes: a signal generation circuit that generates a baseband signal; a coding generation circuit that generates a plurality of coding sequences; a phase rotation circuit that adds a phase rotation based on a part of the plurality of coding sequences to the baseband signal to generate a plurality of transmitted signals after coding multiplexing; and a plurality of transmitting antennas that respectively transmit the plurality of transmitted signals; the plurality of transmitted signals are chirp signals, and the change in the center frequency of the chirp signals makes one cycle in the transmission period of the transmitted signals that is an integral multiple of the code length of the plurality of coding sequences.

[0598] A radar device according to an embodiment of the present disclosure includes: a receiving antenna that receives a reflected wave signal, which is a signal reflected from a target of a transmitted signal that has been coded and multiplexed based on a part of a plurality of coding sequences; and a receiving circuit that determines aliasing in the Doppler frequency domain of the reflected wave signal based on another coding sequence different from the part of the plurality of coding sequences.

[0599] In an embodiment of the present disclosure, the receiving circuit determines the aliasing within a range that is a multiple of the code length of the coding sequence compared to the Doppler analysis range for the reflected wave signal.

[0600] In an embodiment of the present disclosure, the receiving circuit performs coding separation processing on the reflected wave signal based on the determination result of the aliasing and the part of the coding sequences.

[0601] The entire disclosure of the specification, drawings, and abstract of the Japanese patent application No. 2019-208153 filed on November 18, 2019, is incorporated herein by reference.

[0602] Industrial Applicability

[0603] The present disclosure is suitable as a radar device for detecting a wide-angle range.

[0604] Explanation of Reference Numerals

[0605] 10, 10a, 10b, 10c, 10d Radar device

[0606] 100, 100b, 100c, 100d, 100e, 100f Radar transmission unit

[0607] 101, 101b, 101c, 101d Radar transmission signal generation unit

[0608] 102 Modulation signal generation unit

[0609] 103, 103b, 103c, 103d VCO

[0610] 104, 104c, 104d Encoding Generation Unit

[0611] 105, 105f Phase Rotation Unit

[0612] 106, 106e, 106f Transmission Antenna

[0613] 107, 107c, 107d Transmission Frequency Control Unit

[0614] 108, 108f Beam Weight Generation Unit

[0615] 109 Beam Weight Multiplication Unit

[0616] 110 Phase Addition Unit

[0617] 200, 200a, 200b, 200c, 200d Radar Receiver

[0618] 201 Antenna System Processing Unit

[0619] 202 Receiving Antenna

[0620] 203 Receiving Radio Unit

[0621] 204 Mixer Unit

[0622] 205 LPF

[0623] 206 Signal Processing Unit

[0624] 207 AD Conversion Unit

[0625] 208 Beat Frequency Analysis Unit

[0626] 209 Output Switching Unit

[0627] 210 Doppler Analysis Unit

[0628] 211 CFAR Unit

[0629] 212, 212a, 212c, 212d Aliasing Judgment Unit

[0630] 213, 213a, 213c, 213d Encoding Multiplexing and Demultiplexing Unit

[0631] 214, 214b, 214c, 214d Direction Estimation Unit

[0632] 215 Phase Correction Unit

Claims

1. A radar device, characterized in that, Comprising: A signal generation circuit that generates a baseband signal; An encoding generation circuit that generates a plurality of orthogonal code sequences; A phase rotation circuit that adds a phase rotation based on a part of the orthogonal code sequences among the plurality of orthogonal code sequences to the baseband signal to generate a plurality of transmitted signals after encoding multiplexing; And A plurality of transmitting antennas that respectively transmit the plurality of transmitted signals, The code lengths of the plurality of orthogonal code sequences are greater than the encoding multiplexing number for the plurality of transmitted signals, The plurality of orthogonal code sequences include multiple pairs of orthogonal code sequences in which the encoding elements of one of the odd-numbered encoding elements and the even-numbered encoding elements are the same, and the encoding elements of the other of the odd-numbered encoding elements and the even-numbered encoding elements have opposite signs; In the case where there are two or more other orthogonal code sequences different from the part of the orthogonal code sequences among the plurality of orthogonal code sequences, the two or more different other orthogonal code sequences are respectively orthogonal code sequences that are not the multiple pairs of orthogonal code sequences.

2. The radar device according to claim 1, wherein The plurality of orthogonal code sequences are orthogonal code sequences, and the code length is a power of 2.

3. The radar device according to claim 1, wherein The plurality of orthogonal code sequences are pseudo-orthogonal code sequences, the code length is different from a power of 2, and is a value obtained by adding 1 to the encoding multiplexing number.

4. The radar device according to claim 1, wherein Each of the plurality of transmitting antennas transmits a transmitted signal that is a transmitted signal with a different phase rotation added for each positioning performed by the radar device.

5. The radar device according to claim 1, wherein The code lengths of the plurality of orthogonal code sequences are different for each positioning performed by the radar device.

6. The radar device according to claim 1, wherein The transmission periods of the plurality of transmitted signals are different for each positioning performed by the radar device.

7. The radar device according to claim 1, wherein Each of the plurality of transmitting antennas has a sub-array structure.

8. The radar device according to claim 1, wherein, Further comprising: A receiving antenna that receives a reflected wave signal, which is a signal reflected by a target of a transmitted signal that has been encoded and multiplexed based on a part of the orthogonal code sequences among the plurality of orthogonal code sequences; And A receiving circuit that determines the aliasing in the Doppler frequency domain of the reflected wave signal based on other orthogonal code sequences different from the part of the orthogonal code sequences among the plurality of orthogonal code sequences.

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