Outer rotor permanent magnet synchronous motor position sensorless control system and control method

Abstract

The application belongs to the field of position sensorless control system and control method of outer rotor permanent magnet synchronous motor, and is based on the component -ψ f sinθ e and ψ f cosθ e of the permanent magnet flux linkage in the extended back electromotive force component in the voltage equation under the two-phase static αβ coordinate system as the reconstructed component of the sliding mode observer, and the estimated permanent magnet flux linkage component is subsequently sent into the bilinear extended state observer phase-locked loop with compensation mechanism, and the estimated disturbance is sent into the variable observer coefficient bilinear extended state observer phase-locked loop with compensation mechanism, which is responsible for the output of the estimated rotor position and the estimated speed. The application can run in the full operating range including the zero low speed range, has good dynamic performance and anti-disturbance ability, and can also provide accurate rotor position information under the action of slope disturbance.

Description

Technical Field

[0001] This invention belongs to the field of sensorless permanent magnet synchronous motors, and particularly relates to a sensorless control system and control method applicable to the entire operating range of an external rotor permanent magnet synchronous motor, as well as a maximum torque-current ratio control system using the sensorless control system and control method. Background Technology

[0002] External rotor permanent magnet synchronous motors (PMSMs) are widely used in mining conveyor belts, new energy vehicles, ball mills, and other fields due to their high efficiency, energy saving, simple structure, and large starting torque. However, the harsh working environments in these fields result in high sensor failure rates. Sensor failures can lead to the loss of control of the entire system, increasing costs and reducing robustness. To address these issues, sensorless control algorithms have been proposed. Among them, sliding mode observers are widely used in sensorless control of PMSMs due to their simple structure, insensitivity to system parameter changes, and strong resistance to external disturbances. Traditional sliding mode observers use back electromotive force (EMF) as the reconstructed quantity. The back EMF is proportional to the motor's electrical angular velocity. When the motor operates at rated speed or in the medium-to-high speed range, the back EMF is relatively large, and traditional sliding mode observers can achieve good observation performance. However, when the motor operates at zero speed, the back EMF is small, resulting in a low signal-to-noise ratio, making it difficult to extract accurate rotor position information. To address these issues, academia and industry have conducted extensive research, and existing solutions mainly include the following:

[0003] 1. Open-loop control is used during the motor start-up phase. When the motor speed rises to the medium-high speed stage, it switches to closed-loop control. At this time, the motor back electromotive force is large, and the sliding mode observer can obtain better observation performance.

[0004] 2. Use high-frequency signal injection in the zero-low speed stage, and switch to sliding mode observer in the medium-high speed stage.

[0005] However, the two methods mentioned above have some insurmountable drawbacks: the open-loop control performance is poor, making it difficult to meet the requirements of applications with high control precision, while the high-frequency signal injection method has unavoidable noise problems. At the same time, during the switching process, fluctuations in speed and torque are inevitable, which may even lead to switching failure in some cases, and thus cannot meet the requirements of some applications with high control precision.

[0006] In conclusion, the problem of poor observation performance of traditional sliding mode observers at low speeds has not been improved, and the switching control scheme introduces new challenges. How to enable sliding mode observers to operate effectively at low motor speeds is an urgent problem to be solved.

[0007] Meanwhile, traditional sliding mode observers suffer from chattering. The academic and industrial communities have conducted extensive research on suppressing chattering, and existing solutions mainly include the following:

[0008] 1. Boundary Layer Method: Traditional sliding mode observers use a sign function as the switching function, which results in high-frequency components in the reconstructed quantity. A low-pass filter is needed to filter out these high-frequency components. However, the use of a low-pass filter leads to amplitude attenuation and phase delay in the reconstructed quantity, increasing the error in the output rotor position and speed information. To avoid using a low-pass filter while suppressing chattering, existing studies often use the sigmoid function, tanh function, or sat function instead of the sign function, which effectively suppresses chattering.

[0009] 2. Filter method: Using filters such as second-order generalized integrators, complex coefficient filters, and notch filters to filter the α-axis back electromotive force e. α β-axis back electromotive force e β Filtering is performed to retain the fundamental component of the back EMF while filtering out the high-frequency signal components.

[0010] 3. Nonlinear sliding surface method: High-order sliding surfaces, including super-spiral sliding surfaces, integral sliding surfaces, terminal sliding surfaces, fractional-order sliding surfaces, and non-singular terminal fast sliding surfaces, are used to replace traditional sliding surfaces. High-order sliding surfaces generally contain integral elements, which can reduce steady-state error and effectively suppress chattering.

[0011] In conclusion, the first and third schemes described above are universal and effectively suppress chattering. However, the robustness and disturbance rejection capability of the first scheme decrease after replacing the sign function. Meanwhile, the second scheme requires the frequency of the motor's electrical angular velocity; at low speeds, the accuracy of the sliding mode observer based on back EMF in estimating the electrical angular velocity is poor, and the filter's filtering effect is also poor, only showing good filtering performance at medium to high speeds. It is worth noting, however, that the magnitude of chattering is closely related to the sliding mode gain k; reducing the range of k values ​​that meet stability requirements is necessary to fundamentally suppress chattering.

[0012] In addition, phase-locked loops are often widely used in sliding mode observers based on back electromotive force due to their simple structure and excellent performance. However, this structure has problems such as a 180° estimation error when the motor reverses, poor dynamic performance, system bandwidth that varies with the permanent magnet flux linkage and the electric angular velocity of the motor, and poor disturbance rejection capability.

[0013] As can be seen from the above, the main problems of sliding mode observers based on back EMF are that they cannot be used in the low-speed range, there is a 180° estimation error when reversing, the chattering phenomenon is serious, the anti-disturbance ability is poor, and the dynamic performance is poor. Therefore, a feasible solution is urgently needed. Summary of the Invention

[0014] Purpose of the invention:

[0015] This invention provides a sensorless control system and method for an external rotor permanent magnet synchronous motor across its entire operating range. Its purpose is to solve problems such as observer failure, severe chattering, poor dynamic performance, poor anti-disturbance capability, and 180° estimation error in reversal of the sliding mode observer based on back EMF at zero low speed, and to realize a sensorless algorithm across the entire operating range.

[0016] Technical solution:

[0017] A sensorless control system for an external rotor permanent magnet synchronous motor includes a closed-loop sliding mode observer for the permanent magnet flux linkage component and a bilinear extended state observer phase-locked loop with a compensation mechanism. The bilinear extended state observer phase-locked loop with compensation mechanism consists of a fixed observer coefficient extended state observer phase-locked loop and a variable observer coefficient extended state observer phase-locked loop. The output of the sliding mode observer for the permanent magnet flux linkage component is connected to the input of both the fixed observer coefficient extended state observer phase-locked loop and the variable observer coefficient extended state observer phase-locked loop. The output of the fixed observer coefficient extended state observer phase-locked loop is connected to the input of the variable observer extended state coefficient observed phase-locked loop. The output of the variable observer coefficient extended state observer phase-locked loop is connected to the sliding mode observer for the permanent magnet flux linkage component, a Park current coordinate transformation module, and a speed PI controller.

[0018] A control method for a sensorless control system of an external rotor permanent magnet synchronous motor, wherein the direct-axis current i d Cross-axis current i q α-axis voltage u α β-axis voltage u β α-axis current i α β-axis current i β Estimated electrical angular velocity from the output of a bilinear extended state observer phase-locked loop with compensation mechanism Estimating electrical angle All inputs are fed into the sliding mode observer of the permanent magnet flux linkage component. After internal calculations within the sliding mode observer of the permanent magnet flux linkage component, the reconstructed variable -ψ is output. f sinθ e and ψ f cosθ e Reconstructing variable -ψ f sinθ e and ψ f cosθ e The input is fed into a bilinear extended state observer phase-locked loop with compensation mechanism, and the output of the bilinear extended state observer phase-locked loop with compensation mechanism is the estimated electric angular velocity of the motor. In addition, the number of pole pairs of the rotor permanent magnet synchronous motor is used to estimate the mechanical angular velocity ω. m .

[0019] Furthermore, the internal calculations of the sliding mode observer for the permanent magnet flux linkage component are as follows:

[0020] -ψ t sinθ e The α-axis voltage u output by the Clark voltage coordinate transformation module during the reconstruction process α Subtract the output of the sliding mode observer based on the permanent magnet flux linkage component and the estimated electric angular velocity from the output of the phase-locked loop of the variable observer coefficient extended state observer. The product of the absolute values, the difference, and then multiplied by the reciprocal of the direct-axis inductance, 1 / L d get Same as including tracking differentiator The reconstructed estimates of the first physical quantity are added together, and the result is obtained. After that Subtract the variable observer coefficients from the estimated electrical angular velocity output of the extended state observer phase-locked loop. The β-axis current i output by the Clark current coordinate transformation module β Multiply the product of the inductance and the cross-axis inductance, take the difference, and divide by the value of the inductance (L). d -L q ) / L d ,get The reciprocal of its quotient with the stator resistance and the direct-axis inductance - R s / L d and α-axis estimation of current To get the product by adding them together Then, by integrating the result, the α-axis estimated current is obtained. The result The α-axis current i output by the Clark current coordinate transformation module α Get by doing bad things Will The estimated permanent magnet flux linkage component -ψ is obtained by multiplying the result by the sliding mode gain k using the sigmoid function. f sinθ e ;

[0021] ψ f cosθ e During the reconstruction process, the β-axis voltage u output by the Clark voltage coordinate transformation module β Subtract the output of the sliding mode observer based on the permanent magnet flux linkage component and the estimated electric angular velocity from the phase-locked loop output of the variable observer. The product of the absolute values, the difference, and then multiplied by the reciprocal of the direct-axis inductance, 1 / L dget Same as including tracking differentiator Subtract the reconstructed estimates of the second physical quantity, and you get the result. After that Add the estimated electrical angular velocity from the output of the variable observer extended state observer phase-locked loop The α-axis current i output by the Clark current coordinate transformation module α Multiply the product of the inductance and the cross-axis inductance, take the difference, and divide by the value of the inductance (L). d -L q ) / L d ,get The reciprocal of its quotient with the stator resistance and the direct-axis inductance - R s / L d and To get the product by adding them together Then, integrating the result yields the β-axis estimated current. The result The β-axis current i output by the Clark current coordinate transformation module β Get by doing bad things Will The estimated permanent magnet flux linkage component ψ is obtained by multiplying the result by the sliding mode gain k using the sigmoid function. f cosθ e .

[0022] Furthermore, the sliding mode gain k is taken as the value of the permanent magnet flux linkage ψ. f The value is 1.5 to 2 times higher.

[0023] Furthermore, the internal operation of the phase-locked loop of the fixed observer coefficient expansion state observer is as follows: the reconstructed variable -ψ output by the sliding mode observer based on the permanent magnet flux linkage component is... f sinθ e same cos operation Multiply to get The reconstructed variable ψ from the sliding mode observer output based on the permanent magnet flux linkage component. f cosθ e same The sine operation is Multiply to get Will and Invert each and then add them together to get It is sent to the compensation mechanism switch of the double fourth-order linear extended state observer phase-locked loop, and the output of the compensation mechanism switch is divided by the permanent magnet flux linkage ψ. fThe data is then fed into a fourth-order linear extended state observer, where the observer coefficients β are... 01 β 02 β 03 β 04 Set to 4ω respectively o 6ω o 2 4ω o 3 ω o 4 , where ω o For a fixed constant, the perturbation finally estimated by the output of the phase-locked loop of the fixed observer coefficient extended state observer is sent to the input side of the phase-locked loop of the variable observer coefficient extended state observer.

[0024] The internal operation of the variable observer coefficient extended state observer phase-locked loop is as follows: the reconstructed variable -ψ output by the sliding mode observer based on the permanent magnet flux linkage component is... f sinθ e same cos operation Multiply to get The reconstructed variable ψ from the sliding mode observer output based on the permanent magnet flux linkage component. f cosθ e same The sine operation is Multiply to get Will and Invert each and then add them together to get It is sent to the compensation mechanism switch of the double fourth-order linear extended state observer phase-locked loop, and the output of the compensation mechanism switch of the variable observer coefficient extended state observer phase-locked loop is divided by the permanent magnet flux linkage ψ. f The data is then fed into a fourth-order linear extended state observer, where the observer coefficients β are... 05 β 06 β 07 β 08 They are respectively 4(a|ω m |+b), 6(a|ω m |+b) 2 、4(a|ω m |+b) 3 、(a|ω m |+b) 4 Where a and b are both constants greater than zero, the variable observer coefficients of the extended state observer phase-locked loop (17) outputs the estimated electrical angular velocity. Estimate the mechanical angular velocity ω m Estimating rotor position

[0025] Furthermore, the compensation mechanism switch uses the following method to compensate for rotor angle estimation errors: In the input quantity... Under the condition, the compensation mechanism switch output In input volume Under the condition, the compensation mechanism switch output In input volume Under the condition, the compensation mechanism switch output

[0026] A maximum torque-to-current ratio control system using a sensorless control system for an external rotor permanent magnet synchronous motor is disclosed. The output of a reference speed generator and the output of the sensorless control system are both connected to the input of a speed PI controller. The output of the speed PI controller is connected to the input of a maximum torque-to-current ratio controller. The output of the maximum torque-to-current ratio controller and the output of a Park current coordinate transformation module are both connected to the inputs of the q-axis current PI controller and the d-axis current PI controller. The outputs of the q-axis current PI controller and the d-axis current PI controller are connected to the input of an inverse Park voltage coordinate transformation module. The output of the inverse Park voltage coordinate transformation module is connected to the input of a space vector pulse width converter. The input terminal of the SVPWM modulation module is connected, the output terminal of the SVPWM module is connected to the input terminal of the inverter, the output terminal of the inverter is connected to the input terminal of the voltage / current sensor, the output terminal of the voltage / current sensor is connected to the input terminal of the Clark current coordinate transformation module, the Clark voltage coordinate transformation module and the external rotor permanent magnet motor respectively, the output terminal of the Clark current coordinate transformation module is connected to the input terminal of the Park current coordinate transformation module and the sensorless control system respectively, the output terminal of the Clark voltage coordinate transformation module is connected to the input terminal of the sensorless control system, and the Park current coordinate transformation module is connected in a closed loop with the sensorless control system.

[0027] The beneficial effects of this invention are:

[0028] 1. To address a series of issues with the sensorless control algorithm based on back EMF and combined with phase-locked loop, such as poor estimation accuracy, severe chattering, poor disturbance rejection, 180° estimation error during reversal, poor dynamic performance, and system bandwidth variation with permanent magnet flux linkage and motor angular velocity, the sliding mode observer based on permanent magnet flux linkage uses the component of permanent magnet flux linkage in the αβ coordinate system as the reconstruction quantity. This results in a relatively stable reconstruction quantity across the entire operating range, including the zero-low speed range, and provides accurate rotor position information across the entire speed range, thus overcoming the problem of poor estimation accuracy in the zero-low speed range.

[0029] 2. When the permanent magnet flux linkage component is used as the reconstruction quantity, the sliding mode gain k is reduced to 1 / ω of the sliding mode observer based on the back electromotive force. e This effectively suppresses chattering, provides a smoother permanent magnet flux component waveform, avoids the use of low-pass filters, and maintains strong robustness.

[0030] 3. The compensation mechanism switch in the dual fourth-order linear extended state observer phase-locked loop is used to overcome the problem of 180° angle estimation error in traditional phase-locked loops when the external rotor permanent magnet motor reverses. The principle of the compensation mechanism switch is: to perform angle compensation in the interval where there is a 180° estimation error, and not to perform compensation operation in the interval where the estimation error is zero, so as to overcome the problem of 180° estimation error in traditional phase-locked loop motor reversal.

[0031] 4. A dual fourth-order linear extended state observer phase-locked loop with compensation mechanism is used to extract rotor position and speed information from the flux linkage components of permanent magnets. This dual fourth-order linear extended state observer phase-locked loop with compensation mechanism consists of two parts: a fixed observer coefficient observer phase-locked loop and a variable observer coefficient observer phase-locked loop. The fixed observer coefficient observer phase-locked loop estimates the main disturbance and outputs the estimated main disturbance to the variable observer coefficient observer phase-locked loop, which is responsible for the final output of the estimated speed and estimated rotor position.

[0032] 5. This invention does not employ a switching control strategy. Instead, it uses a sliding mode observer based on permanent magnet flux throughout the entire operating range, including the zero-low speed range, thus avoiding the speed and load torque fluctuation problems caused by the switching process in existing methods. Attached Figure Description

[0033] Figure 1 This is a schematic diagram of the sensorless control structure of the external rotor permanent magnet synchronous motor of the present invention;

[0034] Figure 2 This is a schematic diagram of the sliding mode observer principle of the external rotor permanent magnet synchronous motor based on the permanent magnet flux component of the present invention;

[0035] Figure 3 This is a schematic diagram of the phase-locked loop principle of the dual fourth-order linear extended state observer with compensation mechanism of the present invention;

[0036] Figure 4 This is a schematic diagram of the compensation mechanism switch of the present invention;

[0037] Figure 5 A comparison chart showing the speed tracking of an external rotor permanent magnet synchronous motor using the sensorless control algorithm of this invention;

[0038] Figure 6A speed tracking error diagram for an external rotor permanent magnet synchronous motor using the sensorless control algorithm of this invention;

[0039] Figure 7 A comparison diagram of position tracking of an external rotor permanent magnet synchronous motor using the sensorless control algorithm of this invention;

[0040] Figure 8 The diagram shows the position tracking error of an external rotor permanent magnet synchronous motor using the sensorless control algorithm of this invention.

[0041] Explanation of reference numerals in the attached figures:

[0042] 1. Reference speed generator; 2. Speed ​​PI controller; 3. Maximum torque-to-current ratio controller; 4. q-axis current PI controller; 5. d-axis current PI controller; 6. Inverse Park voltage coordinate transformation module; 7. Space vector pulse width modulation (SVPWM) module; 8. Inverter; 9. Voltage / current sensor; 10. External rotor permanent magnet synchronous motor; 11. Clark current coordinate transformation module; 12. Clark voltage coordinate transformation module; 13. Park; 14. Sensorless control system; 15. Sliding mode observer based on permanent magnet flux linkage component; 16. Fixed observer coefficient extended state observer phase-locked loop; 17. Variable observer coefficient extended state observer phase-locked loop; 18. Bilinear extended state observer phase-locked loop with compensation mechanism; 19. First physical quantity reconstruction estimation part; 20. Second physical quantity reconstruction estimation part; 21. Compensation mechanism switch. Detailed Implementation

[0043] The present invention will be described in more detail below with reference to the accompanying drawings.

[0044] This control method mainly consists of a sliding mode observer 15 based on the permanent magnet flux linkage component in the extended back EMF in a two-phase stationary αβ coordinate system, and a bilinear extended state observer phase-locked loop 18 with a compensation mechanism. Utilizing the relatively stable characteristic of the permanent magnet flux linkage in the extended back EMF across the entire operating range, the sliding mode control algorithm reconstructs the component -ψ of the permanent magnet flux linkage in the two-phase stationary αβ coordinate system. f sinθ e and ψ f cosθ eThe reconstructed permanent magnet flux linkage component is then fed into a dual fourth-order linear extended state observer 18 with a compensation mechanism. The dual fourth-order linear extended state observer 18 comprises two parts: a fixed-observer-coefficient fourth-order extended state observer 16 and a variable-observer-coefficient fourth-order extended state observer 17. The fixed-observer-coefficient fourth-order extended state observer 16 is responsible for estimating the main disturbances and feeding these estimates into the variable-observer-coefficient fourth-order extended state observer 17. The variable-observer-coefficient fourth-order extended state observer 17 is responsible for outputting the estimated speed and rotor position information of the final external rotor permanent magnet synchronous motor 10. Its observer coefficients contain quantities that are linearly related to the motor's electrical angular velocity. In the zero-speed and low-speed stages, the observer coefficients are small, effectively suppressing chattering; in the medium- and high-speed stages, the observer coefficients are large, providing better dynamic performance and disturbance rejection performance. The phase-locked loop compensation mechanism switch 21 compensates for the angular error in the interval with a 180° estimation error, overcoming the problem of a 180° estimation error when the motor reverses. The above control strategy avoids the switching problem between the two existing sensorless algorithms, and realizes a sensorless control algorithm for the entire operating range, including the zero-low speed range.

[0045] In this invention, the letters have the following meanings: i d For direct-axis current, i q For quadrature-axis current, u α Let i be the voltage along the α-axis. α Let u be the α-axis current. β For the β-axis voltage, i β Let ψ be the β-axis current. f For permanent magnet flux linkage, ω m To estimate the mechanical angular velocity, To estimate the electric angular velocity, To estimate the electrical angle, -ψ f sinθ e ψ is the variable to be reconstructed along the α axis. f cosθ e L is the variable for reconstructing the β-axis. d For a direct-axis inductor, L q It is a quadrature axis inductor. Estimate the current for the α-axis. Estimate the current along the β axis, θ e Let ω be the electrical angle. e Let ω be the electric angular velocity. m *Reference speed, i N * For the total reference current, i d * i is the d-axis reference current. q * U is the q-axis reference current. d* U is the d-axis reference voltage. q * U is the q-axis reference voltage. abc For three-phase voltage, i abc Let k be the three-phase current and k be the sliding mode gain. This is the reconstruction estimation part for the first physical quantity. For the reconstruction estimation part of the second physical quantity, β 01 β 02 β 03 β 04 For the fixed observer coefficients, the extended state observer phase-locked loop coefficients, β 05 β 06 β 07 β 08 The variable observer coefficients are used to expand the state observer phase-locked loop coefficients.

[0046] This invention proposes a sensorless control system for the entire operating range based on the flux linkage component of a permanent magnet, as shown in the schematic diagram below. Figure 1 As shown in Figure 14, the sensorless control system 14 includes a closed-loop sliding mode observer 15 for the permanent magnet flux linkage component and a bilinear expansion state observer phase-locked loop 18 with a compensation mechanism. The bilinear expansion state observer phase-locked loop 18 with a compensation mechanism consists of a fixed observer coefficient observer phase-locked loop 16 and a variable observer coefficient observer phase-locked loop 17. The output of the sliding mode observer 15 for the permanent magnet flux linkage component is connected to the input of the fixed observer coefficient observer phase-locked loop 16 and the variable observer coefficient observer phase-locked loop 17, respectively. The output of the fixed observer coefficient observer phase-locked loop 16 is connected to the input of the variable observer coefficient observer phase-locked loop 17, and the output of the variable observer coefficient observer phase-locked loop 17 is connected to the sliding mode observer 15 for the permanent magnet flux linkage component, the Park current coordinate transformation module 13, and the speed PI controller 2, respectively.

[0047] A control method for a sensorless control system of an external rotor permanent magnet synchronous motor across its entire operating range, wherein the direct-axis current i d Cross-axis current i q α-axis voltage u α β-axis voltage u β α-axis current i α β-axis current i β The estimated electrical angular velocity output of the bilinear extended state observer phase-locked loop 18 with compensation mechanism Estimating electrical angle All inputs are fed into the sliding mode observer (5) of the permanent magnet flux linkage component, and the reconstructed variable -ψ is output from the internal calculation of the sliding mode observer (15) of the permanent magnet flux linkage component. f sinθ e and ψ fcosθ e Reconstructing variable -ψ f sinθ e and ψ f cosθ e The input is fed into a bilinear extended state observer phase-locked loop 18 with compensation mechanism, and the output of the bilinear extended state observer phase-locked loop 18 with compensation mechanism is the estimated electric angular velocity of the motor. The difference between the given speed generated at the output of the reference speed generator 1 and the given speed is input to the speed PI controller 2.

[0048] Figure 2 This is a schematic diagram of the sliding mode observer 15 based on the permanent magnet flux linkage component of the external rotor permanent magnet synchronous motor 10 of the present invention. The input variables are: the direct-axis current i output by the Park current coordinate transformation module 13. d The quadrature-axis current i output by Park current coordinate transformation module 13 q The α-axis voltage u output by Clark voltage coordinate transformation module 12 α The β-axis voltage u output by Clark voltage coordinate transformation module 12 β The α-axis current i output by Clark current coordinate transformation module 11 α The β-axis current i output by Clark current coordinate transformation module 11 β The estimated electric angular velocity output by the fourth-order linear extended state observer 17 with variable observer coefficients. The motor estimated electrical angle output by the fourth-order linear extended state observer 17 with variable observer coefficients. The output variables of the sliding mode observer 15 based on the permanent magnet flux linkage component are: -ψ f sinθ e and ψ f cosθ e Among them, the sliding mode observer 15 based on permanent magnet flux linkage selects extended back electromotive force. As reconstruction components, the extended back EMF components along the α-axis and β-axis are reconstructed respectively. and -ψ f sinθ e and ψ f cosθ e The component, and will be estimated by sliding mode observer 15 based on permanent magnet flux linkage component -ψ f sinθ e and ψ f cosθ e The component is fed into the double fourth-order linear extended state observer phase-locked loop 18 with compensation mechanism.

[0049] The specific implementation method of the sliding mode observer 15 based on the permanent magnet flux linkage component is as follows:

[0050] The internal calculations of the sliding mode observer 15 for the permanent magnet flux linkage component are as follows: using extended back electromotive force. and -ψ f sinθ e and ψ t cosθ e Components are used as reconstructed variables, and in and During the reconstruction process, a tracking differentiator (TD) is used. The rotor position and speed information required for the variable reconstruction are obtained through feedback from the variable observer coefficient observer phase-locked loop 17 in the next stage. According to Lyapunov stability judgment, the sliding mode gain k that meets the stability requirements is greater than the permanent magnet flux linkage ψ. f The value is generally the magnetic flux linkage ψ of the permanent magnet. f The sliding mode gain k is 1.5 to 2 times that of the boundary layer function. A smaller sliding mode gain k is beneficial for suppressing chattering, while a larger sliding mode gain k is beneficial for improving dynamic performance. The boundary layer function is set to the sigomoid function.

[0051] -ψ f sinθ e During the reconstruction process, the α-axis voltage u output by Clark voltage coordinate transformation module 12 α Subtract the output of the sliding mode observer 15 based on the permanent magnet flux linkage component and the estimated electric angular velocity output of the fourth-order linear extended state observer 17 with variable observer coefficients. The product of absolute values, the difference, and then multiplied by 1 / L d get Same as including tracking differentiator The first physical quantity reconstruction estimation part 19 is added together, and the result is obtained after addition. After that Subtract the estimated electric angular velocity from the output of the fourth-order linear extended state observer 17 with variable observer coefficients. The β-axis current i output by Clark current coordinate transformation module 11 β Multiply by (L) d -L q ) / L d ,get Compare it with -R s / L d and To get the product by adding them together Then, it is integrated to obtain... The result The α-axis current i output by Clark current coordinate transformation module 11 αGet by doing bad things Will The estimated permanent magnet flux linkage component -ψ is obtained by multiplying the result by the sliding mode gain k using the sigmoid function. f sinθ e ;

[0052] ψ f cosθ e During the reconstruction process, the β-axis voltage u output by Clark voltage coordinate transformation module 12 β Subtract the output of the sliding mode observer 15 based on the permanent magnet flux linkage component and the estimated electric angular velocity output of the fourth-order linear extended state observer 17 with variable observer coefficients. The product of absolute values, the difference, and then multiplied by 1 / L d get Same as including tracking differentiator Subtracting the second physical quantity reconstruction estimate from part 20, we get the result. After that Adding the estimated electric angular velocity from the output of the fourth-order linear extended state observer 17 with variable observer coefficients The α-axis current i output by Clark current coordinate transformation module 11 α Multiply by (L) d -L q ) / L d ,get Compare it with -R s / L d and To get the product by adding them together Then, it is integrated to obtain... The result The β-axis current i output by Clark current coordinate transformation module 11 β Get by doing bad things Will The estimated permanent magnet flux linkage component ψ is obtained by multiplying the result by the sliding mode gain k using the sigmoid function. f cosθ e .

[0053] The bilinear extended state observer phase-locked loop 18 with compensation mechanism includes a fourth-order linear extended state observer 16 with fixed observer coefficients and a fourth-order extended state observer phase-locked loop 17 with variable observer coefficients and compensation mechanism, and the output -ψ of the sliding mode observer 15 based on the permanent magnet flux linkage component. f sinθ e and ψ f cosθ eThe data is fed into a fixed-observer coefficient observer phase-locked loop 16 and a variable-observer coefficient fourth-order extended state observer phase-locked loop 17 with a compensation mechanism for parallel computation. The fixed-observer coefficient fourth-order linear extended state observer 16 is responsible for estimating disturbances such as main disturbances, motor parameter changes, load torque, and unmodeled errors, and sends the estimated disturbances to the variable-observer coefficient fourth-order extended state observer phase-locked loop 17 with a compensation mechanism. The variable-observer coefficient fourth-order extended state observer phase-locked loop 17 with a compensation mechanism is responsible for the final output of estimated speed and estimated rotor position information. It has good dynamic performance and anti-disturbance capability, and can provide accurate rotor position and speed information under ramp-type disturbances and ramp-given speeds. Through the above three main modules: the sliding mode observer 15 based on permanent magnet flux linkage components, the fixed-observer coefficient observer phase-locked loop 16, and the variable-observer coefficient fourth-order linear state observer phase-locked loop 17 with a compensation mechanism, a sensorless control algorithm covering the entire operating range, including the zero-low speed range, is realized.

[0054] Figure 3 This is a schematic diagram of the phase-locked loop 18 of the dual fourth-order linear extended state observer with compensation mechanism of the present invention. The input variables are: the direct-axis current i output by the Park current coordinate transformation module 13. d The quadrature-axis current i output by Park current coordinate transformation module 13 q The reconstructed variable output by the sliding mode observer 15 based on the permanent magnet flux linkage component: -ψ f sinθ e and ψ f cosθ e .

[0055] The specific implementation of the dual fourth-order linear extended state observer phase-locked loop 18 with compensation mechanism is as follows:

[0056] The dual fourth-order linear extended state observer phase-locked loop 18 with compensation mechanism mainly consists of two parts. First, the implementation method of the first part, the implementation method of the fixed observer coefficients phase-locked loop 16, is described. The input variables of the fixed observer coefficients fourth-order linear extended state observer phase-locked loop 16 include: the direct-axis current i output by the Park current coordinate transformation module 13. d The quadrature-axis current i output by Park current coordinate transformation module 13 q The reconstructed variable output by the sliding mode observer 15 based on the permanent magnet flux linkage component: -ψ f sinθ e and ψ f cosθ e First, the reconstructed variable -ψ output by the sliding mode observer 15 based on the permanent magnet flux linkage component is... f sinθ e same cos operation Multiply to get The reconstructed variable ψ output by the sliding mode observer 15 based on the permanent magnet flux linkage component f cosθ e same cos operation Multiply to get Will and Invert each and then add them together to get It is sent to the compensation mechanism switch21 of the dual fourth-order linear extended state observer phase-locked loop. Figure 4 ), where the input of the compensation mechanism switch21 satisfies Under the given conditions, the compensation mechanism outputs switch21. The input to the compensation mechanism switch21 satisfies Under the given conditions, the compensation mechanism outputs switch21. The input to the compensation mechanism switch21 satisfies Under the given conditions, the compensation mechanism outputs switch21. The output of switch 21, which is the compensation mechanism of the fourth-order linear expansion state observer phase-locked loop 16, is divided by the permanent magnet flux linkage ψ. f The data is then fed into a fourth-order linear extended state observer, where the observer coefficients β are... 01 β 02 β 03 β 04 Set to 4ω respectively o 6ω o 2 4ω o 3 ω o 4 , where ω o With fixed constants, the output of the fixed observer coefficients phase-locked loop 16 is the final estimated disturbance, which is then sent to the input side of the variable observer coefficients fourth-order linear extended state observer 17.

[0057] The second part of the dual fourth-order linear extended state observer phase-locked loop 18 with compensation mechanism is implemented as follows: the input variables of the variable observer coefficient fourth-order linear extended state observer 17 include: the direct-axis current i output by the Park current coordinate transformation module 13. d The quadrature-axis current i output by Park current coordinate transformation module 13 q The reconstructed variable output by the sliding mode observer 15 based on the permanent magnet flux linkage component: -ψ f sinθ e and ψ f cosθe The estimated perturbation is obtained from the output of the fixed-mode observer coefficients phase-locked loop 16. First, the reconstructed variable -ψ from the output of the sliding mode observer 15 based on the permanent magnet flux linkage component is... f sinθ e same cos operation Multiply to get The reconstructed variable ψ output by the sliding mode observer 15 based on the permanent magnet flux linkage component f cosθ e same cos operation Multiply to get Will and Invert each and then add them together to get The compensation mechanism of sending it to the dual fourth-order linear extended state observer phase-locked loop is switch21. Figure 4 In the compensation mechanism switch21, the input quantity satisfies Under the given conditions, the compensation mechanism outputs switch21. The input to the compensation mechanism switch21 satisfies Under the given conditions, the compensation mechanism outputs switch21. The input to the compensation mechanism switch21 satisfies Under the given conditions, the compensation mechanism outputs switch21. The compensation mechanism of the fourth-order linear extended state observer 17 with variable observer coefficients: the output of switch21 divided by the permanent magnet flux linkage ψ f The data is then fed into a fourth-order linear extended state observer, where the observer coefficients β are... 05 β 06 β 07 β 08 They are respectively 4(a|ω m |+b), 6(a|ω m |+b) 2 、4(a|ω m |+b) 3 、(a|ω m |+b) 4 Where a and b are both constants greater than zero, the fourth-order linear extended state observer 17 with variable observer coefficients outputs an estimated rotational speed. Estimate the mechanical angular velocity ω m and estimate rotor position

[0058] like Figure 1As shown, a maximum torque-to-current ratio control system using the sensorless control method of the present invention is described. The output terminals of the reference speed generator 1 and the sensorless control system 14 are both connected to the input terminal of the speed PI controller 2. The output terminal of the speed PI controller 2 is connected to the input terminal of the maximum torque-to-current ratio controller 3. The output terminal of the maximum torque-to-current ratio controller 3 and the output terminal of the Park current coordinate transformation module 13 are both connected to the input terminals of the q-axis current PI controller 4 and the d-axis current PI controller 5. The output terminals of the q-axis current PI controller 4 and the d-axis current PI controller 5 are connected to the input terminal of the inverse Park voltage coordinate transformation module 6. The output terminal of the inverse Park voltage coordinate transformation module 6 is connected to the input terminal of the space vector pulse width modulation (SVPWM) module 7. The input terminal of the inverter is connected to the input terminal of the space vector pulse width modulation (SVPWM) module 7, the output terminal of the inverter 8 is connected to the input terminal of the voltage / current sensor 9, the output terminal of the voltage / current sensor 9 is connected to the input terminal of the Clark current coordinate transformation module 11, the Clark voltage coordinate transformation module 12 and the external rotor permanent magnet motor 10, respectively, the output terminal of the Clark current coordinate transformation module 11 is connected to the input terminal of the Park current coordinate transformation module 13 and the sensorless control system 14, the output terminal of the Clark voltage coordinate transformation module 12 is connected to the input terminal of the sensorless control system 14, and the Park current coordinate transformation module 13 is connected in a closed loop with the sensorless control system 14.

[0059] The control method of the maximum torque-to-current ratio control system for the external rotor permanent magnet synchronous motor is as follows: the reference speed ω output by reference speed generator 1... m * The estimated rotational speed ω output by the sensorless control algorithm 14 m The difference is calculated and then input to the speed PI controller 2. The speed PI controller 2 outputs the total reference current i. N * i N * The input is fed into the maximum torque current ratio controller 3, and the output of the maximum torque current ratio controller 3 includes two parts: the d-axis reference current i. d * and q-axis reference current i q * d-axis reference current i d * The actual d-axis current i output by Park current coordinate transformation module 13 d The difference is subtracted and input to the d-axis current PI controller 5, and the q-axis reference current i q * The actual q-axis current i output by Park current coordinate transformation module 13 qThe difference is subtracted and input to the q-axis current PI controller 4. The d-axis current PI controller 5 and the q-axis current PI controller 4 output u respectively. d * u q * In the reverse Park voltage coordinate transformation module 6, the θ output by the sensorless control algorithm 14 is used in conjunction with the transformation. e Perform inverse Park coordinate transformation; inverse Park voltage coordinate transformation module 6 outputs u α u β The output value of the Space Vector Pulse Width Modulation (SVPWM) module 7 is sent to the inverter 8, and the output value of the inverter 8 is sent to the voltage / current sensor 9; the voltage / current sensor 9 outputs i abc The Clark current coordinate transformation module 11 outputs i α i β The outputs are respectively sent to the Park current coordinate transformation module 13 and the sensorless control algorithm 14. The Park current coordinate transformation module 13 works in conjunction with the sensorless control algorithm 14 to output θ. e Output d-axis actual current i d q-axis actual current i q And the maximum torque current ratio of the controller 3 output i d * i q * Perform differential calculations separately. Voltage / current sensor 9 outputs u. abc The output u of Clark voltage coordinate transformation module 12 is then transferred to Clark voltage coordinate transformation module 12. α u β The i output of the Park current coordinate transformation module 13 d i q and the output i of the Clark current coordinate transformation module α i β The input is fed into the sensorless control algorithm 14 for calculation; the output value of the voltage / current sensor 9 is output to the external rotor permanent magnet motor 10.

[0060] The sensorless control algorithm for external rotor permanent magnet motors provided by this invention has a wide operating range. The sliding mode gain required by the sliding mode observer based on the permanent magnet flux linkage component is usually much smaller than that of the traditional sliding mode observer based on back EMF. It effectively suppresses chattering from the bottom layer and has a smoother waveform without using a low-pass filter. Meanwhile, the bilinear extended state observer proposed in this invention has a wider operating range and stronger anti-disturbance capability. The fixed observer coefficient fourth-order linear extended state observer phase-locked loop estimates the main disturbances and sends the estimated main disturbances to the variable observer coefficient fourth-order linear extended state observer phase-locked loop. The above measures reduce the observation burden of the variable observer coefficient phase-locked loop, and can provide relatively accurate rotor position and speed information even with a smaller observer coefficient. At the same time, the variable observer coefficient setting provides better dynamic performance and anti-interference capability near the rated speed, while effectively suppressing chattering near zero low speed. Appropriately reducing dynamic performance to suppress chattering ensures that the estimated speed does not have the opposite sign to the actual speed, thus achieving good observation performance at low speeds. This significantly widens the operating range of the sensorless control algorithm and provides good tracking performance across the entire operating range.

[0061] The sensorless control system and control algorithm for the external rotor permanent magnet synchronous motor proposed in this invention were simulated in Matlab / Simulink software. The load torque of the external rotor permanent magnet synchronous motor was 30000 N·m. It was specified that the speed of the external rotor permanent magnet synchronous motor was positive when rotating forward and negative when rotating in reverse. At 0s, the given reference speed was +75 r / min. Starting from 1.5s, the given reference speed gradually decreased with a changing trend of 25 r / min. When the given reference speed reached -75 r / min, the given reference speed no longer changed. Figure 5 This is a comparison chart showing the speed tracking of an external rotor permanent magnet synchronous motor using the sensorless control algorithm of this invention. Figure 5 It can be seen that the sensorless control algorithm proposed in this invention has good speed tracking capability across the entire operating range. Figure 6 The speed tracking error diagram of the external rotor permanent magnet synchronous motor using the sensorless control algorithm of this invention is shown below. Figure 6 It can be seen that during the starting phase of the external rotor permanent magnet synchronous motor, the speed estimation error remains within the range of -0.5 r / min to 0.6 r / min, and after the start-up is completed, the speed estimation error remains within the range of -0.2 r / min to 0.2 r / min. Figure 7 This is a comparison chart of position tracking of an external rotor permanent magnet synchronous motor using the sensorless control algorithm of this invention. Figure 7 It can be seen that the sensorless control algorithm proposed in this invention has good position tracking capability across the entire operating range; Figure 8This is a position tracking error diagram of an external rotor permanent magnet synchronous motor using the sensorless control algorithm of this invention. Figure 8 It can be seen that during the startup phase, the position estimation error remains within the range of -0.01rad to 0.06rad, and after startup, the position estimation error is within the range of 0.025rad to 0.0425rad, indicating good position tracking performance.

Claims

1. A control method for a sensorless control system of an external rotor permanent magnet synchronous motor, characterized in that: The sensorless control system (14) includes a closed-loop sliding mode observer (15) for the permanent magnet flux component and a bilinear extended state observer phase-locked loop (18) with a compensation mechanism. The bilinear extended state observer phase-locked loop (18) with a compensation mechanism consists of a fixed observer coefficient fourth-order extended state observer (16) and a variable observer coefficient fourth-order extended state observer (17). The output of the sliding mode observer (15) for the permanent magnet flux component is connected to the input of the fixed observer coefficient fourth-order extended state observer (16) and the variable observer coefficient fourth-order extended state observer (17), respectively. The output of the fixed observer coefficient fourth-order extended state observer (16) is connected to the input of the variable observer coefficient fourth-order extended state observer (17), and the output of the variable observer coefficient fourth-order extended state observer (17) is connected to the sliding mode observer (15) for the permanent magnet flux component, the Park current coordinate transformation module (13), and the speed PI controller (2), respectively. The direct-axis current i d Cross-axis current i q α-axis voltage u α β-axis voltage u β α-axis current i α β-axis current i β The estimated electric angular velocity output by the bilinear extended state observer phase-locked loop (18) with compensation mechanism Estimating electrical angle All inputs are fed into the sliding mode observer (15) of the permanent magnet flux linkage component, and the reconstructed variable -ψfsinθ is output after the internal calculation of the sliding mode observer (15) of the permanent magnet flux linkage component. e and ψfcosθ e Reconstruct the variable -ψfsinθ e and ψfcosθ e The input is fed into a bilinear extended state observer phase-locked loop (18) with compensation mechanism, and the output of the bilinear extended state observer phase-locked loop (18) with compensation mechanism is the estimated electric angular velocity of the motor. In addition, the number of pole pairs of the rotor permanent magnet synchronous motor is used to estimate the mechanical angular velocity ω. m ; The internal operation of the fourth-order extended state observer (16) with constant observer coefficients is as follows: the reconstructed variable -ψfsinθ output by the sliding mode observer (15) based on the permanent magnet flux linkage component is used. e same cos operation Multiply to get The reconstructed variable ψfcosθ output by the sliding mode observer (15) based on the permanent magnet flux linkage component is... e same The sine operation is Multiply to get Will and Invert each and then add them together to get The output of the compensation mechanism switch (21) of the double fourth-order linear extended state observer phase-locked loop is divided by the permanent magnet flux linkage ψf and then sent to the fourth-order linear extended state observer. The observer coefficient β of the fourth-order linear extended state observer is... 01 β 02 β 03 β 04 Set to 4ω respectively o 6ω o 2 4ω o 3 ω o 4 , where ω o For a fixed constant, the output of the fourth-order extended state observer (16) with fixed observer coefficients is the final estimated disturbance to the input side of the fourth-order extended state observer (17) with variable observer coefficients; The internal operation of the fourth-order extended state observer (17) with variable observer coefficients is as follows: the reconstruction variable -ψfsinθ output by the sliding mode observer (15) based on the permanent magnet flux linkage component is used. e same cos operation Multiply to get The reconstructed variable ψfcosθ output by the sliding mode observer (15) based on the permanent magnet flux linkage component is... e same The sine operation is Multiply to get Will and Invert each and then add them together to get The output of the compensation mechanism switch (21) of the phase-locked loop of the double fourth-order linear extended state observer is divided by the permanent magnet flux linkage ψf and then sent to the fourth-order linear extended state observer. The observer coefficient β of the fourth-order linear extended state observer is... 05 β 06 β 07 β 08 They are respectively 4(a|ω m |+b), 6(a|ω m |+b) 2 、4(a|ω m |+b) 3 、(a|ω m |+b) 4 Where a and b are both constants greater than zero, the fourth-order extended state observer (17) with variable observer coefficients outputs an estimated electric angular velocity. Estimate the mechanical angular velocity ω m Estimating rotor position 2. The control method for a sensorless control system of an external rotor permanent magnet synchronous motor according to claim 1, characterized in that: The internal calculations of the sliding mode observer (15) for the permanent magnet flux linkage component are as follows: -ψ f sinθ e During the reconstruction process, the α-axis voltage u output by the Clark voltage coordinate transformation module (12) α Subtract the output of the sliding mode observer (15) based on the permanent magnet flux linkage component and the estimated electric angular velocity output of the fourth-order extended state observer (17) with variable observer coefficients. The product of the absolute values, the difference, and then multiplied by the reciprocal of the direct-axis inductance, 1 / L d get Same as including tracking differentiator The first physical quantity reconstruction estimation part (19) is added together, and the result is obtained. After that The estimated electric angular velocity output by subtracting the variable observer coefficients from the fourth-order extended state observer (17) The β-axis current i output by the Clark current coordinate transformation module (11) β Multiply the product of the inductance and the cross-axis inductance, take the difference, and divide by the value of the inductance (L). d -L q ) / L d ,get The reciprocal of its quotient with the stator resistance and the direct-axis inductance - R s / L d and α-axis estimation of current Adding the products together yields Then, by integrating the result, the α-axis estimated current is obtained. The result The α-axis current i output by the Clark current coordinate transformation module (11) α Get by doing bad things Will The estimated permanent magnet flux linkage component -ψfsinθ is obtained by multiplying the result by the sliding mode gain k using the sigmoid function. e ; ψ f cosθ e During the reconstruction process, the β-axis voltage u output by the Clark voltage coordinate transformation module (12) β Subtract the output of the sliding mode observer (15) based on the permanent magnet flux linkage component and the estimated electric angular velocity output of the fourth-order extended state observer (17) with variable observer coefficients. The product of the absolute values, the difference, and then multiplied by the reciprocal of the direct-axis inductance, 1 / L d get Same as including tracking differentiator Subtract the second physical quantity reconstruction estimate (20) to obtain the result. After that The estimated electric angular velocity is added to the output of the fourth-order extended state observer (17) with variable observer coefficients. The α-axis current i output by the Clark current coordinate transformation module (11) α Multiply the product of the inductance and the cross-axis inductance, take the difference, and divide by the value of the inductance (L). d -L q ) / L d ,get The reciprocal of its quotient with the stator resistance and the direct-axis inductance - R s / L d and Adding the products together yields Then, integrating the result yields the β-axis estimated current. The result The β-axis current i output by the Clark current coordinate transformation module (11) β Get by doing bad things Will The estimated permanent magnet flux linkage component ψ is obtained by multiplying the result by the sliding mode gain k using the sigmoid function. f cosθ e .

3. The control method for a sensorless control system of an external rotor permanent magnet synchronous motor according to claim 2, characterized in that: The sliding mode gain k is taken as the permanent magnet flux linkage ψ f The value is 1.5 to 2 times higher.

4. The control method for a sensorless control system of an external rotor permanent magnet synchronous motor according to claim 1, characterized in that: The compensation mechanism switch(21) uses the following method to compensate for rotor angle estimation errors: In the input quantity... Under the given conditions, the compensation mechanism switch(21) outputs In input volume Under the given conditions, the compensation mechanism switch(21) outputs In input volume Under the given conditions, the compensation mechanism switch(21) outputs 5. A maximum torque-to-current ratio control system using the control method of the sensorless control system for an external rotor permanent magnet synchronous motor according to claim 1, characterized in that: The output of the reference speed generator (1) and the output of the sensorless control system 14 are both connected to the input of the speed PI controller (2). The output of the speed PI controller (2) is connected to the input of the maximum torque current ratio controller (3). The output of the maximum torque current ratio controller (3) and the output of the Park current coordinate transformation module (13) are both connected to the inputs of the q-axis current PI controller (4) and the d-axis current PI controller (5). The outputs of the q-axis current PI controller (4) and the d-axis current PI controller (5) are connected to the input of the inverse Park voltage coordinate transformation module (6). The output of the inverse Park voltage coordinate transformation module (6) is connected to the input of the space vector pulse width modulation (SVPWM) module (7). The output of the WM module (7) is connected to the input of the inverter (8). The output of the inverter (8) is connected to the input of the voltage / current sensor (9). The output of the voltage / current sensor (9) is connected to the input of the Clark current coordinate transformation module (11), the Clark voltage coordinate transformation module (12), and the external rotor permanent magnet motor (10), respectively. The output of the Clark current coordinate transformation module (11) is connected to the input of the Park current coordinate transformation module (13) and the sensorless control system (14), respectively. The output of the Clark voltage coordinate transformation module (12) is connected to the input of the sensorless control system (14). The Park current coordinate transformation module (13) and the sensorless control system (14) are connected in a closed loop.

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