A full-bridge converter with zero voltage and zero current switching

Abstract

The invention discloses a full-bridge converter with zero voltage and zero current switching, belonging to the technical field of power electronics. In this scheme, the transformer primary side adopts a full-bridge structure composed of four MOS tubes, and the secondary side adopts a coupled inductor structure to suppress the output current ripple and can greatly reduce the capacity and volume of the output filter capacitor; in an ideal state, the output current ripple is zero and the output current ripple suppression effect is not affected by the duty cycle change; by adding a switch tube on the secondary side and controlling its working sequence, the leading arm of the converter can work in a zero voltage switch (ZVS) state, and the lagging arm can work in a zero current switch (ZCS) state, thereby solving the problems of difficulty in realizing soft switching of the lagging arm of the traditional zero voltage switch phase-shifted full-bridge converter and the existence of circulating current on the primary side.

Description

Technical Field

[0001] The present invention relates to the technical field of power electronics, and more specifically to a full-bridge converter with zero voltage and zero current switching. Background Art

[0002] At present, most of the high-precision power supplies are linear power supplies. The advantage of linear power supplies is that it is easy to achieve low output current ripple. However, linear power supplies will inevitably have problems such as low efficiency, heat dissipation difficulties, and low power density. These defects will become more obvious as the power level of the power supply increases. Therefore, research on high-power, high-precision switching power supplies is very meaningful. At present, the use of an output-side staggered parallel structure in high-power switching power supplies is a conventional method to reduce output current ripple. In addition, the use of a current doubler rectifier circuit can also reduce output current ripple. The above two methods can effectively reduce output current ripple when corresponding to a certain fixed duty cycle. However, the duty cycle of a switching power supply with adjustable output voltage cannot be fixed. In most output voltage ranges, its output current ripple suppression effect is poor. Therefore, it is of great significance to develop a converter that can effectively reduce output current ripple and whose output current ripple suppression effect is not affected by duty cycle changes.

[0003] In high-power DC / DC switching power supplies, phase-shifted full-bridge converters are widely used due to their simple soft switching conditions and high efficiency. Conventional phase-shifted full-bridge converters use the parasitic capacitance of MOS tubes combined with phase-shift control methods to achieve zero voltage switching (ZVS) of the leading arm and the lagging arm. However, conventional phase-shifted full-bridge converters have the following disadvantages:

[0004] 1. It is difficult to achieve soft switching in the existing lagging arm when the load current is small;

[0005] 2. The existing primary side has circulating current in the freewheeling stage, which will reduce efficiency;

[0006] 3. The existing system has the problem of duty cycle loss.

[0007] Therefore, how to effectively solve the above problems of the phase-shifted full-bridge converter is of great significance. Summary of the invention

[0008] The present invention aims to reduce the output current ripple of the phase-shifted full-bridge converter and improve the above three disadvantages of the converter, and proposes a full-bridge converter with zero voltage and zero current switching. The technical solution adopted is as follows:

[0009] A zero voltage and zero current switching full bridge converter includes four primary side switch tubes Q a -Q d , secondary switch tube Q, primary MOS tube parasitic body diode D a -D d、Primary MOS tube parasitic capacitance C a -C d , secondary MOS tube parasitic diode D, capacitor C 1 , capacitor C 2 , output filter capacitor C, load resistor R, transformer primary series resonant inductor L r , coupled inductor L 1a , coupled inductor L 1b 、Inductance L 2 , characterized in that: the primary side of the transformer adopts a full-bridge structure, the primary-to-secondary turns ratio of the transformer is n, the secondary side of the transformer adopts a full-wave rectification method, and the center tap of the transformer is connected to two coupled inductors L 1a and L 1b The same-named ends of the two coupled inductors L 1a and L 1b The turns ratio is N 1 :N 2 , the two coupled inductors L 1a and L 1b The same-named ends are electrically connected.

[0010] Furthermore, the inductor L 2 With coupled inductor L 1b in series, the capacitor C 2 One end is connected to L 2 and the other end is connected to ground, the capacitor C 1 One end is connected to two coupled inductors L 1a and L 1b The same-name end of the capacitor C 1 The other end is connected to the secondary switch tube Q, and the secondary switch tube Q is connected to the capacitor C 1 and capacitor C 2 The secondary side switch tube Q has an anti-parallel body diode D.

[0011] Furthermore, the upper and lower taps of the transformer secondary side are respectively connected to the rectifier diode D 1 and D 2 , the output filter capacitor C and the load resistor R are connected in parallel and are located at the end of the circuit.

[0012] Furthermore, the primary switch tube Q a -Q d MOS tube is used as the switch tube.

[0013] Furthermore, the control method of the full-bridge converter is as follows: the primary switch tube Q a -Q d Using the phase shift control method, the primary switch tube Q a And the primary switch tube Qb As the leading arm, the primary switch tube Q c And the primary switch tube Q d as the lagging arm.

[0014] Furthermore, the primary switch tube Q a And the primary switch tube Q b The driving waveform is a complementary square wave signal with a duty cycle of 50%. c And the primary switch tube Q d The driving waveform is also a complementary square wave signal with a duty cycle of 50%.

[0015] Furthermore, the primary switch tube Q a -Q d There is a dead zone between the two driving signals on the same bridge arm. a The driving signal phase is ahead of the primary switch tube Q d Further, the secondary switch tube Q and the primary switch tube Q a or Q b The conduction time is the same, after the primary current of the transformer drops to zero and the primary switch tube Q c and Q d Before shutting down, turn off the secondary switch tube Q, and its switching frequency is the primary switch tube Q a -Q d twice as much.

[0016] Compared with the prior art, the advantages of the present invention are:

[0017] (1) The zero voltage and zero current switching full-bridge converter proposed in the present invention adopts a coupled inductor structure on the output side, which can effectively reduce the output current ripple and greatly reduce the output filter capacitance.

[0018] (2) It can make the primary switch tube Q a -Q b Working in zero voltage switching (ZVS) state, the primary switch tube Q c -Q d Working in zero current switching (ZCS) state, the primary switch tube Q c -Q d The problem of difficulty in achieving ZVS.

[0019] (3) When entering the freewheeling state, the primary current flows through the secondary capacitor C 1 and C 2 Under the action of the upper voltage, it quickly drops to zero, thus avoiding the problem of circulating current on the primary side that reduces efficiency.

[0020] (4) During freewheeling, the secondary capacitor C 1 and C2 It will bear the secondary side voltage when the secondary side switch tube Q is turned on, compensating for the duty cycle loss problem to a certain extent. BRIEF DESCRIPTION OF THE DRAWINGS

[0021] Figure 1 The full-bridge converter topology proposed by the present invention;

[0022] Figure 2 The control timing and main waveform diagram of the converter;

[0023] Figure 3 It is a two-port network equivalent to the output filter circuit;

[0024] Figure 4 It is the various working mode diagrams of the converter;

[0025] Figure 5 is the equivalent circuit diagram of the converter;

[0026] Figure 6 It is the main voltage and current simulation waveform when the converter is working;

[0027] Figure 7 Q is the primary switch tube a -Q b Realize zero voltage turn-on simulation waveform;

[0028] Figure 8 Q is the primary switch tube c -Q d Realize zero current shutdown simulation waveform;

[0029] Fig. 9 is the current i L1a ,i L1b ,i L1a 'Simulation waveform. DETAILED DESCRIPTION

[0030] The following will combine the drawings in the embodiments of the present invention to clearly and completely describe the technical solutions in the embodiments of the present invention. Obviously, the described embodiments are only part of the embodiments of the present invention, not all the embodiments. All other embodiments obtained by ordinary technicians in this field without creative work based on the embodiments of the present invention belong to the scope of protection of the present invention.

[0031] In the description of the present invention, it should be noted that the terms "upper", "lower", "inner", "outer", "top / bottom" and the like indicate positions or positional relationships based on the positions or positional relationships shown in the drawings, and are only for the convenience of describing the present invention and simplifying the description, rather than indicating or implying that the device or element referred to must have a specific position, be constructed and operated in a specific position, and therefore cannot be understood as limiting the present invention. In addition, the terms "first" and "second" are used for descriptive purposes only and cannot be understood as indicating or implying relative importance.

[0032] In the description of the present invention, it should be noted that, unless otherwise clearly specified and limited, the terms "installed", "provided with", "mounted / connected", "connected", etc. should be understood in a broad sense. For example, "connected" can be a fixed connection, a detachable connection, or an integral connection; it can be a mechanical connection or an electrical connection; it can be a direct connection, or it can be indirectly connected through an intermediate medium, or it can be the internal communication of two components. For ordinary technicians in this field, the specific meanings of the above terms in the present invention can be understood according to specific circumstances.

[0033] Embodiment 1:

[0034] A zero voltage and zero current switching full bridge converter includes four primary side switch tubes Q a -Q d , secondary switch tube Q, primary MOS tube parasitic body diode D a -D d 、Primary MOS tube parasitic capacitance C a -C d , secondary MOS tube parasitic diode D, capacitor C 1 , capacitor C 2 , output filter capacitor C, load resistor R, transformer primary series resonant inductor L r , coupled inductor L 1a , coupled inductor L 1b 、Inductance L 2 , characterized in that: the primary side of the transformer adopts a full-bridge structure, the primary-to-secondary turns ratio of the transformer is n, the secondary side of the transformer adopts a full-wave rectification method, and the center tap of the transformer is connected to two coupled inductors L 1a and L 1b The same-name end of the two coupled inductors L 1a and L 1b The turns ratio is N 1 :N 2 , two coupled inductors L 1a and L 1b The same-named ends are electrically connected.

[0035] Furthermore, the inductor L 2 With coupled inductor L1b The capacitor C is connected in series 2 One end is connected to L 2 and the other end is connected to ground, capacitor C 1 One end is connected to two coupled inductors L 1a and L 1b The same-name end, capacitor C 1 The other end is connected to the secondary switch tube Q, and the secondary switch tube Q is connected to the capacitor C 1 and capacitor C 2 Connected, the secondary side switch tube Q has an anti-parallel body diode D.

[0036] Furthermore, the upper and lower taps of the transformer secondary are connected to the rectifier diode D 1 and D 2 , the output filter capacitor C and the load resistor R are connected in parallel and are located at the end of the circuit.

[0037] Furthermore, the primary switch tube Q a -Q d MOS tube is used as the switch tube.

[0038] Furthermore, the control method of the full-bridge converter is as follows: the primary switch tube Q a -Q d Using the phase shift control method, the primary switch tube Q a And the primary switch tube Q b As the super forearm, the primary switch tube Q c And the primary switch tube Q d as the lagging arm.

[0039] Furthermore, the primary switch tube Q a And the primary switch tube Q b The driving waveform is a complementary square wave signal with a duty cycle of 50%, and the primary switch tube Q c And the primary switch tube Q d The driving waveform is also a complementary square wave signal with a duty cycle of 50%.

[0040] Furthermore, the primary switch tube Q a -Q d There is a dead zone between the two driving signals on the same bridge arm. a The driving signal phase is ahead of the primary switch tube Q d The driving signal.

[0041] Furthermore, the secondary switch tube Q and the primary switch tube Q a or Q b The conduction time is the same, after the primary current of the transformer drops to zero and the primary switch tube Q c and Q dBefore shutting down, turn off the secondary switch tube Q, and its switching frequency is the primary switch tube Q a -Q d twice as much.

[0042] Embodiment 2:

[0043] The present invention will be further described in detail below in conjunction with the accompanying drawings.

[0044] The converter topology proposed by the present invention is as follows: Figure 1 As shown, the specific implementation method mainly includes two parts: an output current ripple suppression method and a full-bridge converter ZVZCS implementation method; specifically includes the following steps:

[0045] Before the analysis, the following assumptions are made: the output voltage is considered constant, and the voltage is V o ; Primary MOS tube Q a -Q d Considering the parasitic body diode and parasitic capacitance, the secondary MOS tube Q only considers the parasitic body diode, and all other devices including the transformer, secondary rectifier diode, inductor and capacitor of the filter circuit and primary resonant inductor are ideal devices; the primary MOS tube parasitic capacitance C a and C b Same size, C c and C d The same size, let C a =C b =C lead , C c =C d =C lag ; The transformer is an ideal transformer and the primary-to-secondary ratio is n; the coupling inductor L 1a and L 1b The turns ratio is N 1 :N 2 , coupling coefficient is k, mutual inductance is L M ; Primary switch tube Q a and Q b The bridge arm where it is located is called the super-forward arm, and the primary switch tube Q c and Q d The bridge arm where it is located is called the lagging arm, and the primary switch tube Q a and Q b The driving waveform is a complementary square wave signal with a duty cycle of 50%. c and Q d The driving waveform is also a complementary square wave signal with a duty cycle of 50%. There is a dead zone between the two driving signals on the same bridge arm. a The driving signal phase is ahead of the primary switch tube Q dThe driving signal can adjust the output voltage by adjusting the phase shift angle; the primary resonant inductor L r Much smaller than the secondary coupling inductance L 1a ; Capacitor C 2 The capacitance value is much larger than the capacitance C 1 The capacitance value, capacitance C 2 The voltage on is constant.

[0046] Example 3: Analysis of output current ripple suppression method

[0047] The secondary output current ripple suppression part is composed of the coupled inductor L 1a and L 1b 、Inductance L 2 And the capacitor C 2 In order to improve the attenuation capability of high-frequency harmonics, a filter capacitor C is added at the output end, but this capacitor does not play a major role in suppressing current ripple; it can be composed of a coupled inductor L 1a and L 1b 、Inductance L 2 And the capacitor C 2 A two-port network is formed as Figure 3 As shown in (a), V o is the output voltage, V cd is the voltage between the same-name end of the coupled inductor and the ground. If the voltage of V cd The influence of V cd is a square wave; in the actual circuit, the coupling inductor has a coupling coefficient. Let the inductor L 1a and inductor L 1b The turns ratio is N 1 :N 2 , assuming the coupling coefficient is k, the mutual inductance can be calculated as You can Figure 3 The two-port network shown in (a) is equivalent to Figure 3 (b), that is, using an ideal transformer (turn ratio N 1 :N 2 ) and mutual inductance L M To replace Figure 3 (a) in the coupled inductor; then the mutual inductance L M Depend on Figure 3 The primary side of the ideal transformer in (b) is equivalent to the secondary side as follows Figure 3 (c) shows; the equivalent mutual inductance is L M ' indicates that, and

[0048] Combine the following Figure 3 (c) Analyze the current suppression principle and determine the inductance L 2 The value of; first assume that the current i L1aThe ripple is zero, and then according to the assumption, the current i L1a The conditions that need to be met for the ripple to be zero; if the current i L1a The ripple is zero, then Figure 3 The current ripple on the secondary winding of the ideal transformer in (c) is also zero, that is, there is no AC current; so the current i L1b The AC current flows through the inductor L 2 and L M ', based on the above analysis combined Figure 3 (c) The following equation can be listed:

[0049]

[0050]

[0051] According to formula (1) and formula (2), we can deduce L 2 The values ​​are:

[0052]

[0053] Assume that the capacitor C 2 If the capacitor voltage is large enough, it will be a constant value, and due to the inductance i L1a 、i L1b And the average value of the voltage across L2 in steady state is zero, so the capacitor C 2 The voltage value is the output voltage Vo. At this time, according to Figure 3 (c) Write the KVL equation for the two ports to get V 1 =V 2 , formula (3) can be further simplified as:

[0054]

[0055] According to the above analysis, we can get that if L 2 Satisfying equation (4) can achieve inductance L 1a The current ripple on the L1a ,i L1b ,i L1a Waveform like Figure 2 As shown, L1b The average current is zero, and the current i L1b and current i L1a The current pulsation on the inductor has opposite changing trends and can cancel each other out. The output current can be obtained by adding the two together. The output current is the inductor L 1a The current i L1aBecause the current ripple has become very small after passing through the coupled inductor filtering network (ideally, the current ripple is even zero), the value of the output filter capacitor C can be set smaller. The conventional filter capacitor C is about several hundred μF, but a small capacitor of tens of μF can be used in the converter proposed in the present invention.

[0056] It is worth noting that Figure 2 t in 4 -t 6 During this period, the MOS tube Q is turned on, causing the capacitor C 1 and C 2 The transformer secondary voltage is taken on, so it will cause t 3 -t 6 During the period L1b and i L1a 'A small fluctuation appears on the waveform; however, the changing trends of the two small fluctuations are still opposite and can offset each other; the specific reasons for the fluctuations will be explained in Example 3.

[0057] Example 4: Process Analysis of Full-Bridge Converter Implementing ZVZCS

[0058] The control timing and working waveform of the converter are as follows: Figure 2 As shown, the converter has 18 working modes in one working cycle. Since the positive and negative half cycles are symmetrical, only the first 9 working modes are analyzed here.

[0059] (1) Mode 1(t 0 -t 1 ), the primary switch tube Q a and Q d The diode D 1 Conducting diode D 2 As of now, the body diode D of MOS tube Q is turned on; at this time, the working process of the circuit is as follows Figure 4 (a) As shown in the previous analysis of the current ripple suppression method, the capacitor C 2 The voltage value is considered to be a constant value, and the voltage magnitude is the output voltage V o ; In addition, due to the inductance L 1a Much larger than the resonant inductance L r , and after the current ripple is filtered by the coupled inductor, the output current ripple is ideally 0, so the output current is considered to be a constant value I o ;according to Figure 4 (a) The equivalent circuit diagram can be drawn as follows Figure 5 As shown in (a), it can be found that the capacitance C 1 and the primary resonant inductor L r Resonance occurs; according to the equivalent circuit diagram, the primary current i in this mode can be obtained pand capacitor C 1 Voltage across the terminals v C1 as follows:

[0060]

[0061]

[0062] In the formula, n is the turns ratio between the primary and secondary sides of the transformer, V in is the input voltage, v C1 is the capacitance C 1 The voltage across the terminals, i p (t 0 ) indicates t 0 Moment i p The value of v C1 (t 0 ) indicates t 0 Time v C1 The value of .

[0063] At this time, the inductance L 1a The voltage across the two ends is the transformer secondary voltage V cd With output voltage V o The difference, V cd -V o According to the working principle of coupled inductors, the inductance L 1b The voltage across the terminals is N 2 (V s -V o ) / N 1 , if the inductance L 2 The voltage at both ends and the current flowing through take the associated reference direction, then the inductance L 2 Voltage value across the two ends v L2 and the current flowing through the inductor L 2 The current i L1b as follows:

[0064]

[0065]

[0066] In the formula, i L1b (t 0 ) indicates t 0 Moment i L1b The value of .

[0067] Depend on Figure 4 (a) It can be seen that i L1a =i L1a '+i L1b , according to the coupled inductor voltage-current relationship:

[0068]

[0069]

[0070] Then we can get i L1a 'for:

[0071]

[0072] In the formula, i L1a (t 0 ) indicates t 0 Moment i L1a The value of .

[0073] (2) Mode 2 (t 1 -t 2 ), at t 1 At this moment, the capacitor C 1 and resonant inductor L r The resonance process ends, and the body diode D of the MOS tube Q is reversely cut off; at this time, the working process of the circuit is as follows Figure 4 (b) In this mode, the current flowing through the inductor L 1a , L 1b , L 2 The variation of current and t 0 -t 1 Exactly the same.

[0074] (3) Mode 3 (t 2 -t 3 ), at t 2 At this moment, the primary switch tube Q a Turn off; at this time the circuit works as follows Figure 4 (c) shows the equivalent circuit diagram. Figure 5 (b) As shown; due to Q a Parallel capacitor C a The voltage across the two ends cannot change suddenly, so Q a It is zero voltage shutdown; in this mode, energy is transferred from the primary side to the secondary side, so the secondary side filter inductor can be equivalent to the primary side, plus the capacitor C a and C b Very small, so t 2 -t 3 The time is very short, so it can be considered that the primary current i p The size remains unchanged, so the capacitor C a and C b The voltage across the two ends can be considered to change linearly; the capacitor C a , C b The voltage across the two ends can be expressed as:

[0075]

[0076]

[0077] t 3 Moment, v ca (t 3 )=V in , v cb (t 3 )=0. Substituting into (13) we get:

[0078]

[0079] Where i p (t 2 ) indicates t 2 Moment i p The value of .

[0080] (4) Mode 4 (t 3 -t 4 ), t 3 At this moment, the capacitor C b The discharge is completed, the diode D b Natural conduction, switch tube Q b The voltage at both ends is zero, creating conditions for zero voltage switching. At this time, the primary and secondary voltages on the transformer are both 0. 3 -t 4 During this period, the primary current i p By D b and Q d The circuit works as follows: Figure 4 (d) is shown. Inductance L 1a The voltage across the two ends is -V o , inductance L 1b The voltage across the two ends is -N 2 V o / N 1 , inductance L 2 Voltage value across the two ends v L2 and the current flowing through the inductor L 2 The current i L1b as follows:

[0081]

[0082]

[0083] According to the voltage-current relationship of the coupled inductor, we can get i L1a 'for:

[0084]

[0085] In the formula, i L1b (t 3 ) indicates bar t 3Moment i L1b The value of i L1a '(t 3 ) indicates bar t 3 Moment i L1a ' value.

[0086] (5) Mode 5(t 4 -t 5 ), t 4 At this moment, the switch tube Q b Zero voltage is turned on, and at the same time, the secondary switch tube Q is turned on; the circuit working process is as follows Figure 4 (e) shows the equivalent circuit diagram. Figure 5 (c) as shown. Capacitor C 1 With the resonant inductor L r Resonance occurs, and the primary current drops rapidly. 5 The primary current drops to zero at the moment. p and capacitor C 1 Voltage across the terminals v C1 as follows:

[0087]

[0088]

[0089] In the formula, i p (t 4 ) indicates t 4 Moment i p The value of v C1 (t 4 ) indicates t 4 Time v C1 The value of .

[0090] (6) Mode 6(t 5 -t 6 ), in this mode, the primary current remains at 0, and the load current is supplied by the capacitor C 1 and C 2 Provided, capacitor C 1 The voltage of diode D decreases linearly. 1 , D 2 The circuit works as follows: Figure 4 (f) shows that the equivalent circuit is Figure 5 (d) as shown.

[0091] (7) Mode 7(t 6 -t 7 ), t 6 At this moment, the switch tube Q is turned off, and the secondary rectifier diode D 1 , D 2 At the same time, the freewheeling current is turned on, and the circuit works as follows Figure 4 (g) as shown.

[0092] (8) Mode 8(t 7 -t 8 ), t 7 Time switch tube Q d Zero current shutdown, after dead time t 7 -t 8 , t 8 Time switch tube Q c Zero current switching; the circuit working process is as follows Figure 4 (h) shown.

[0093] (9) Mode 9(t 8 -t 9 ), at t 8 -t 9 During this time period, the primary current increases rapidly, but since the primary current is not enough to match the load current, the secondary rectifier diode D 1 , D 2 Continue to conduct the freewheeling current at the same time, the equivalent circuit diagram is as follows Figure 4 (i) As shown; during this period V ab Equal to the input voltage V in However, since the two rectifier diodes are turned on at the same time, the transformer secondary voltage V cd is zero, so there will be a duty cycle loss, but compared with the traditional ZVS phase-shifted full-bridge converter, the duty cycle loss is smaller, because the traditional phase-shifted full-bridge converter needs to reduce the primary current from the forward direction to zero and then increase it in the reverse direction at this stage, but the improved converter has a primary current of zero at the beginning of this stage, and the primary current can directly increase in the reverse direction during this stage; and at t 4 -t 6 During the period, the transformer secondary voltage is v C1 +V o , the transformer secondary voltage is C 1 and C 2 To some extent, this makes up for the 8 -t 9 The duty cycle of the time period is lost.

[0094] In t 4 -t 6 Inductance L during the period 1a 、Inductance L 1b And the inductor L 2 The variation law of the upper current and t 0 -t 1 Same time period; t 6 -t 9 Inductance L during the period 1a 、Inductance L 1b And the inductor L2 The variation law of the upper current and t 3 -t 4 The time period is the same; based on the above analysis of each mode, it can be found that at t 3 -t 6 Current i in the time period L1b ,i L1a 'A small groove will appear on the

[0095] Embodiment 5:

[0096] In order to further illustrate the correctness and feasibility of the present invention, a simulation is performed on the converter of the present invention in combination with a specific example. The simulation parameters in this example are: DC input voltage V in The voltage is 400V, the power level is 1.5kW, and the switching frequency is 200kHz. Capacitor C 1 0.2μF, C 2 4.7μF, L 1a 40μH, L 1b 10μH, L 2 The output filter capacitor C is 5μH and 68μF.

[0097] Figure 6 The figure shows the driving signal, transformer primary and secondary voltages, transformer primary current, and capacitor C 1 The voltage across the two ends and the current through C 1 The current simulation waveform of the transformer primary current is higher than the leading arm switch tube Q. a or Q b After it is turned on, it drops rapidly to 0, thus eliminating the circulating current on the primary side of the transformer. From the voltage waveform on the secondary side of the transformer, it can be seen that during the period when the switch tube Q is turned on, the capacitor C 1 and C 2 It takes on the secondary voltage of the transformer, thereby reducing the degree of duty cycle loss.

[0098] Figure 7 In order to achieve the zero voltage turn-on waveform of the leading arm, the voltage across the switch tube has dropped to 0 before the driving signal arrives.

[0099] Figure 8 This is the lagging arm zero current shutdown waveform. It can be seen that the current flowing through the switch tube is 0 before the lagging arm switch shutdown signal arrives. The use of ZVZCS solves the problem of difficulty in achieving soft switching in the lagging arm of the conventional ZVS phase-shifted full-bridge converter.

[0100] Fig. 9 Shown is i L1a ,i L1b ,i L1a 'The waveform, i L1b ,i L1a'The waveform pulsation suppresses each other, and the addition of the two can get i L1a , that is, the output current; it can be seen that the output current ripple is almost zero. This proves that the present invention has the advantage of extremely low output current ripple.

[0101] In summary, this example proves that the present invention can well realize the leading arm ZVS and lagging arm ZCS, thereby solving the problem of difficulty in realizing soft switching of the lagging arm and the problem of circulating current in the primary side of the transformer, and when the secondary side switch tube Q is turned on, C 1 and C 2 The voltage of the secondary side is taken up, which makes up for the duty cycle loss problem to a certain extent. At the same time, this example also proves that the invented converter has extremely low output current ripple and has good application prospects in the field of high-power and high-precision switching power supplies.

[0102] The above is only a preferred specific implementation manner of the present invention; but the protection scope of the present invention is not limited thereto; any technician familiar with the technical field within the technical scope disclosed by the present invention; any equivalent replacement or change based on the technical solution and improved concept of the present invention shall be covered by the protection scope of the present invention.

Claims

1. A full-bridge converter with zero voltage and zero current switching, comprising four primary switch tubes Qa-Qd, a secondary switch tube Q, a primary MOS tube parasitic body diode Da-Dd, a primary MOS tube parasitic capacitor Ca-Cd, a secondary MOS tube parasitic body diode D, a capacitor C1, a capacitor C2, an output filter capacitor C, a load resistor R, a transformer primary series resonant inductor Lr, a coupled inductor L1a, a coupled inductor L1b, and an inductor L2, characterized in that: The primary side of the transformer adopts a full-bridge structure, the turns ratio of the primary side to the secondary side of the transformer is n, the secondary side of the transformer adopts a full-wave rectification mode, the center tap of the transformer is connected to the same-name ends of the two coupled inductors L1a and L1b, the turns ratio of the two coupled inductors L1a and L1b is N1:N2, and the same-name ends of the two coupled inductors L1a and L1b are electrically connected; One end of the inductor L2 is connected in series with the opposite-name end of the coupling inductor L1b, one end of the capacitor C2 is connected to the other end of L2, and the other end of C2 is connected to the ground, one end of the capacitor C1 is connected to the same-name ends of the two coupling inductors L1a and L1b, the other end of the capacitor C1 is connected to the source of the secondary switch tube Q, the drain of the secondary switch tube Q is connected to one end of the capacitor C2, the secondary switch tube Q has an anti-parallel body diode D, and the opposite-name end of L1a is connected to the output positive end; The upper and lower taps of the secondary side of the transformer are connected to the cathode of the rectifier diode D1 and the cathode of D2 respectively. The output filter capacitor C and the load resistor R are connected in parallel and are located at the end of the circuit.

2. A zero voltage and zero current switching full bridge converter according to claim 1, characterized in that: The primary switch tubes Qa-Qd use MOS tubes as switch tubes.

3. The zero voltage and zero current switching full-bridge converter according to claim 1, characterized in that: The control method of the full-bridge converter is a control method in which the primary switch tubes Qa-Qd adopt a phase shift, the primary switch tubes Qa and Qb serve as leading arms, and the primary switch tubes Qc and Qd serve as lagging arms.

4. The zero voltage and zero current switching full-bridge converter according to claim 1, characterized in that: The driving waveforms of the primary switch tubes Qa and Qb are complementary square wave signals with a duty cycle of 50%, and the driving waveforms of the primary switch tubes Qc and Qd are also complementary square wave signals with a duty cycle of 50%.

5. The zero voltage and zero current switching full-bridge converter according to claim 1, characterized in that: There is a dead zone between the two driving signals on the same bridge arm of the primary switch tube Qa-Qd, and the phase of the driving signal of the primary switch tube Qa is ahead of the driving signal of the primary switch tube Qd.

6. A zero voltage and zero current switching full bridge converter according to claim 1, characterized in that: The secondary switch tube Q is turned on at the same time as the primary switch tube Qa or Qb. The secondary switch tube Q is turned off after the primary current of the transformer drops to zero and before the primary switch tubes Qc and Qd are turned off. Its switching frequency is twice that of the primary switch tubes Qa-Qd.

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