Transformer-based high power density resonant boost unit for offshore wind power

Abstract

The present invention discloses a transformer-based offshore wind power high power density resonant boost unit, comprising a resonant capacitor, a transformer, a first diode, a second diode, a first capacitor and a second capacitor; the first diode, the second diode, the first capacitor and the second capacitor constitute a rectifier bridge, the anode of the first diode is connected to the cathode of the second diode, the cathode of the first diode is connected to the positive electrode of the first capacitor, the anode of the second diode is connected to the negative electrode of the second capacitor, the negative electrode of the first capacitor is connected to the positive electrode of the second capacitor, one end of the resonant capacitor is connected to one end of the primary winding of the transformer, and the secondary winding of the transformer is respectively connected to the midpoint of the diode bridge arm and the midpoint of the capacitor bridge arm of the rectifier bridge. Compared with the traditional voltage doubling rectifier unit, the present invention has the advantages of higher power upper limit, more stable gain characteristics, and more suitable for high-frequency applications under the same application scenario.

Description

Technical Field

[0001] The present invention relates to the technical field of offshore wind power boost conversion, and in particular to a transformer-based offshore wind power high power density resonant boost unit. Background Art

[0002] As the scale of offshore wind farm operation increases, the DC collection method has more development advantages and application markets than the traditional AC collection method, because it does not have the problem of multi-machine synchronization, does not require a large and bulky power frequency transformer, and has only two power transmission lines. In order to speed up the construction of offshore wind power, it is necessary to reduce the cost and difficulty of system transportation, installation, and maintenance. Therefore, it is necessary to further reduce the weight and volume of the system, and accordingly increase the operating switching frequency of the converter. Since the voltage gain of the traditional (Pulse Width Modulation PWM) type transformer is coupled with power and switching frequency, as the switching frequency increases, the power operating upper limit of the isolation transformer will decrease accordingly, thereby reducing the gain size at the same power level. In order to achieve high-frequency and lightweight systems while further improving power density, it is necessary to model and analyze the power transmission mechanism of high-frequency transformers and quantify the coupling constraints of power, gain, and frequency, and optimize and improve the boost topology accordingly to make it suitable for high-frequency applications. Summary of the invention

[0003] The purpose of the present invention is to address the defects of transformers in high-frequency unidirectional boosting applications, such as high coupling of power and gain and low upper power operating limit, and to propose a transformer-based offshore wind power high-power density resonant boosting unit with precise design of resonant capacitor capacitance, stable voltage gain and higher power upper limit.

[0004] To achieve the above-mentioned purpose, the technical solution provided by the present invention is: a transformer-based offshore wind power high power density resonant boost unit, comprising a resonant capacitor, a transformer, a first diode, a second diode, a first capacitor and a second capacitor; the first diode, the second diode, the first capacitor and the second capacitor constitute a rectifier bridge, wherein the anode of the first diode is connected to the cathode of the second diode, the cathode of the first diode is connected to the positive electrode of the first capacitor, the anode of the second diode is connected to the negative electrode of the second capacitor, the negative electrode of the first capacitor is connected to the positive electrode of the second capacitor, one end of the resonant capacitor is connected to one end of the primary winding of the transformer, and the secondary winding of the transformer is respectively connected to the midpoint of the diode bridge arm and the midpoint of the capacitor bridge arm of the rectifier bridge.

[0005] Furthermore, the turns ratio of the secondary primary winding of the transformer is n:1.

[0006] Furthermore, the resonant capacitor adopts a non-polarized capacitor, and the maximum capacitance value C maxThe following formula must be satisfied under two-level AC square wave input:

[0007]

[0008] Where D is the duty cycle of the positive half cycle of the two-level AC input square wave; L′ k is the parasitic leakage inductance of the transformer; f S is the switching frequency; R is the parasitic resistance value.

[0009] Furthermore, under the two-level AC square wave input, the output gain M and the maximum output power P OMax Satisfies the following formula:

[0010]

[0011]

[0012] In the formula, is the resonance attenuation coefficient, R is the parasitic resistance, L′ k is the parasitic leakage inductance of the transformer, C1′ is the capacitance value of the resonant capacitor, f S is the switching frequency, n is the turns ratio of the transformer's secondary winding to the primary winding, V inpp is the peak-to-peak value of the input AC square wave, P O is the output power size.

[0013] Compared with the prior art, the present invention has the following advantages and beneficial effects:

[0014] 1. In high-frequency scenarios, the voltage gain is greater.

[0015] 2. In high-frequency scenarios, the operating power limit is higher.

[0016] 3. Since the booster unit can transmit greater power under the same transformer parameters, the power density can be further improved. Combined with high-frequency applications, the lightweight design of the offshore wind power booster station can be further realized to reduce the cost and difficulty of system transportation, installation and maintenance. BRIEF DESCRIPTION OF THE DRAWINGS

[0017] Figure 1 It is a circuit diagram of the high power density resonant boost unit of the present invention.

[0018] Figure 2 It is an equivalent circuit diagram of an implementation case of the present invention.

[0019] Figure 3 Figure 2 is the key voltage and current waveforms in a switching cycle in the implementation case.

[0020] Figure 4aThis is one of the circuit mode diagrams of the implementation case within a switching cycle.

[0021] Figure 4b This is the second circuit modal diagram of the implementation case within a switching cycle.

[0022] Figure 4c This is the third circuit modal diagram of the implementation case within a switching cycle.

[0023] Figure 4d This is the fourth circuit modal diagram of the implementation case within a switching cycle.

[0024] Figure 5a Simplified diagram of the resonant tank in one of the modes for the implementation example.

[0025] Figure 5b This is a simplified diagram of the resonance tank in mode 3 of the implementation example.

[0026] Figure 6a This is the experimental waveform diagram of the maximum output power of the PWM type boost unit under a duty cycle of 0.4.

[0027] Figure 6b This is the experimental waveform diagram of the maximum output power of the implementation case under a duty cycle of 0.4.

[0028] Figure 7a This is the experimental waveform diagram of the maximum output power of the PWM type boost unit under a duty cycle of 0.5.

[0029] Figure 7b This is the experimental waveform diagram of the maximum output power of the implementation case under a duty cycle of 0.5.

[0030] Figure 8a This is the experimental waveform diagram of the maximum output power of the PWM type boost unit under a duty cycle of 0.6.

[0031] Figure 8b This is the experimental waveform diagram of the maximum output power of the implementation case under a duty cycle of 0.6.

[0032] Fig. 9 It is a comparison diagram of the power gain curves of the same PWM type boost unit in the implementation case. DETAILED DESCRIPTION

[0033] The present invention is further described in detail below in conjunction with embodiments and drawings, but the embodiments of the present invention are not limited thereto.

[0034] See also Figure 1As shown, this embodiment provides a transformer-based offshore wind power high power density resonant boost unit, comprising a resonant capacitor C1, a transformer T, a first diode D1, a second diode D2, a first capacitor C2 and a second capacitor C3; the first diode D1, the second diode D2, the first capacitor C2 and the second capacitor C3 constitute a rectifier bridge, wherein the anode of the first diode D1 is connected to the cathode of the second diode D2, the cathode of the first diode D1 is connected to the positive electrode of the first capacitor C2, the anode of the second diode D2 is connected to the negative electrode of the second capacitor C3, the negative electrode of the first capacitor C2 is connected to the positive electrode of the second capacitor C3, one end of the resonant capacitor C1 is connected to one end of the primary winding of the transformer T, and the secondary winding of the transformer T is respectively connected to the midpoint of the diode bridge arm and the midpoint of the capacitor bridge arm of the rectifier bridge.

[0035] Specifically, the turns ratio of the secondary primary winding of the transformer T is n:1 resonant capacitor.

[0036] Specifically, the resonant capacitor C1 is a non-polar capacitor, and the maximum capacitance value C max The following formula must be satisfied under two-level AC square wave input:

[0037]

[0038] Where D is the duty cycle of the positive half cycle of the two-level AC input square wave; L k ′ is the parasitic leakage inductance of the transformer; f S is the switching frequency; R is the parasitic resistance value.

[0039] Specifically, under the two-level AC square wave input, the output gain M and the maximum output power P OMax Satisfies the following formula:

[0040]

[0041]

[0042] In the formula, C1′ is the capacitance value of the resonant capacitor, V inpp is the peak-to-peak value of the input AC square wave, P O is the output power size.

[0043] The equivalent circuit diagram of the specific implementation case is as follows Figure 2 As shown, the input side is connected to a full-bridge inverter structure. It includes a DC input power supply DC, a first switch tube S1 and its anti-parallel diode D S1 , the second switch tube S2 and its anti-parallel diode D S2 , the third switch tube S3 and its anti-parallel diode D S3, the fourth switch tube S4 and its anti-parallel diode D S4 The source and drain of the first switch tube S1 and the second switch tube S2 are connected, the source and drain of the third switch tube S3 and the fourth switch tube S4 are connected, the drain of the first switch tube S1 and the third switch tube S3 are connected, and the source of the second switch tube S2 and the fourth switch tube S4 are connected; the positive and negative poles of the DC input power supply DC are respectively connected to the upper and lower ends of the inverter bridge. The resonant capacitor C1 and the primary winding of the transformer T are respectively connected to the midpoints of the two bridge arms in the full-bridge inverter structure. The output load Load is connected to the upper and lower ends of the rectifier bridge. The transformer T uses a leakage inductor L k , Excitation inductance L m With the ideal transformer T r Model replacement, L m is the magnetizing inductance of transformer T, L k is the leakage inductance of transformer T, ideal transformer T r The secondary-primary turns ratio is n:1, and R is the parasitic resistance.

[0044] The following will analyze the implementation case to clarify the design process and gain and power characteristics of the resonant capacitor C1.

[0045] 1) Modal analysis

[0046] Figure 3 Draw the key waveform diagram of the implementation case under stable working conditions.

[0047] GS1, GS2, GS3, and GS4 represent the drive signals of four switch tubes (S1, S2, S3, and S4) respectively; ab 、v cd Respectively represent the primary and secondary voltages of transformer T; i C1 Represents the current flowing through the resonant capacitor C1; u C1 Represents the voltage on the resonant capacitor C1. Figure 3 Detailed analysis of the circuit's working status:

[0048] a. Stage t0~t1 Figure 4a In this stage, the first switch tube S1 and the fourth switch tube S4 start to conduct under the action of the driving signal, the first diode D1 is turned on, and the primary side of the transformer T transmits power to the secondary side; since the secondary side is clamped by the second capacitor C2 at this time, the resonant capacitor C1 will be connected with the transformer leakage inductance L k Resonance occurs; the second diode D2 is turned off due to reverse voltage; the third capacitor C3 and the secondary side of the transformer T jointly supply power to the load Load; when the leakage inductance L k When the resonant capacitor C1 completes half of the resonant cycle, the leakage inductance L k The current returns to zero and this phase ends.

[0049] b. Phase t1~DTS like Figure 4b In this stage, the first switch tube S1 and the fourth switch tube S4 are still turned on, the first diode D1 and the second diode D2 are turned off by the reverse voltage, and the power transmission between the primary and secondary sides of the transformer T is terminated; the resonant capacitor C1 and the leakage inductance L k 、Inspiration m Resonance occurs due to the excitation L m The inductance is very large, and the total reactance in the resonant circuit is very large, so it can be considered that no current flows through the resonant circuit at this time; the load Load is powered by the second capacitor C2 and the third capacitor C3 in series; when the driving signals of the first switch tube S1 and the fourth switch tube S4 fall to zero, this stage ends.

[0050] c. Stage DT S ~t2 Figure 4c At this stage, the second switch tube S2 and the third switch tube S3 start to conduct under the action of the driving signal, the second diode D2 is turned on, and the primary side of the transformer T transmits power to the secondary side; since the secondary side is clamped by the third capacitor C3 at this time, the resonant capacitor C1 will be connected with the transformer leakage inductance L k Resonance occurs; the first diode D1 is turned off due to reverse voltage; the second capacitor C2 and the secondary side of the transformer T jointly supply power to the load Load; when the leakage inductance L k When the resonant capacitor C1 completes half of the resonant cycle, the leakage inductance L k The current returns to zero and this phase ends.

[0051] d. Stage t2~T S like Figure 4d In this stage, the second switch tube S2 and the third switch tube S3 are still turned on, the first diode D1 and the second diode D2 are turned off by the reverse voltage, and the power transmission between the primary and secondary sides of the transformer T is terminated; the resonant capacitor C1 and the leakage inductance L k 、Inspiration m Resonance occurs due to the excitation L m The inductance is very large, and the total reactance in the resonant circuit is very large, so it can be considered that no current flows through the resonant circuit at this time; the load Load is powered by the second capacitor C2 and the third capacitor C3 in series; when the driving signals of the second switch tube S2 and the third switch tube S3 fall to zero, this stage ends.

[0052] 2) Gain and power analysis

[0053] To simplify the analysis of voltage gain, we first make the following definition (a) and make reasonable assumptions (b, c, d):

[0054] a. Duty cycle D = T on / T S , where T s is the circuit's duty cycle, T onis the conduction time of the switch tubes S1 and S4, that is, the time when their drive signals are at a high level;

[0055] b. The capacitance of the second capacitor C2 and the third capacitor C3 is large enough so that the voltage at their terminals remains constant during the switching cycle;

[0056] c. All the resistances in the circuit are concentrated and superimposed on the parasitic resistance R in the transformer winding;

[0057] d. Due to stage t1~DT S and t2~T S The resonant reactance is very large, so it can be considered that the current flowing through the resonant capacitor C1 in these two stages is zero, and thus the voltage on the resonant capacitor C1 remains unchanged in these two stages.

[0058] According to hypothesis d, we focus on analyzing phases 0 to t1 and DT S ~t2, and the excitation sensor L m In these two stages, the circuit is considered open. Figure 5a and Figure 5b The resonant circuit diagrams of these two stages are given respectively.

[0059] For stage t0~t1, according to Figure 5a From Kirchhoff's voltage law (KVL), we can get:

[0060] u C1 +u Lk +u R =V in1 -V o1 / n (1)

[0061] Among them, V in1 、V o1 They are the voltages of the primary winding and the secondary winding of the transformer T at this stage, respectively. Figure 4a It can be seen that V in1 、V o1 are equal to the voltage of the input DC power supply DC and the voltage on the second capacitor C2 respectively; u Lk 、u R They are leakage inductance L k and the voltage across the parasitic resistance R.

[0062] According to the characteristic equations of inductance and capacitance, the above equation can be converted into the following differential equation:

[0063] Solving the above differential equation yields:

[0064]

[0065] in, The resonant frequency is used to determine the upper limit of the capacitance of the resonant capacitor C1, and the corresponding t1 = π / ω is the time of the resonant process. R≤4L k ′ / C1′ is used to ensure that the circuit can work in the expected mode in an underdamped state; E1 and E2 are two unknown coefficients.

[0066] Similarly for stage DT S ~t2, according to Figure 5b You can get:

[0067]

[0068] Where V in2 、V o2 They are the voltages of the primary winding and the secondary winding of the transformer T at this stage, respectively. Figure 4c It can be seen that V in2 、V o2 are equal to the voltage of the input DC power supply DC and the voltage on the third capacitor C3 respectively; E3 and E4 are two unknown coefficients; and t2=DT S +π / ω.

[0069] Combination Figure 3 You can get:

[0070]

[0071] Then we can get the maximum capacitance C of the resonant capacitor C1. max :

[0072]

[0073] The gain characteristics and maximum operating power will be further discussed below in conjunction with steady-state conditions.

[0074] According to the assumption d and the periodic steady-state condition, we can get:

[0075] u C1 (0) = u C1 (DT S +π / ω) (7)

[0076] Combining (3)(4)(7), we can get:

[0077]

[0078] Based on the assumption of continuity of d and capacitor voltage, we can obtain:

[0079] u C1 (π / ω)=u C1 (DT S ) (9)

[0080] Combining (3)(4)(9), we can get:

[0081]

[0082] Combining (8) and (10) we can get:

[0083] E1+E3=0(11)

[0084] Further, according to the condition that the initial current value at the beginning of resonance is 0, we can obtain:

[0085]

[0086] Combining (3)(4)(11)(12) we can get:

[0087]

[0088]

[0089] Combining (8)(10)(11)(13)(14) and the circuit, the output voltage can be obtained, that is, the voltage V of the second capacitor C2 o1 and the voltage V of the third capacitor C3 o2 The algebraic sum of :

[0090] V O =V o1 +V o2 =n×[V in1 +V in2 +E1(1-λ)] (15)

[0091] in,

[0092] According to the current relationship of the secondary side of the transformer T, the average current I of the first diode D1 can be obtained. DO1 :

[0093]

[0094] Among them, f S is the switching frequency, which is equal to the switching period T S The reciprocal of .

[0095] According to the ampere-second balance condition of the load Load, the average current I of the first diode D1 can be obtained. DO1 and load current I O Relationship:

[0096] I DO1 =I O (17)

[0097] Consider the output load power PO :

[0098] P O =V O I O (18)

[0099] Combining (15)(16)(17)(18) we can get:

[0100]

[0101] Among them, V inpp is the peak-to-peak value of the transformer primary winding voltage, which is equal to V in1 and V in2 The algebraic sum of .

[0102] From (19), we can get:

[0103]

[0104] Correspondingly, the voltage gain M can be expressed as:

[0105]

[0106] Consider the phase t1~DT S and t2~T S In order to ensure that the circuit works as expected, the voltage of the secondary winding of the transformer T needs to be lower than the voltage of this mode at the end of the previous mode. Correspondingly, we can get:

[0107] -u C1 (0)-V in2 ≤V o1 / n (22)

[0108] u C1 (π / ω)-V in1 ≤V o2 / n (23)

[0109] Solving the above equation, we can get the maximum output power of the circuit:

[0110]

[0111] 3) Experimental verification and performance comparison

[0112] The present invention is now experimentally verified, and the parameters of each component are shown in Table 1. In order to verify the advantages of the present invention, the same PWM type boost unit structure is used as a control object, and its structure is consistent with the implementation case here, and the difference is that the capacitance of the resonant capacitor C1 is larger (the voltage u C1It can be regarded as a constant value during the switching cycle. In the experiment, 20μF is used. It will not be affected by the leakage inductance L of the transformer T in each switching mode. k Resonance occurs. Its gain M PWM With power P OPWM The characteristics can be derived as:

[0113]

[0114]

[0115] The specific experimental waveform is as follows Figure 6a , Figure 6b , Figure 7a , Figure 7b , Figure 8a and Figure 8b shown.

[0116] Table 1 Parameter selection of each component in the boost unit of the present invention

[0117] element Parameter value / model DC input power supply voltage / V 25 Switching tube operating frequency / kHz 100 Duty cycle design range D 0.4～0.6 <![CDATA[The first, second, third, and fourth switching transistors (S1, S2, S3, S4)]]> IRFP4110PBF <![CDATA[First and second diodes (D1, D2)]]> STTH6004W <![CDATA[The exciting inductance L of transformer T m and the leakage inductance L k / μH]]> 2000；5.4 Transformer T secondary primary turns ratio n 8:16 <![CDATA[Resonant capacitance, first and second capacitors (C1, C2, C3) / μF]]> 0.3；100；100

[0118] When the duty cycle is set to 0.4, the maximum power experimental waveforms of the boost unit of the present invention and the PWM boost unit structure are as follows: Figure 6a and Figure 6b As shown, by comparing the output voltage, it can be seen that the output voltage of the boost unit of the present invention is 82V, which is significantly greater than the 59V of the PWM type boost unit; when the duty cycle is set to 0.5, the maximum power experimental waveforms of the boost unit of the present invention and the PWM type boost unit structure are as follows Figure 7a and Figure 7b As shown in the figure, the output voltage of the PWM boost unit structure is only about 2V higher than that of the 0.4 duty cycle, but it is still significantly lower than the 82V output voltage of the boost unit of the present invention; and when the duty cycle increases to 0.6, the experimental waveform Figure 8a and Figure 8b It can be seen that the output voltage of the PWM type boost unit structure is about 2V lower than that of the 0.5 duty cycle, while the output voltage of the boost unit of the present invention remains unchanged at 82V and is significantly higher than the output voltage of the PWM type boost unit structure; in order to more comprehensively compare the performance of the two structures, Fig. 9 The power gain curves of the two structures are compared, from which it can be seen that the power upper limit of the boost unit of the present invention is about 115W, which is about twice the maximum output power of 58W of the PWM type boost unit, and the boost unit of the present invention has a more stable and higher voltage gain characteristic within the full power range; similarly, the experimental data is basically consistent with the theoretical calculation within the error range, which further verifies the reliability of the theoretical analysis.

[0119] In summary, the experimental results are consistent with the theoretical analysis and verify the correctness of the aforementioned analysis. Compared with the traditional PWM boost structure, the present invention has a more stable gain and a larger power upper limit at a higher switching frequency of 100kHz. Accordingly, the power size can be further increased without increasing the volume. Therefore, combined with high-frequency applications, the lightweight design of the offshore wind power booster station can be further realized to reduce the cost and difficulty of system transportation, installation, and maintenance.

[0120] The above is only a preferred embodiment of the present invention, and the present invention is not limited to the above embodiment. During the implementation process, there may be changes in the boost structure and other circuit models, drive signals and related parameters. If the changes or deformations of the present invention do not deviate from the spirit and scope of the present invention and fall within the scope of the claims and equivalent technologies of the present invention, the present invention is also intended to include these changes and deformations.

Claims

1. A transformer-based offshore wind power high power density resonant boost unit, characterized in that: The invention comprises a resonant capacitor (C1), a transformer (T), a first diode (D1), a second diode (D2), a first capacitor (C2) and a second capacitor (C3); the first diode (D1), the second diode (D2), the first capacitor (C2) and the second capacitor (C3) constitute a rectifier bridge, wherein the anode of the first diode (D1) is connected to the cathode of the second diode (D2), the cathode of the first diode (D1) is connected to the positive electrode of the first capacitor (C2), the anode of the second diode (D2) is connected to the negative electrode of the second capacitor (C3), the negative electrode of the first capacitor (C2) is connected to the positive electrode of the second capacitor (C3), one end of the resonant capacitor (C1) is connected to one end of the primary winding of the transformer (T), and the secondary winding of the transformer (T) is respectively connected to the midpoint of the diode bridge arm and the midpoint of the capacitor bridge arm of the rectifier bridge; The resonant capacitor (C1) is a non-polar capacitor, and the maximum capacitance value C max The following formula must be satisfied under two-level AC square wave input: (1)； Where D is the duty cycle of the positive half cycle of the two-level AC input square wave; is the parasitic leakage inductance of the transformer; f S is the switching frequency; R is the parasitic resistance value; Under two-level AC square wave input, the output gain With the maximum output power P OMax Satisfies the following formula: (2)； (3)； In the formula, is the resonance attenuation coefficient, , is the capacitance value of the resonant capacitor, f S is the switching frequency, n is the turns ratio of the transformer's secondary winding to the primary winding, V inpp is the peak-to-peak value of the input AC square wave, P O is the output power size.

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