Phase Angle Synchronization Method for Bidirectional Wireless Charging System Based on SOGI Phase-Locked Loop

Abstract

Frequency deviation of the circuits on both sides of the coil in a bidirectional wireless power transfer system will cause power oscillation of the system. Since it is difficult to solve this problem due to the delay in the synchronization method using wireless communication, achieving phase angle synchronization on both sides of the coil without communication is a difficult problem that needs to be solved urgently in the bidirectional wireless power transfer system. The present invention discloses a phase angle synchronization method for a bidirectional wireless charging system based on a SOGI phase-locked loop. It adopts a phase angle detection technology based on a SOGI (Second-Order Generalized Integrator) phase-locked loop, establishes a steady-state mathematical model, derives the vector relationship between voltages to obtain the reference phase angle difference, tracks the phase angle difference through a double integral controller, and inputs it into a phase shift modulator to generate a driving voltage, which can achieve bidirectional seamless switching. Finally, the present invention constructs a simulation model based on MATLAB / Simulink to verify that the proposed phase synchronization method has good steady-state and dynamic performance.

Description

Technical Field

[0001] The present invention relates to a phase angle synchronization method for a bidirectional wireless charging system based on a SOGI phase-locked loop. Background Art

[0002] With the booming development of the new energy industry, electric vehicles are further replacing fuel vehicles as the main force of new energy vehicles. The wireless charging technology for electric vehicles will become the main charging method for future electric vehicles due to its advantages such as low cost, reliability, safety, and portability. With the rise of smart distribution grids and the energy Internet, the power battery of electric vehicles, as a good distributed power source with a large energy storage capacity, has received extensive attention for its energy interaction technology with the power grid. The bidirectional inductive power transfer system (BD-IPT) can achieve energy interaction between vehicles and the grid and has functions such as power distribution and peak shaving and valley filling, which will play an important role in the development of future smart grids.

[0003] In order to maximize power transfer, the BD-IPT system often needs to add a compensation circuit. Commonly used compensation circuits include SS, PP, LCL, and LCC, etc. Among them, the LCC compensation circuit has advantages such as high transmission power and strong anti-offset ability. And in the national standard GB_T 38775.1-2020 "Wireless Charging System for Electric Vehicles - General Requirements", the LCC compensation circuit is selected as the standard circuit. Therefore, the present invention adopts a symmetric LCC topology on both sides, and the system resonance frequency is 85 kHz.

[0004] In the BD-IPT system using the above compensation circuit, the power transmitted by the system is related to the phase angle difference of the fundamental voltages on both sides. Due to electrical isolation, the frequency deviation of the control boards on both sides will cause the phase angle difference to change rapidly, resulting in system power oscillation. However, wireless communication cannot effectively solve the problem due to delays and other issues. Therefore, it has become an urgent problem to achieve phase angle synchronization on both sides without using wireless communication.

[0005] There are mainly three phase angle synchronization schemes without communication: 1) Using an additional winding to detect the induced voltages or currents on both sides, which requires a complex hardware circuit and also increases the cost; 2) Utilizing the relationship between the indirect electrical quantity and the phase angle difference, and indirectly controlling the phase angle difference change by detecting and controlling this electrical quantity. This method is very sensitive to changes in the parameters of passive components and requires a complex calculation process; 3) Directly measuring the phase angle of the AC physical quantity and performing control. This method generally uses a low-pass filter and a zero-crossing detection circuit, and the accuracy of phase angle measurement is relatively low in the case of high harmonic distortion rate and frequency detuning, and needs to be further improved.

[0006] Therefore, it is necessary to design a new phase angle synchronization method. Summary of the Invention

[0007] The technical problem to be solved by the present invention is to provide a phase angle synchronization method for a bidirectional wireless charging system based on a SOGI phase-locked loop. The phase angle synchronization method for the bidirectional wireless charging system based on the SOGI phase-locked loop introduces a control closed-loop to track the change of the phase angle difference and realizes the phase angle synchronization between the primary side and the secondary side.

[0008] The technical solution of the invention is as follows:

[0009] A phase angle synchronization method for a bidirectional wireless charging system based on a SOGI phase-locked loop uses a controller to control the phase-shift modulation H-bridge to realize the real-time phase angle synchronization control of the bidirectional wireless charging system based on the SOGI phase-locked loop;

[0010] The given value of the controller is The feedback quantity of the controller is The output quantity of the controller is θ rec ;

[0011] The phase-shift modulation H-bridge is the actuator corresponding to the controller; the phase-shift modulation H-bridge has 2 output parameters: θ rec and β rec ; β rec is the phase-shift angle of the secondary full-bridge converter.

[0012] Wherein:

[0013] is the reference value of the phase angle difference between u st and u si ; u si is the mutually induced voltage, which cannot be directly sampled in practice, u si = u st -(1-ω 2 L2C sr )u sr , V si = jωMI po ;

[0014] is the phase angle difference between u st and u si ;

[0015] u sr is the real-time voltage across the capacitor Csr, and Csr can be called the secondary compensation capacitor;

[0016] u stis the real-time voltage across the capacitor Cst. Cst is called the secondary resonant capacitor. Similarly, Lf2 is called the secondary resonant inductor, and L2 is the self-inductance of the secondary coil. On the left, Lf1 is the primary resonant inductor, L1 is the self-inductance of the primary coil, Cpt is the primary resonant capacitor, and Cst is the primary compensation capacitor. M is the mutual inductance of the coupled coils;

[0017] θ rec is the phase angle of the secondary voltage Vso;

[0018] β rec is the phase shift angle of the secondary voltage, which can change the duty cycle of the square wave voltage to adjust the equivalent voltage amplitude.

[0019] The controller mentioned above is a PI double-integral controller. The double-integral controller is a PI controller plus an integral link. The purpose of choosing the double-integral controller is that the phase angle error is generated by integrating the frequency error. If the frequency difference is regarded as constant, then the phase angle difference is not a step signal, and simple PI control cannot track the phase angle difference without steady-state error. Therefore, in the present invention, the frequency is tracked without steady-state error through a PI controller, and then the phase angle difference is obtained through an integrator to achieve the steady-state error-free tracking of the phase angle difference.

[0020] The calculation of u si and u st After sampling, the phase angles are obtained respectively through the software SOGI phase-locked loop in the DSP, and then the difference is made to obtain u si = u st -(1 - ω 2 L2C sr )u sr , u rt is the real-time voltage across the secondary compensation capacitor C sr , which is obtained by direct sampling;

[0021] L2 is the self-inductance value of the secondary side of the coupled inductor. In the embodiment of the present invention, when the frequency is 85 kHz, then ω is the fundamental angular frequency, and its value is: 2 * pi * 85000.

[0022] The formula is used to calculate

[0023] V st is the effective value of u st , where △θ ref = ±90°, which changes according to the reference direction of the power flow; Vso is the effective value of the fundamental component of the square wave voltage output by the secondary full-bridge converter. Vso is a constant value during actual operation.

[0024] β recFor adjusting the duty cycle of the secondary square-wave voltage, in this patent, βrec is determined to be 180° later, that is, no phase shift is performed, but the phase shift angle can be adjusted to regulate the voltage.

[0025] The main circuit of the bidirectional wireless charging system based on the SOGI phase-locked loop includes a primary inverter, a charging coil, and a secondary inverter connected in sequence; both the primary inverter and the secondary inverter adopt full-controlled full-bridge converters.

[0026] Beneficial effects:

[0027] In a bidirectional wireless power transmission system, the frequency deviation between the circuits on both sides of the coil will cause system power oscillation. The synchronization method using wireless communication has difficulty solving this problem due to delay. Therefore, achieving phase angle synchronization between the two sides of the coil without communication is an urgent problem to be solved in the bidirectional wireless power transmission system. The present invention discloses a phase angle synchronization method for a bidirectional wireless charging system based on a SOGI phase-locked loop. It adopts a phase angle detection technology based on a SOGI (second-order generalized integrator) phase-locked loop, establishes a steady-state mathematical model, deduces the vector relationship between voltages to obtain the reference phase angle difference, tracks the phase angle difference through a double-integral controller, and inputs it into a phase-shift modulator to generate a driving voltage, which can achieve bidirectional seamless switching. Finally, the present invention builds a simulation model based on MATLAB / Simulink to verify that the proposed phase synchronization method has good steady-state and dynamic performance.

[0028] The method of directly measuring the phase angle of AC physical quantities has strong anti-offset ability and a simple calculation process without using additional hardware circuits. Therefore, aiming at the problems of high harmonic distortion rate and low phase angle detection accuracy in the LCC topology, the present invention proposes a phase angle synchronization method for a bidirectional wireless charging system based on a SOGI phase-locked loop. The present invention uses a SOGI phase-locked loop to detect the phase angles of the induced voltage and the capacitor voltage (both are sinusoidal signals), determines the relationship between the phase angle difference and the effective voltage value by establishing a steady-state model and a vector relationship diagram, obtains the reference value of the secondary side voltage phase angle, and introduces a control closed-loop to track the change of the phase angle difference, realizing the phase angle synchronization between the primary side and the secondary side. Description of the drawings

[0029] Figure 1 It is the structural diagram of the system main circuit;

[0030] Figure 2 It is the schematic diagram of the equivalent circuit;

[0031] Figure 3 It is the phase angle synchronization control block diagram;

[0032] Figure 4 It is the SOGI control block diagram;

[0033] Figure 5 It is the voltage relationship phasor diagram;

[0034] Figure 6 is the voltage waveform when the system is running forward;

[0035] Figure 7 is the voltage waveform when the system is running in reverse;

[0036] Figure 8 is the power waveform when the system switches seamlessly between forward and reverse. Detailed implementation method

[0037] The present invention will be further described in detail below with reference to the accompanying drawings and specific embodiments:

[0038] Embodiment 1: As Figures 1 to 8 , a phase angle synchronization method for a bidirectional wireless charging system based on a SOGI phase-locked loop, which uses a controller to control a phase-shifted modulation H-bridge to achieve real-time phase angle synchronization control of the bidirectional wireless charging system based on the SOGI phase-locked loop;

[0039] The given value of the controller is The feedback quantity of the controller is The output quantity of the controller is θ rec ;

[0040] The phase-shifted modulation H-bridge is the actuator corresponding to the controller; the phase-shifted modulation H-bridge has 2 output parameters: θ rec and β rec ; β rec is the phase shift angle of the secondary full-bridge converter.

[0041] Among them:

[0042] is the reference value of the phase angle difference between u st and u si ;

[0043] is the phase angle difference between u st and u si ;

[0044] u sr is the real-time voltage across the capacitor Csr;

[0045] u st is the real-time voltage across the capacitor Cst;

[0046] θ rec is the phase angle of the secondary voltage Vso;

[0047] β rec is the phase shift angle of the secondary voltage.

[0048] The controller is a PI double-integral controller.

[0049] Calculation of u si and u st After sampling, the phase angles are obtained respectively by the software SOGI phase-locked loop in the DSP, and then the difference is calculated to obtain

[0050] Using the formula Calculate

[0051] V st is the effective value of u st where △θ ref = ±90°; Vso is the effective value of the fundamental component of the square-wave voltage output by the secondary full-bridge converter.

[0052] β rec is 180°.

[0053] The main circuit of the bidirectional wireless charging system based on the SOGI phase-locked loop includes a primary inverter, a charging coil, and a secondary inverter connected in sequence; both the primary inverter and the secondary inverter adopt full-controlled full-bridge converters.

[0054] Figure 3 In [0000323], RMS is the identifier for effective value calculation.

[0055] I. Steady-state mathematical model of the BD-IPT system

[0056] The main circuit of the BD-IPT system is as Figure 1 shown. The two sides of the system are DC sources V dc1 and V dc2 , and square-wave voltages are obtained through the full-controlled full-bridge converters on both sides. The fundamental components of the square-wave voltages are V pi and V so . Energy is transferred through the LCC compensation circuit and the coupled inductor. Among them, the primary LCC circuit consists of a resonant inductor L f1 , a resonant capacitor C pt , a compensation capacitor C pr , and the primary self-inductance L1 of the coil. The secondary LCC circuit consists of a resonant inductor L f2 , a resonant capacitor C st , a compensation capacitor C sr , and the secondary self-inductance L2 of the coil. The mutual inductance of the coupled inductor is M.

[0057] The compensation circuit operates at the resonant frequency of 85 kHz, and its relationship is:

[0058]

[0059] The system operates at the resonant frequency point. The present invention adopts the fundamental wave approximation method to establish a steady-state mathematical model.

[0060] The square-wave voltage output by the primary full-bridge circuit is regarded as the reference voltage V pi ∠0°. The induced voltages V on both sides of the coupled inductor si and V po have the following corresponding relationship, where I po and I si are the coil currents passing through the primary and secondary sides of the coupled inductor respectively:

[0061] V si = jωMI po (3)

[0062] V po = jωMI si (4)

[0063] Perform a delta transformation on the above main circuit, convert the original T-type network into a π-type, and the circuit parameters remain unchanged after equivalence, as Figure 2 shown. Among them, the capacitors C pr and C sr are equivalent to the coil self-inductances L1 and L2 according to the following relationship.

[0064]

[0065] So far, the present invention has established a steady-state mathematical model of BD-IPT, and the phase synchronization method will be proposed based on the above model in the following text.

[0066] II. Phase Angle Synchronization Method Based on SOGI Phase-Locked Loop

[0067] 3.1 Equivalent Transformation and Analysis of Synchronization Problem

[0068] According to Kirchhoff's law and the equivalent π-type circuit, as Figure 2 , the primary input current I pi and the coil self-inductance current I po are as follows:

[0069]

[0070] Among them:

[0071] L pt_eq = L pt = L f1 (9)

[0072] C pt_eq = C pt

[0073] Combined with (1), it can be simplified at steady state as:

[0074]

[0075] Similarly, due to the symmetric topology on both sides, there is a similar corresponding relationship for the secondary-side current. The proof method is as above.

[0076]

[0077] where L st = L pt , C st = C pt Hereinafter, L st is used to represent.

[0078] Express the secondary-side output voltage as V so ∠θ°. Then the expression of the output active power is as follows (here V pi and V so represent the effective value of the fundamental voltage):

[0079]

[0080] It can be clearly seen that the positive or negative of the phase angle difference θ between the voltages on both sides determines the direction of power flow transfer. To ensure that the system transfers the maximum active power and operates at unity power factor, the phase angle difference needs to be maintained at ±90°. For the control of the active power magnitude, the present invention adopts a phase-shift modulation method to change the equivalent voltage amplitudes on both sides so as to achieve the goal of controlling the power magnitude.

[0081] It should be noted that from (3) and (11), for a BD-IPT system with a symmetric LCC compensation network, the phase of the primary-side coil current I po lags behind the primary-side input voltage V pi by 90°. Then the secondary-side induced voltage V si is in phase with the fundamental component of the primary-side input voltage V pi . In the steady state, this relationship is not affected by the fundamental component of the secondary-side voltage V si . Therefore, the problem of voltage phase synchronization on both sides can be transformed into the problem of phase synchronization between V si and V so .

[0082] However, since V so is a square-wave voltage and the phase angle parameter cannot be accurately obtained, the present invention selects indirect electrical quantities (the secondary-side induced voltage V si and the secondary-side capacitor voltage V st ) for phase angle detection, obtains the phase angle difference, and introduces a control closed-loop.

[0083] Since i si and i st are still sine waves even when the phases are not synchronized during the operation of the system, so u si , u sr and u stThey are all sine waves and can be expressed as:

[0084]

[0085] Among them, V si , V st , V sr and I si are the effective values of the fundamental wave components of u si , u st , u sr and i si respectively.

[0086] Sample u sr and u st through the high-frequency sampling circuit, and calculate u si according to the following relationship:

[0087] u si = u st -(1 - ω 2 L2C sr )u sr (16)

[0088] Because (16) only involves the product transformation between voltage quantities, and u sr and u st use the same conditioning circuit and are arranged nearby, the phase shift caused by the sampling and conditioning circuit will not affect this algorithm.

[0089] In summary, the present invention detects the phase angles of the secondary side induced voltage u si and the secondary side capacitor voltage u st , and controls the phase synchronization of V si and V so to solve the problem of the phase of the two control boards.

[0090] 3.2 Phase Angle Synchronization Method

[0091] The present invention will implement the phase angle synchronization method based on the above relationship. The proposed phase angle synchronization method is as Figure 3 shown. Introduce u st and u si into the SOGI phase-locked loop to detect the phase angle difference in real time, calculate the steady-state vector relationship of the fundamental wave effective values V st , V si of u st , V si and the secondary side output voltage V so , and feedback the real-time phase angle difference to obtain the reference value The secondary side phase reference signal is generated through closed-loop double integral control and input into the H-bridge. The phase-shift modulation method is adopted to obtain the driving signal for the switching tube.

[0092] 1. Phase angle detection method of SOGI phase-locked loop

[0093] To solve the problem of phase angle detection error caused by frequency offset and harmonics, the present invention uses a SOGI phase-locked loop for phase angle detection. The obtained u st and u si are input into the SOGI phase-locked loop to obtain the phase angle difference between the two, denoted as

[0094] The generalized second-order integral PLL (Second-Order Generalized Integrator-Based PLL, SOGI-PLL), also known as the quadrature integrator, is widely used in solving the detection problem of single-phase sinusoidal signals. The SOGI-QSG module creates virtual quadrature signals, which are input into the PLL through a second-order integral circuit and dq transformation. The gain of the SOGI in the frequency domain at the resonant frequency point is very large, and the bandwidth is small. Without passing through a low-pass filter, it can effectively filter out voltage harmonics. The frequency estimated by the PLL will be fed back to the SOGI-QSG to adapt to frequency changes. In the case of frequency offset, the phase angle can still be accurately tracked.

[0095] 2. Phase angle synchronization method

[0096] After obtaining the real-time phase angle difference , it is also necessary to find its reference value to establish a control closed-loop.

[0097] At steady state, the relationship between the fundamental components of V so and V st can be clearly obtained as follows:

[0098] V so =V st +jωL f2 ·I so (17)

[0099] Combining (13) and (17), we can get:

[0100] V st =V so +V si (18)

[0101] Its relationship can be represented by a phasor diagram, as shown in Figure 5 .

[0102] As can be seen from the above, to control the phase angle difference between the two sides of the voltage to be ±90° to achieve the purpose of maximum active power output, it is necessary to control V si and Vso The phase angle difference is ±90°. V si , V st and V so satisfy the relationship of the sine theorem, and it is also very easy to obtain the fundamental effective value of V st and V so . With the fundamental effective value, the phase angle difference reference value

[0103]

[0104] where Vθ ref = ±90°, and its change is based on the reference direction of the power flow.

[0105] Through the SOGI phase-locked loop, the phase angle difference between V si and V st is tracked in real time. The obtained is subtracted from the phase angle difference reference value to obtain an error value. Then, the error value is input into a double-integral loop controller. The loop controller consists of a proportional-integral (PI) controller, an integrator, and a clipping function, and then a drive signal is output through a phase-shifted modulation full-bridge circuit.

[0106] The purpose of selecting the double-integral controller is that the phase angle error is generated by integrating the frequency error. If the frequency difference is regarded as constant, then the phase angle difference is not a step signal, and simple PI control cannot perform static-error-free tracking of the phase angle difference. Therefore, in the present invention, the PI controller is used to perform static-error-free tracking of the frequency, and then the phase angle difference is controlled by the integrator. Finally, the phase angle difference si between V st and V is tracked to the phase angle difference reference value . The voltages Vθ on both sides are stabilized at ±90°, and the output power reaches the stable maximum power point, realizing phase angle synchronization.

[0107] 3. Phase-Shifted Modulation Method

[0108] The present invention adopts a phase-shifted modulation method. The phase-shifted modulation method has advantages such as simple implementation method and a wider voltage gain range. It outputs a three-level square wave voltage by time-shifting the drive signal, and adjusts the duty cycle by adjusting the phase-shift angle to change the equivalent voltage amplitude, so as to achieve the purpose of adjusting the active power.

[0109] It should be noted that phase-shifted modulation will cause the phase angle difference on both sides to shift, making the phase angle difference on both sides unable to be stabilized at ±90°. The system will generate reactive power, increasing the current stress on the device. Therefore, a certain phase compensation is required, as shown in the following formula:

[0110]

[0111] The driving signal of the present invention is input into the phase-shifted full-bridge converter through the controller, and the phase-shift angle β is input at the same time. rec In the present invention, the phase-shift angle is fixed at 180°, and the duty cycle of the square wave is 50%.

[0112] Explanation of model and principle issues

[0113] 1. The system establishes a steady-state mathematical model through the equivalent conversion of T-type - π-type.

[0114] Model parameter description: Equations 5 - 6, which are subsequently represented by L st and unified representation is used because their inductance values are the same.

[0115]

[0116] Model principle: Equations 7 - 14. The characteristic formula 14 representing the model elaborates that the power flow is controlled by the phase angle difference θ between the voltages on both sides, and the phase angle difference θ needs to be controlled within ±90° (the goal of the phase synchronization problem).

[0117] The secondary output voltage is expressed as V so ∠θ°. Then the expression of the output active power is as follows (here V pi and V so represent the fundamental voltage effective values):

[0118]

[0119] 2. Transformation of the phase synchronization problem

[0120] Equations 3 and 11 indicate that the fundamental component of the secondary induced voltage V si is in phase with the fundamental component of the primary input voltage V pi . In the steady state, this relationship is not affected by the fundamental component of the secondary voltage V si . The phase synchronization problem of the voltages on both sides (V pi and V so ) can be transformed into the phase synchronization problem of V si and V so . (Transformation of the phase synchronization problem. This part is based on the analysis of the above steady-state model)

[0121] 3. Phase angle detection

[0122] Sample u st and u sr , and obtain u si in real time through Equation 16 (all three are transient values). Obtain the phase angle difference st between u si and u through the SOGI phase-locked loop (SOGI needs to detect real-time transient values). Corresponding to Figure 3in the upper left part

[0123] u si = u st -(1 - ω 2 L2C sr )u sr (16)

[0124] 4. Phase angle synchronization control closed - loop

[0125] Equations 17 - 19 illustrate how to obtain u st and u si phase angle difference reference value corresponds to Figure 3 the steady - state vector calculation in (this part uses the above - mentioned steady - state model). Introduce a double - integral control closed - loop and phase - shift modulation to control the secondary full - bridge converter to achieve phase - angle synchronization of the voltages on both sides.

[0126] At steady state, it can be clearly obtained that V so and V st The relationship between the fundamental components is as follows:

[0127] V so = V st + jωL f2 ·I so (17)

[0128] V st = V so + V si (18)

[0129] Then the phase - angle difference reference value .

[0130]

[0131] IV. Experimental simulation results

[0132] The present invention builds a simulation platform based on MATLAB / simulink to verify the phase - angle synchronization method of the proposed bidirectional wireless charging system. The parameter settings in the simulation are shown in Table 1:

[0133] Table 1: Simulation parameters of the bidirectional wireless charging system

[0134]

[0135] The actual operation results of the bidirectional wireless charging system are as Figures 6 - 8 shown.

[0136] Figure 6 For the waveform diagram of the voltages on both sides when the system is running forward, it can be observed that the voltages V pi and V soThe phase angle difference is stabilized at 90°, Figure 7 It is the waveform diagram of the voltages on both sides when the system runs in the reverse direction. It can be observed that voltage V pi and V so The phase angle difference is stabilized at -90°, Figure 8 The output power waveform when the system switches from forward operation to reverse operation. It can be found that the system realizes seamless and smooth switching between forward and reverse directions within 4 ms. The actual operation results can fully prove the effectiveness of the method of the present invention.

Claims

1. A phase angle synchronization method for a bidirectional wireless charging system based on a SOGI phase-locked loop, characterized in that The main circuit of the bidirectional wireless charging system based on the SOGI phase-locked loop includes a primary inverter, a charging coil, and a secondary inverter connected in sequence; both the primary inverter and the secondary inverter adopt full-controlled full-bridge converters; The two sides of the system are DC sources V dc1 and V dc2 . The square-wave voltage is obtained through the full-controlled full-bridge converters on both sides. The fundamental components of the square-wave voltage are V pi and V so . The energy is transferred through the LCC compensation circuit and the coupled inductor. Among them, the primary LCC circuit consists of the resonant inductor L f1 , the resonant capacitor C pt , the compensation capacitor C pr and the self-inductance L1 of the primary coil. The secondary LCC circuit consists of the resonant inductor L f2 , the resonant capacitor C st , the compensation capacitor C sr and the self-inductance L2 of the secondary coil. The mutual inductance of the coupled inductor is M; A controller is used to control the phase-shift modulation H-bridge to achieve real-time phase angle synchronization control of the bidirectional wireless charging system based on the SOGI phase-locked loop; The set value of the controller is The feedback value of the controller is The output value of the controller is θ rec ; The phase-shifted modulation H-bridge is the actuator corresponding to the controller; the phase-shifted modulation H-bridge has two output parameters: θ rec and β rec ; θ rec is the phase angle of the secondary side voltage Vso; β rec is the phase-shift angle of the secondary full-bridge converter; Wherein: is u st and u si phase angle difference reference value; is u st and u si phase angle difference; u st is the real-time voltage across capacitor Cst; u si is the mutual inductance induced voltage; The controller is a PI double-integral controller; The calculation of si and st After sampling, the phase angles are obtained by the software SOGI phase-locked loop in the DSP respectively, and then the difference is calculated to obtain Adopt the formula Calculate V st is the effective value of u st , where △θ ref = ±90°; Vso is the effective value of the fundamental component of the square-wave voltage output by the secondary full-bridge converter.

2. The phase angle synchronization method of the bidirectional wireless charging system based on the SOGI phase-locked loop according to claim 1, characterized in that β rec is 180°.

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