A Constant Conduction Time Critical Conduction Mode Control Method for Boost-Type Power Factor Correction Circuit

Abstract

The present invention discloses a constant-on-time current critical continuous control method for a single-phase full-bridge rectifier in the field of power electronics and electrical engineering technology. The sampled signal of the output voltage of the single-phase full-bridge rectifier and the given reference voltage are jointly fed into an adder, and the voltage error signal generated after passing through the adder is fed into a voltage controller; the output signal of the voltage controller is fed into a constant-on-time CRM generator, and the pulse control signal of the high-frequency switching tube with an equivalent duty cycle is generated by the constant-on-time CRM generator. The present invention has the performance of simple control structure, fast response speed and easy stability; the algorithm can be realized by a low-cost digital control chip, and the performance-price ratio is high.

Description

Technical Field

[0001] The present invention relates to the field of power electronics and electrical engineering, and particularly relates to a control method for a single-phase full-bridge rectifier. Background Art

[0002] Power Factor Correction (PFC) can effectively reduce the harmonic distortion of the grid-side current and reactive power, and reduce the pollution of the grid by harmonics. Therefore, it is widely used in rectifier circuits. The traditional active power factor correction rectifier consists of an uncontrolled rectifier bridge and a Boost boost circuit. However, with the increase of the power level, the loss of the rectifier bridge greatly hinders the improvement of the system efficiency. Therefore, many PFC topologies without rectifier bridges have emerged. Among them, the single-phase full-bridge PFC circuit has the advantages of low common-mode interference, few devices, and high device utilization rate, so it has good application prospects.

[0003] In order to pursue the highest possible efficiency, the working mode and control method of the circuit have become two aspects that researchers focus on. The working mode of the circuit is divided into three types according to whether the current on the energy storage inductor in the circuit is continuous: discontinuous conduction mode (DCM), continuous conduction mode (CCM), and critical conduction mode (CRM).

[0004] The discontinuous conduction mode (DCM) control structure is simple and easy to implement. However, at the same power output, the turn-off loss of the switching tube in this working mode is large, the circuit efficiency is low, and it is difficult to further improve the grid-side power factor. Therefore, it is suitable for applications with high cost requirements and low power.

[0005] The continuous conduction mode (CCM) generally adopts the average current control scheme more. At the same output power, the peak value of the switching tube current is small, the effective value of the current is small, the circuit efficiency is high, and the power factor is high. However, the average current control method is complex and costly; moreover, if a common Si MOSFET is used as the switching tube to work at high frequencies, the reverse recovery characteristics of its body diode may cause unnecessary losses to the circuit and even burn out the switching tube.

[0006] The critical conduction mode (CRM) is commonly used in medium and high frequency cases because it is easy to achieve soft switching. The efficiency of this working mode is high, and the grid-side power factor is high. However, using the traditional average current control also faces problems such as complex structure and high cost.

[0007] In addition, in order to achieve the critical continuous operation of the inductor current, many scholars have proposed to add a current zero-crossing detection circuit to detect whether the inductor current crosses zero, which will undoubtedly further increase the research cost and reduce the circuit reliability. Summary of the Invention

[0008] Aiming at the deficiencies existing in the prior art, the present invention provides a constant-on-time current critical continuous control method for a Boost-type power factor correction circuit, which reduces the circuit complexity and improves the reliability.

[0009] The purpose of the present invention is achieved as follows: A constant-on-time current critical continuous control method for a Boost-type power factor correction circuit:

[0010] 1) The sampling signal of the output voltage of the single-phase full-bridge rectifier and the given reference voltage are jointly sent to an adder, and the voltage error signal generated after passing through the adder is sent to a voltage controller;

[0011] 2) The output signal of the voltage controller is sent to a constant-on-time CRM generator, and the constant-on-time CRM generator generates a pulse control signal for the high-frequency switching tube with an equivalent duty cycle. The constant-on-time CRM generator is composed of a turn-off time calculator and a PFM modulator; the output signal of the voltage controller is the turn-on time signal of the high-frequency switching tube in the single-phase full-bridge rectifier. This signal is divided into three paths. The first path is sent to the turn-off time calculator, and the turn-off time signal of the high-frequency switching tube is obtained by the output of the turn-off time calculator. This turn-off time signal and the second path turn-on time signal are jointly sent to an adder to obtain the switching period signal of the high-frequency switching tube. This switching period signal and the third path turn-on time signal are jointly sent to the PFM modulator, and the output of the PFM modulator is the output of the constant-on-time CRM generator.

[0012] As a further limitation of the present invention, the pulse control signal generated by the PFM modulator generates a drive signal after passing through a drive circuit, which is used to control the on and off of the high-frequency switching tube of the single-phase full-bridge rectifier, so as to control the current on the energy storage inductor in the rectifier to always work in the critical continuous mode, making the single-phase full-bridge rectifier output a stable DC voltage waveform.

[0013] As a further limitation of the present invention, the turn-off time calculator calculates the turn-off time signal of the high-frequency switching tube according to the turn-on time signal of the high-frequency switching tube output by the voltage controller, and applies the volt-second balance principle to the energy storage inductor in the rectifier.

[0014] As a further limitation of the present invention, the transfer function of the switching frequency f of the high-frequency switching tube of the single-phase full-bridge rectifier and the output voltage V o is:

[0015]

[0016] where, P o is the output power of the rectifier, v in_rms is the effective value of the AC input voltage, I LLet \(i\) be the inductor current of the rectifier, \(L\) be the rectifier inductor, \(C\) be the rectifier output capacitor, and \(R\) be the rectifier equivalent load.

[0017] As a further limitation of the present invention, the transfer function of the output voltage sampling and conditioning circuit is:

[0018]

[0019] where \(k\) vf is the voltage loop sampling gain, and \(T\) sp is the sampling period.

[0020] As a further limitation of the present invention, the transfer function of the voltage controller is:

[0021]

[0022] where \(k\) pv and \(k\) iv are the two control parameters of the PI compensator.

[0023] As a further limitation of the present invention, the conversion function of the conduction time \(t\) on of the high-frequency switching transistor and the switching frequency \(f\):

[0024]

[0025] where \(t\) off is the turn-off time of the high-frequency switching transistor, and \(v\) in is the input AC power supply voltage.

[0026] The present invention has the following advantages:

[0027] According to the fixed conduction time of the high-frequency switching transistor in the rectifier, the volt-second balance principle is applied to the energy storage inductor to calculate the turn-off time of the high-frequency switching transistor corresponding to different moments, so that the current on the energy storage inductor always operates in the critical continuous mode without adding an additional current zero-crossing detection circuit, reducing the circuit complexity and being beneficial to improving the reliability; only a single voltage controller is required to achieve a high power factor on the grid side of the rectifier, with a simple control structure, fast response speed, and easy stability; the algorithm can be implemented by a low-cost digital control chip, so it has high cost performance.

[0028] Other advantages and effects of the present invention will be further described below.

[0029] The terms involved in the present invention: PFM (Pulse Frequency Modulation), CRM (Critical Conduction Mode), volt-second balance. Brief Description of the Drawings

[0030] To more clearly illustrate the technical solutions in the embodiments of the present invention or the prior art, the following will briefly introduce the drawings required for the description of the embodiments or the prior art. Obviously, the drawings in the following description are only the embodiments of the present invention. For those of ordinary skill in the art, without creative efforts, other drawings can also be obtained according to the provided drawings.

[0031] Figure 1 It is a control schematic diagram of the present invention;

[0032] Figure 2 It is a schematic diagram of the application circuit composition of the present invention;

[0033] Figure 3 It is a schematic diagram of the voltage loop control principle of the present invention;

[0034] Figure 4 It is a Bode diagram of the system after adding a PI compensator to the present invention;

[0035] Figure 5 It is the input voltage v in the PSIM simulation of the present invention in and the input current i L waveform diagram;

[0036] Figure 6 It is the output voltage V waveform diagram in the PSIM simulation of the present invention o waveform diagram;

[0037] Figure 7 It is the local waveform diagram of the input current i and the drive PFM1B of the high-frequency switching tube S2 within a positive half-cycle of an AC input in the PSIM simulation of the present invention L and the drive PFM1B of the high-frequency switching tube S2;

[0038] Figure 1 Symbol names in

[0039]

[0040]

[0041] Figure 2 Symbol names in

[0042]

[0043] Others are the same as Figure 1 the symbol descriptions in.

[0044] Figure 3 Symbol names in

[0045]

[0046] Others are the same as Figure 1Symbol description.

[0047] Figure 4 Symbol description in:

[0048] dB is the amplitude and fc is the cut-off frequency.

[0049] Figure 5 and Figure 6 Symbols in:

[0050] Time is time.

[0051] Figure 7 Symbol description in:

[0052] PFM1B is the drive signal of the high-frequency switching transistor S2, PFM1A is the drive signal of the high-frequency switching transistor S1, f is the switching period of the high-frequency switching transistor, and others are the same as Figure 2 Symbol description in. Specific implementation mode

[0053] Next, the technical solutions in the embodiments of the present invention will be clearly and completely described in conjunction with the accompanying drawings in the embodiments of the present invention. Obviously, the described embodiments are only a part of the embodiments of the present invention, rather than all the embodiments. All other embodiments obtained by those of ordinary skill in the art based on the embodiments of the present invention without creative efforts shall fall within the protection scope of the present invention.

[0054] As Figure 1 , Figure 2 shown, the component composition of the present invention:

[0055] Single-phase full-bridge rectifier main circuit 1, sampling conditioning circuit 2, voltage controller 3, constant-on-time CRM generator 4, drive circuit 5, where the constant-on-time CRM generator 4 is composed of a turn-off time calculator 6 and a PFM modulator 7, and the voltage controller 3 and the constant-on-time CRM generator 4 can be implemented by a digital control chip. The single-phase full-bridge rectifier main circuit 1 includes an AC input power supply AC, two high-frequency switching transistors S1 and S2, parasitic capacitances C s1 and C s2 of the two high-frequency switching transistors, parasitic body diodes D s1 and D s2 of the two high-frequency switching transistors, a storage inductor L, an output capacitor Co, and an equivalent load resistance R. The voltage across the equivalent load resistance R is the output voltage V o of the single-phase full-bridge rectifier. The output voltage V o is sampled by the sampling conditioning circuit 2 to obtain a rectifier output voltage sampling signal v of , which is sent to an adder together with a given reference voltage v ref to form a voltage error signal v eAs the input signal of the voltage controller 3, the output signal of the voltage controller 3 is the conduction time signal t of the high-frequency switching tube of the rectifier on , and this signal is sent to the constant conduction time CRM generator 4. In the constant conduction time CRM generator 4, two links of the turn-off time calculator 6 and the PFM modulator 7 are completed, and then the pulse control signal d of the high-frequency switching tube with an equivalent duty cycle is output. This signal forms a drive signal after passing through the drive circuit 5, which is used to control the on and off of the high-frequency switching tubes S1 and S2 in the main circuit 1 of the single-phase full-bridge rectifier, so as to realize the constant conduction time current critical continuous control process of the single-phase full-bridge rectifier.

[0056] The present invention realizes a constant conduction time current critical continuous control method for a Boost-type power factor correction circuit. The sampling signal v of the rectifier output voltage of and the given reference voltage v ref are jointly sent to the adder. After subtracting the two, a voltage error signal v e is sent to the voltage controller 3. The output signal of the voltage controller 3 is sent to the constant conduction time CRM generator 4. The constant conduction time CRM generator 4 generates a pulse control signal d of the high-frequency switching tube with an equivalent duty cycle. The constant conduction time CRM generator 4 is composed of a turn-off time calculator 6 and a PFM modulator 7. The output signal of the voltage controller 3 is the conduction time signal t of the high-frequency switching tube in the rectifier on , and this signal t on is divided into three paths in total. The first path is sent to the turn-off time calculator 6, and the turn-off time signal t of the high-frequency switching tube is obtained by the output of the turn-off time calculator 6 off , and the turn-off time signal t off and the second path conduction time signal t on are jointly sent to the adder to obtain the switching period signal T of the high-frequency switching tube s , and the switching period signal T s and the third path conduction time signal t on are jointly sent to the PFM modulator 7. The output of the PFM modulator 7 is the output of the constant conduction time CRM generator 4.

[0057] The pulse control signal d generated by the PFM modulator 7 forms a drive signal after passing through the drive circuit 5, which is used to control the on and off of the high-frequency switching tubes S1 and S2 in the rectifier, so that the main circuit 1 of the single-phase full-bridge rectifier outputs a stable DC voltage V o .

[0058] The following gives a further description in combination with the modeling process and control method design.

[0059] The design and calibration of the voltage controller need to utilize the structural characteristics of the rectifier dynamic equation. Therefore, the state-space averaging method can be used to establish the small-signal model of the single-phase full-bridge rectifier and obtain its AC small-signal state equation. Since Figure 2 the main circuit 1 of the single-phase full-bridge rectifier is equivalent to a Boost circuit in both the positive and negative half-cycles of the AC input power supply, its AC small-signal state equation can be derived based on the Boost circuit.

[0060] Assume that the devices such as the switch tube S, energy storage inductor L, output capacitor C, and equivalent load R in the Boost circuit are all ideal devices, and the dead time of the switch is ignored. According to Kirchhoff's voltage and current laws, the Boost circuit is analyzed using the state-space averaging method.

[0061] When S is conducting, the inductor voltage and capacitor current are

[0062]

[0063] When S is off, the inductor voltage and capacitor current are

[0064]

[0065] Among them, u L (t) is the instantaneous value of the inductor voltage, i L (t) is the instantaneous value of the inductor current, v in (t) is the instantaneous value of the input AC power supply voltage, i o (t) is the instantaneous value of the capacitor current, v o (t) is the instantaneous value of the rectifier output voltage, which is also the instantaneous value of the capacitor voltage.

[0066] Now assume that the switching frequency ripples of the state variables (capacitor voltage v o and inductor current i L ) of the circuit are very small and can be ignored. Then the instantaneous values of the state variables are approximately equal to the average values. Assume that the perturbation frequency is low enough. Within one switching period, the average value is close to the DC component and is approximately constant, that is

[0067]

[0068] Among them, <u L (t)> is the average value of the inductor voltage, <v in (t)> is the average value of the input AC voltage, <v o (t)> is the average value of the rectifier output voltage, T s is the switching period, and d is the duty cycle of the switch tube S.

[0069] Also, because the average value of the inductor voltage can be expressed as

[0070]

[0071] The average value of the inductor voltage can be obtained from Equations (3) and (4) as

[0072]

[0073] The average value of the capacitor current is also expressed in the same way as

[0074]

[0075] where <i o >(t)> is the average value of the capacitor current, and <i L >(t)> is the average value of the inductor current.

[0076] Also, since the average value of the inductor voltage can be expressed as

[0077]

[0078] The average value of the capacitor current can be obtained from Equations (6) and (7) as

[0079]

[0080] The state-average equation (large-signal model) is then

[0081]

[0082] Both the average variables and the control variable d contain DC components and low-frequency small-signal components. To obtain the low-frequency small-signal model, the large-signal model needs to be decomposed into DC components and low-frequency small-signal components. Let

[0083]

[0084] where, I L , V o , V in , D are the DC components of the inductor current, output voltage, input AC supply voltage, and duty cycle respectively; are the low-frequency small-signal components of the inductor current, output voltage, input AC supply voltage, and duty cycle respectively.

[0085] Substituting Equation (10) into Equation (9) gives the state equation as

[0086]

[0087] The DC terms and AC terms on both sides of Equation (11) are equal correspondingly. Thus, the quiescent operating point is obtained as

[0088]

[0089] The AC small-signal state equation is

[0090]

[0091] In Equation (13), the small-signal product term is a non-linear term and belongs to the second-order infinitesimal, which has a minimal impact on the system. Remove it from the equation and linearize the equation. The resulting small-signal analytical model is

[0092]

[0093] Assume that the initial values of each state variable are zero. Perform Laplace transform on Equation (14) to obtain

[0094]

[0095] Design the turn-off time calculator 6: Since the output signal of the voltage controller 3 is the conduction time t of the high-frequency switching tube of the rectifier on , then by applying the volt-second balance principle to the energy storage inductor in the rectifier, the turn-off time t of the high-frequency switching tube can be calculated off as

[0096]

[0097] Adding the conduction time t on and the turn-off time t off can obtain the switching period T s as

[0098] T s = t on + t off (17)

[0099] Taking the reciprocal of the switching period gives the switching frequency f as

[0100]

[0101] Feed the conduction time signal t of the high-frequency switching tube output by the voltage controller 3 on and the switching period signal T of the high-frequency switching tube calculated by Equation (17) s into the PFM modulator 7 together, and then the PFM modulator 7 outputs the pulse control signal d of the high-frequency switching tube with an equivalent duty cycle.

[0102] For single voltage loop control, the relationship between the conduction time of the switching tube and the output power of the circuit can be expressed as

[0103]

[0104] where, P o is the output power of the rectifier, v in_rms is the effective value of the AC input voltage.

[0105] Also, since the duty cycle D, the conduction time t of the high-frequency switching transistor on , and the switching period T s are related as

[0106]

[0107] Then, combining Equation (18) and Equation (19), the relationship between the duty cycle D and the switching frequency f can be obtained as

[0108]

[0109] Taking the differential on both sides, we can get

[0110]

[0111] Expressed in terms of low-frequency small signals as

[0112]

[0113] Substituting Equation (23) into Equation (15), the transfer function G vf (s) of the switching frequency with respect to the output voltage can be obtained as

[0114]

[0115] Method for tuning the parameters of the voltage controller: Using Mathcad software, tune the parameters of the designed voltage loop and conduct a stability check.

[0116] According to Figure 3 the voltage loop control schematic diagram, PI compensator modulation is adopted.

[0117] The transfer function of the output voltage sampling and conditioning circuit is

[0118]

[0119] where k vf is the voltage loop sampling gain, and T sp is the sampling period.

[0120] The transfer function of the PI compensator is

[0121]

[0122] where k pv and k iv are the two control parameters of the PI compensator.

[0123] The open-loop transfer function of the system after adding the PI compensator is

[0124] G v(s) = K(s)·G vc (s)·G vf (s)·TL(s) (27)

[0125] Among them, TL(s) is the conversion function of the conduction time of the high-frequency switching tube and the switching frequency, which can be calculated according to Equations (16)-(21) and is shown in Equation (28).

[0126]

[0127] In the single-phase rectifier circuit, there is a 100Hz voltage ripple on the DC side. Therefore, it is advisable to set the cut-off frequency at 50Hz and the phase margin at 45 degrees. The parameters of the voltage controller can be calculated through Equation (29):

[0128]

[0129] Among them, ω is the cut-off frequency and γ is the phase margin.

[0130] Substituting the known design parameters, the control parameters of the voltage controller can be obtained.

[0131] The constant-conduction-time current-critical continuous control method of the Boost-type power factor correction circuit is realized through a digital control chip. According to Figure 1 、 Figure 2 and Figure 3 the schematic diagram, Equations (16)-(21), Equations (24)-(29), and the design parameters determined in the simulation, write the control algorithm program in Visual Studio, build the simulation model in the PSIM software, and connect the simulation model with the program in Visual Studio using the dynamic link library to realize the constant-conduction-time current-critical continuous control method of the Boost-type power factor correction circuit.

[0132] As known from the above description, the described turn-off time calculator 6 is characterized in that, according to the conduction time signal t on of the high-frequency switching tube output by the voltage controller 3, the volt-second balance principle is applied to the energy storage inductor in the rectifier to calculate the turn-off time signal t off of the high-frequency switching tube, without the need for an additional current zero-crossing detection circuit.

[0133] The described PFM modulator 7 is characterized in that the conduction time signal t on of the high-frequency switching tube output by the voltage controller 3 and the calculated switching period signal T s of the high-frequency switching tube are jointly fed into the PFM modulator 7, and then the PFM modulator 7 outputs the pulse control signal d of the high-frequency switching tube with an equivalent duty cycle.

[0134] The constant-on-time current critical continuous control method of the described Boost-type power factor correction circuit is characterized in that a constant-on-time CRM generator 4 is constituted by an off-time calculator 6 and a PFM modulator 7. A pulse control signal d of a high-frequency switching tube is generated by the constant-on-time CRM generator 4, and after passing through a drive circuit 5, a drive signal is generated to control the on-off of the high-frequency switching tube in the rectifier, so as to control the current of the energy storage inductor in the rectifier to work in the critical continuous mode.

[0135] A specific embodiment of the present invention is as follows:

[0136] The main circuit parameters of the single-phase full-bridge rectifier are as follows: the energy storage inductor L = 300 μH, the output capacitor Co = 660 μF, the output DC voltage Vo = 390 V, the input AC power supply AC: v in = 230sin50t (V), the load resistance R = 190 Ω. The maximum output power P o = 800 W, the switching frequency range f = 20 - 200 kHz. The high-frequency switching tubes S1 and S2 both adopt IPW65R080CFD of Infineon Technologies AG, and the industrial frequency rectifier diodes D1 and D2 both adopt DSI45-08A of YXYS Company. The operational amplifier in the sampling conditioning circuit adopts LMV612 of Texas Instruments Incorporated, and the digital control chip adopts TMS320F280049 of Texas Instruments Incorporated. The sampling conditioning circuit converts the collected output voltage signal into a signal that can be accepted by the CPU and sends it to the program processing unit in the CPU for processing. The fixed on-time of the high-frequency switching tubes S1 and S2 is obtained from the output of the voltage controller in the program. The transfer function of the off-time, switching period, frequency to the output voltage, the conversion function of the on-time of the high-frequency switching tube and the switching frequency, and the control parameters of the voltage controller can be obtained according to Equation (16), Equation (17), Equation (24), Equation (28) and Equation (29) respectively.

[0137] The first stage: Parameter tuning of the voltage controller is carried out in Mathcad software. Add a PI compensator and set the parameters as follows: The voltage loop sampling gain k in Equation (25) vf Converted to the DSP, take k vf = 1.856×10 -3 , set the voltage loop in the DSP to be interrupted at 10 kHz, then T s = 1×10 -4 .

[0138] Then, according to Equation (29), the calculated value of the control parameter is

[0139]

[0140] The value of the control parameter converted to the DSP is

[0141]

[0142] Among them, is the sampling frequency of the voltage loop, that is, 10 kHz.

[0143] The Bode plot of the system after adding the PI compensator is as Figure 4 shown. It can be seen from Figure 4 that the system is stable. The second stage: According to the determined design parameters, build a simulation model in the PSIM simulation software, write a control program in Visual Studio, and use the dynamic link library to connect the program for simulation. The simulation waveform is as Figures 5-7 shown.

[0144] Figure 5 The simulation waveforms of the input voltage v in and the input current i L of the main circuit of the single-phase full-bridge rectifier are given. The blue is the input voltage waveform, and the red is the input current waveform, indicating that the constant-on-time current critical continuous control method of a Boost-type power factor correction circuit of the present invention realizes the function of power factor correction; Figure 6 The simulation waveform of the output voltage V o is given, indicating that the constant-on-time current critical continuous control method of a Boost-type power factor correction circuit of the present invention realizes a stable DC voltage at the output of the rectifier; Figure 7 The partial enlarged waveform diagram of the input current i L and the drive PFM1B of the high-frequency switch tube S2 within a positive half-cycle of the AC input is given, indicating that the constant-on-time current critical continuous control method of a Boost-type power factor correction circuit of the present invention realizes the constant-on-time control of the high-frequency switch tube, and the input current always operates in the critical continuous mode.

[0145] From the above description, a constant-on-time current critical continuous control method for a Boost-type power factor correction circuit of the invention has the following advantages:

[0146] (1) A single-phase full-bridge topology is used to form the rectifier. Compared with the traditional bridge topology, the single-phase full-bridge topology requires fewer devices, has a high device utilization rate, and has low common-mode noise;

[0147] (2) A single voltage loop controller is used to ensure the stability of the output voltage. Compared with the double closed-loop control, the single voltage loop control has a simple structure and high cost performance;

[0148] (3) By using the volt-second balance principle, the circuit always operates in the critical conduction mode, rather than adding an additional current zero-crossing detection circuit, which reduces the circuit complexity and is beneficial to improving the reliability;

[0149] (4) The circuit operates under variable frequency conditions. Compared with traditional fixed-frequency control, variable-frequency operation is more flexible and can better meet the future development needs of this field.

[0150] (5) Digital control is adopted, and the algorithm can be implemented by a low-cost digital control chip, featuring high cost performance.

[0151] The description of the above embodiments is only used to help understand the method and its core idea of the present invention. It should be noted that for those of ordinary skill in the art of this technology, without departing from the principle of the present invention, several improvements and modifications can be made to the present invention, and these improvements and modifications also fall within the protection scope of the claims of the present invention.

Claims

1. A constant-on-time current critical continuous control method for a Boost-type power factor correction circuit, characterized in that: 1) The sampling signal of the output voltage of the single-phase full-bridge rectifier and the given reference voltage are jointly sent to an adder, and the voltage error signal generated after passing through the adder is sent to a voltage controller; 2) The output signal of the voltage controller is sent to a constant-on-time CRM generator, and the constant-on-time CRM generator generates a pulse control signal for the high-frequency switching tube with an equivalent duty cycle. The constant-on-time CRM generator is composed of a turn-off time calculator and a PFM modulator; the output signal of the voltage controller is the turn-on time signal of the high-frequency switching tube in the single-phase full-bridge rectifier. This signal is divided into three paths. The first path is sent to the turn-off time calculator, and the turn-off time signal of the high-frequency switching tube is obtained by the output of the turn-off time calculator. This turn-off time signal and the second path of turn-on time signal are jointly sent to an adder to obtain the switching period signal of the high-frequency switching tube. This switching period signal and the third path of turn-on time signal are jointly sent to the PFM modulator, and the output of the PFM modulator is the output of the constant-on-time CRM generator.

2. The constant-on-time current critical continuous control method for a Boost-type power factor correction circuit according to claim 1, characterized in that: The pulse control signal generated by the PFM modulator generates a drive signal after passing through a drive circuit, which is used to control the on-off of the high-frequency switching tube of the single-phase full-bridge rectifier, so as to control the current on the energy storage inductor in the rectifier to always work in the critical continuous mode, so that the single-phase full-bridge rectifier outputs a stable DC voltage waveform.

3. The constant-on-time current critical continuous control method for a Boost-type power factor correction circuit according to claim 1 or 2, characterized in that: The turn-off time calculator calculates the turn-off time signal of the high-frequency switching tube according to the turn-on time signal of the high-frequency switching tube output by the voltage controller, and applies the volt-second balance principle to the energy storage inductor in the rectifier.

4. The constant on-time current critical continuous control method for the Boost-type power factor correction circuit according to claim 1 or 2, characterized in that: The transfer function of the switching frequency f of the high-frequency switching tube of the single-phase full-bridge rectifier and the output voltage V o is as follows: Among them, P o is the output power of the rectifier, v in_rms is the effective value of the AC input voltage, I L is the inductor current of the rectifier, L is the rectifier inductor, C is the rectifier output capacitor, and R is the rectifier equivalent load.

5. A constant-on-time current critical continuous control method for a Boost-type power factor correction circuit according to claim 4, characterized in that: The transfer function of the output voltage sampling conditioning circuit is: where k vf is the sampling gain of the voltage loop, and T sp is the sampling period.

6. The constant-on-time current critical continuous control method for a Boost-type power factor correction circuit according to claim 5, characterized in that: The transfer function of the voltage controller is: where k pv and k iv are two control parameters of the PI compensator.

7. The constant-on-time current critical continuous control method of a Boost-type power factor correction circuit according to claim 6, characterized in that: Conduction time t of the high-frequency switching transistor on Conversion function with the switching frequency f: Among them, t off is the turn-off time of the high-frequency switching transistor, and v in is the input AC power supply voltage.

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