A variable-frequency asymmetric control method under light load of a full-bridge LLC

Abstract

The application relates to a variable-frequency asymmetric control method of a full-bridge LLC under light load, establishes a full-bridge bidirectional LLC resonant converter topological structure, establishes a total power loss model of an LLC resonant circuit based on the topological structure, takes the duty ratio of the switch tube of the upper bridge arm in the primary side full-bridge inverter circuit as the optimal duty ratio, the switch tube of the lower bridge arm in the primary side full-bridge inverter circuit is complementary conduction, the optimal duty ratio under the light load working condition is calculated based on the minimum total power loss, and the output voltage is adjusted by using the variable-frequency control on the basis of the preset optimal duty ratio. The method has high efficiency under the light load condition, significantly reduces the switching frequency under the light load, and the asymmetric conduction mode can avoid the influence of the direct current bias caused by the variable duty ratio control method in the prior art, and eliminates the freewheeling process of the body diode.

Description

Technical Field

[0001] This invention relates to the field of power electronics technology, specifically to a variable frequency asymmetric control method for a full-bridge LLC under light load. Background Technology

[0002] LLC resonant converters are favored by industry professionals due to their high efficiency, wide soft-switching range, and high power density, and are widely used in electric vehicle charging piles, switching power supplies, photovoltaic energy storage, and other industries. However, according to the working principle and gain characteristics of LLC resonant converters, the gain curve is easily affected by circuit parasitic parameters under light load output, resulting in a non-monotonic gain curve. This phenomenon can affect the loop regulation and control when using only frequency modulation control.

[0003] To address the aforementioned issues, some scholars have proposed a hybrid control method combining frequency conversion and phase shifting (e.g., YKLo, CYLin, MTHsieh, and CYLin, "Phase-shifted full-bridge series-resonant DC-DC converters for wide load variations," IEEE Trans. Ind. Electron., vol. 58, no. 6, pp. 2572–2575, Jun. 2011). This method, under light loads, adjusts the output voltage solely by the duty cycle after the switching frequency reaches its maximum. However, this approach only considers switching losses and output voltage regulation, neglecting key losses such as core losses and conduction losses in magnetic components. Other scholars have proposed a hybrid frequency-duty cycle modulation technique (e.g., Abhishek Awasthi, Snehal Bagawade, and Praveen K. Jain. Analysis of a Hybrid Variable-Frequency-Duty-Cycle-Modulated Low-Q LLC Resonant Converter for Improving the Light-Load Efficiency for a Wide Input Voltage Range. IEEE TRANSACTIONS ONPOWER ELECTRONICS, VOL.36, NO.7, JULY 2021.), which uses time-domain equations to perform a detailed analysis of all circuit losses to ensure that the generated losses are minimized. Although this method can significantly improve the efficiency under light load, the resonant cavity voltage under this method has a DC bias, which can easily lead to transformer core saturation and failure to work. To avoid transformer magnetization, some researchers have proposed a finite bipolar control strategy (e.g., Chen Tianjin, Cao Ya, Cao Zhihui, et al. Optimization of light-load ripple in LLC resonant converter based on composite control [J]. Electrical Drive, 2021, 51(8):34-39. DOI:10.19457 / j.1001-2095.dqcd22371.). This strategy involves controlling the upper and lower switches in the full-bridge arm to operate symmetrically to prevent transformer magnetization, while the diagonal switches (i.e., the paired switches) are controlled asymmetrically to achieve soft switching, which can reduce output voltage ripple. However, this control method results in a longer freewheeling time for the body diode connected in parallel with the switch, leading to significant diode conduction losses and potentially causing insufficient current to achieve zero-voltage start-up of the switch. Summary of the Invention

[0004] This invention discloses a frequency conversion asymmetric control method for a full-bridge LLC under light load. It mainly adopts a frequency conversion-asymmetric control strategy to enable the resonant circuit to always operate at the optimal duty cycle with the least loss under light load conditions, and achieves output voltage stability through frequency control, which can significantly improve the transmission efficiency of LLC resonant converter under light load conditions.

[0005] To achieve the above objectives, the technical solution adopted by the present invention is as follows:

[0006] A variable frequency asymmetric control method for a full-bridge LLC under light load includes:

[0007] A full-bridge bidirectional LLC resonant converter topology is established, which includes a primary-side full-bridge inverter circuit composed of switches Q1 to Q4, a secondary-side full-bridge rectifier circuit composed of switches Q5 to Q8, an LLC resonant cavity connected to the output of the primary-side full-bridge inverter circuit, and a transformer connecting the primary and secondary circuits.

[0008] A total power loss model for the LLC resonant circuit is established. The duty cycle of the upper bridge arm switch in the primary full-bridge inverter circuit is taken as the optimal duty cycle. The switches of the lower bridge arm in the primary full-bridge inverter circuit are complementary and conduct. The optimal duty cycle under different load conditions is determined based on the minimum total power loss.

[0009] To determine the load condition, when the load is lightly loaded, the switching frequency is obtained by dual-loop modulation based on the DC voltage outer loop and the resonant current inner loop. The obtained switching frequency and the optimal duty cycle corresponding to the load are input to the PWM pulse generator to generate the drive signal for the switching transistor in the primary-side full-bridge inverter circuit.

[0010] Furthermore, the light load condition refers to the situation where the current load is ≤50% of the rated load.

[0011] Furthermore, the optimal duty cycle calculation method under the light load condition is as follows:

[0012] (1) Establish the total power loss model according to the following formula:

[0013] P sum =P Lm.core +P Lr.core +P con +P off +P dr

[0014] In the above formula, P sum For the total power loss, P Lm.core For the transformer core power loss, P Lr.core For the core power loss of the resonant inductor, P con For the conduction power loss of the resonant converter, P offFor the turn-off power loss of the resonant converter, P dr This refers to the drive power loss of the switching transistors in the primary-side full-bridge inverter circuit.

[0015] (2) Based on the total power loss model, calculate the total power loss of the resonant circuit under different duty cycles and different load conditions, and determine the optimal duty cycle based on the principle of minimizing the total power loss under light load conditions.

[0016] Furthermore, the optimal duty cycle is determined by the following formula:

[0017]

[0018] In the above formula, I n I is the rated output current of the LLC resonant circuit. o I is the actual output current of the LLC resonant circuit. o / I n This indicates the load percentage.

[0019] Furthermore, the core power loss of the transformer is as follows:

[0020]

[0021] In the above formula, W tFe.T Let f be the mass of the transformer core iron, k, α, and β be the OSE coefficients determined by the material properties, and f be the mass of the transformer core iron. eq For the equivalent frequency, B Lm f is the transformer magnetic flux density. s Where n is the switching frequency, n is the transformer turns ratio, and V is the voltage. O N is the output voltage of the resonant circuit. p A represents the number of turns in the primary winding of the transformer. e D is the cross-sectional area of ​​the transformer core. opt This represents the duty cycle of the upper bridge arm switching transistor in the primary-side full-bridge inverter circuit.

[0022] The core power loss of the resonant inductor is as follows:

[0023]

[0024] In the above formula, W tFe.Lr For the mass of the resonant inductor core iron, v Lr For the resonant inductor voltage, ΔB Lr N is the peak-to-peak magnetic flux density of the resonant inductor. r A is the number of turns of the resonant inductor. r The cross-sectional area of ​​the resonant inductor;

[0025] The turn-off power loss of the resonant converter is as follows:

[0026]

[0027] In the above formula, P off.13 P represents the turn-off power loss of the upper bridge arm switches Q1 and Q3 in the primary-side full-bridge inverter circuit. off.24 V represents the turn-off power loss of the lower bridge arm switches Q2 and Q4 in the primary-side full-bridge inverter circuit. in For the input voltage, I off.13 I is the turn-off current of switching transistors Q1 and Q3. off.24 t represents the turn-off current of switching transistors Q2 and Q4. off L represents the turn-off time of switching transistors Q1-Q4. m For the magnetizing inductance of the transformer, ω r T is the resonant angular frequency. s θ is the switching period, and θ is the phase of the current lagging behind the output voltage of the primary-side full-bridge inverter circuit.

[0028] The expression for M is:

[0029]

[0030] In the above formula, P O This refers to the output power.

[0031] The power loss of the resonant converter is as follows:

[0032]

[0033] In the above formula, P mos.p.con P ds.con These represent the power loss of the switching transistors in the primary-side full-bridge inverter circuit and the power loss of the body diode in the secondary-side full-bridge rectifier circuit, respectively. Lr.cu P T.p.cu P T.s.cu These represent the copper losses of the resonant inductor, the primary side of the transformer, and the secondary side of the transformer, respectively. mos.con R is the on-state resistance of the switching transistor. Lr R p.w R s.w These are the resonant inductance, the equivalent series resistance of the primary side of the transformer, and the secondary side of the transformer, respectively, V F I is the forward voltage drop of the body diode. r.rms I s.rms These are the effective values ​​of the resonant current and the secondary current, respectively.

[0034] The power loss of the switching transistors in the primary-side full-bridge inverter circuit is as follows:

[0035] P dr =8u gs Q g f s

[0036] In the above formula, u gs Q is the gate-source voltage of the switching transistor. g This represents the total gate charge.

[0037] This invention improves the light-load efficiency of a full-bridge LLC resonant circuit by employing a frequency-conversion asymmetric control strategy. This method uses time-domain analysis to perform precise total circuit loss analysis, pre-calculating the optimal duty cycle of the switching transistors under light-load conditions within the required voltage range to minimize the total power loss of the resonant circuit. Specifically, the duty cycle of the primary-side upper bridge arm switching transistor is set to the optimal duty cycle, while the primary-side lower bridge arm switching transistors are complementary in conduction. Then, based on the preset optimal duty cycle, frequency conversion control is used to adjust the output voltage. Therefore, this method exhibits high efficiency under light-load conditions, significantly reducing the switching frequency at light loads. Furthermore, its asymmetric conduction mode avoids the DC bias effect caused by the variable duty cycle control method in existing technologies and eliminates the freewheeling process of the body diode. Attached Figure Description

[0038] Figure 1 This is the dual-bridge bidirectional LLC resonant converter topology constructed in the embodiment;

[0039] Figure 2 for Figure 1 The circuit waveform diagram of the mid-topology using frequency conversion asymmetric control;

[0040] Figure 3 for Figure 1 The circuit equivalent model of the topology during the t0-t1 time period in the first half of the cycle;

[0041] Figure 4 for Figure 1 The circuit equivalent model of the topology during the t1-t2 time period in the first half of the cycle;

[0042] Figure 5 for Figure 1 The circuit equivalent model of the topology during the t2-t3 time period in the first half of the cycle;

[0043] Figure 6a The total loss curve of a full-bridge LLC resonant circuit under 10% load conditions;

[0044] Figure 6b The total loss curve of a full-bridge LLC resonant circuit under 20% load conditions;

[0045] Figure 6c The total loss curve of a full-bridge LLC resonant circuit under 30% load conditions;

[0046] Figure 7 This is a curve showing the optimal duty cycle for each load condition under rated voltage.

[0047] Figure 8 This is a comparison chart of the efficiency of the variable frequency asymmetric control method of the present invention and the traditional variable frequency control method;

[0048] Figure 9 This is a schematic diagram of the principle structure of the variable frequency asymmetric control method of the present invention. Detailed Implementation

[0049] The technical solutions of the present invention will be clearly and completely described below with reference to the accompanying drawings in the embodiments of the present invention.

[0050] This embodiment proposes a frequency conversion asymmetric control method for a full-bridge bidirectional LLC resonant converter circuit, aiming to improve LLC transmission efficiency and stabilize output voltage under light load conditions. This invention combines primary-side dual-arm asymmetric control with frequency conversion control to form frequency conversion asymmetric control. Specifically: First, a loss model of all components in the LLC resonant circuit is established based on time-domain analysis. Based on this loss model, the relationship between load power loss and duty cycle is obtained, and the optimal duty cycle that minimizes total power loss under various load conditions is pre-calculated. The upper switching transistors of the two primary-side arms have the optimal duty cycle, while the lower switching transistors are complementary. Then, the circuit load condition is determined, the preset optimal duty cycle is selected, and frequency conversion control is used to stabilize the output voltage.

[0051] The topology of the full-bridge bidirectional LLC resonant converter in this embodiment is as follows: Figure 1 As shown, Q1 to Q8 are eight MOSFET switches; Q1 to Q4 form a primary-side full-bridge inverter circuit; and Q5 to Q8 form a secondary-side full-bridge rectifier circuit, converting high-frequency AC voltage into DC output voltage. D1 to D8 are the freewheeling diodes (i.e., body diodes) of Q1 to Q8, respectively. oss1 ~C oss8 The junction capacitances of Q1 to Q8 are respectively; the resonant inductance L r Resonant capacitor C r and transformer magnetizing inductance L m This forms an LLC series-parallel resonant cavity; the resonant cavity is connected between output points A and B of the primary-side full-bridge circuit; the primary and secondary circuits are connected via a high-frequency transformer with a turns ratio of n:1; R o C is the load resistance; f For filtering capacitors; V in V is the input voltage. o V is the output voltage. AB V is the voltage at the midpoint of the two primary arms of the bridge. CD V is the voltage at the midpoint of the two arms of the secondary bridge. Lr V is the voltage across the resonant inductor. Cr V is the voltage across the resonant capacitor. Lm I is the excitation inductor voltage;o For output current; i Lr i is the resonant inductor current; Lm i is the magnetizing inductor current; e This represents the primary winding current of the transformer.

[0052] The key waveforms of the above-mentioned full-bridge bidirectional LLC resonant converter topology under light load conditions with frequency conversion asymmetric control are as follows: Figure 2 As shown, in variable frequency asymmetric control, the duty cycles of switching transistors Q1 and Q3 are D. opt The duty cycles of Q2 and Q4 are 1-D. opt . Figure 2 In the middle, T s For the switching period, V GS.Q1 V GS.Q2 V GS.Q3 V GS.Q4 These are the drive signals for switching transistors Q1 to Q4, respectively.

[0053] To establish a total power loss model for the above topology under light load conditions, it is first necessary to combine... Figure 1 and Figure 2 Modal analysis is performed on the circuit's operation during its cycle. Figure 2 It can be seen that each cycle can be divided into two half-cycles, t0-t3 and t3-t6. Since the working principle of the two half-cycles is symmetrical, only the first half-cycle needs to be discussed. According to the circuit's operating state, the first half-cycle can be divided into three working time periods: [t0-t1], [t1-t2], and [t2-t3]. The equivalent circuit models for these three time periods are as follows: Figures 3-5 As shown.

[0054] Mode 1 [t0-t1]: At t0, Q1 turns on with zero voltage, L r and C r Resonance occurs, and the power supply delivers power to the load via Q1 and Q4, V AB =V in v Lm The clamped position is:

[0055] v Lm (t)=nV o (1)

[0056] Resonant inductor current i Lr and resonant inductor voltage v Lr They are respectively:

[0057]

[0058] v Lr (t)=[V in -nV o -vCr (t0)]cosω r (t-t0)+Z r i Lr (t0)sinω r (t-t0) (3)

[0059] in, It is the resonant angular frequency. Given the characteristic impedance, the resonant current at time t0 is...

[0060] Mode 2 [t1-t2]: At time t1, Q1 is turned off, and the resonant current is supplied to the junction capacitance C of Q1 and Q2 respectively. oos1 C oos2 Charging and discharging, v Lm Clamped to nV o Due to the light load condition v Cr Very small, v Lr Approximately -nV o Therefore, i Lr It decreases rapidly. It is worth noting that during this short period, under light load conditions, the active power transferred from the input side to the output side is very small. Meanwhile, when i Lr Reduce to equal i Lm This phase ends when the time comes.

[0061] Mode 3 [t2-t3]: At t2, Q2 turns on with zero voltage, i Lr equals i Lm All secondary rectifier diodes achieve zero-current turn-off. Since all rectifier diodes are off, no input power is transmitted to the output during this period. m With L r C r Resonance occurs, at which point v Lm for:

[0062]

[0063] Due to v under light load conditions Cr (t) is much smaller than nV o Therefore, during this period v Lm (t) can be ignored. Lr equals i Lm And it remains almost unchanged, with a size of:

[0064]

[0065] The core loss of a magnetic element under non-sinusoidal excitation can be obtained by the modified Steinmetz empirical formula:

[0066]

[0067] Among them, W tFe The mass of the magnetic core is given by B; k, α, and β are OSE coefficients determined by material properties; B m f is the peak magnetic flux density; s f is the switching frequency; eq The equivalent frequency is expressed as follows:

[0068]

[0069] Where dB(t) / dt is the rate of change of magnetic flux density; ΔB is the peak-to-peak magnetic flux density of the magnetic core, and ΔB = B max -B min =2B m .

[0070] The losses of each component are analyzed for the three modes mentioned above as follows:

[0071] The duration of operating mode 2 is typically very short, so its effect can be ignored for ease of analysis. The excitation voltage in mode 3 is extremely small and can be neglected; therefore, the transformer core loss P... Lm.core Only mode 1 needs to be considered. The rate of change of the transformer flux density in mode 1 is:

[0072]

[0073] Among them, B Lm (t) represents the transformer magnetic flux density, N p A represents the number of turns in the primary winding of the transformer. e This represents the cross-sectional area of ​​the transformer core.

[0074] Peak-to-peak magnetic flux density ΔB of the transformer core in Mode 1 Lm for:

[0075]

[0076] The equivalent frequency f of the magnetizing inductor voltage eq for:

[0077]

[0078] Combining equations (6), (9), and (10), the core loss of the transformer can be expressed as:

[0079]

[0080] Among them, W tFe.T This refers to the mass of the transformer core iron.

[0081] The resonant inductor voltage in mode 3 is zero, therefore the core loss P of the resonant inductor is zero. Lr.coreOnly mode 1 needs to be considered. The peak-to-peak magnetic flux density ΔB of the resonant inductor Lr It can be obtained from the following formula:

[0082]

[0083] Among them, I Lr.max N is the peak value of the resonant current. r A is the number of turns of the resonant inductor. r Let be the cross-sectional area of ​​the resonant inductor.

[0084] To determine the peak value of the resonant current, the i in equation (2) within half a switching cycle is... Lr (t) can be rewritten in sinusoidal form by a linear combination of cos and sin:

[0085]

[0086] Where θ is i r Lagging behind u AB phase, I Lr.m This represents the peak value of the resonant current.

[0087] The excitation inductor current during the positive half-cycle is:

[0088]

[0089] At time t0, i.e., when t = 0, i Lr (t0)=i Lm (t0), by combining equations (13) and (14), θ is obtained as:

[0090]

[0091] The converter's operating waveform is symmetrical during the positive and negative half-cycles, i.e., V in The energy transferred to the load is equal in both the positive and negative half-cycles, therefore V in one cycle in Total energy emitted E in for:

[0092]

[0093] Among them, f r It is the resonant frequency.

[0094] Load R within one cycle o Total energy consumed E o for:

[0095] E o =2D opt T s P o (17)

[0096] Among them, Po This refers to the output power.

[0097] According to the law of conservation of energy, V in one period in The total energy emitted is equal to the load R o The total energy consumed, i.e., E in =E o Combining equations (15) to (17), the peak value of the resonant current I can be obtained. Lr.m for:

[0098]

[0099] Where M is expressed as:

[0100]

[0101] Peak resonant current I in mode 1 Lr.max.1 for:

[0102]

[0103] The rate of change of magnetic flux density of a resonant inductor is obtained by the following formula:

[0104]

[0105] The resonant inductor voltage of mode 1 is given by equation (3), therefore, P Lr.core The expression is:

[0106]

[0107] Among them, W tFe.Lr The mass of the resonant inductor core iron.

[0108] Primary-side switches Q1-Q4 can achieve zero-voltage turn-on with no turn-on losses. Therefore, the turn-off losses of these switches are one of the main factors affecting the efficiency of the resonant converter. Under frequency conversion-asymmetric control, the LLC resonant converter operates in the low-frequency region, and secondary-side switches Q5-Q8 turn off with zero current, resulting in no turn-off losses. Therefore, only the turn-off losses of the primary-side switches need to be calculated.

[0109] The turn-off currents of switching transistors Q1 and Q3 are:

[0110]

[0111] The turn-off currents of switching transistors Q2 and Q4 are:

[0112]

[0113] According to equations (23) and (24), the turn-off loss of the converter is:

[0114]

[0115] Among them, P off.13 For the turn-off losses of switching transistors Q1 and Q3, P off.24 For the turn-off losses of switching transistors Q2 and Q4, t off The shutdown time is for Q1-Q4.

[0116] The conduction loss of the converter can be expressed as:

[0117]

[0118] Among them, P mos.p.con P ds.con P represents the conduction losses of the primary-side MOSFET and the secondary-side body diode, respectively. Lr.cu P T.p.cu P T.s.cu These represent the copper losses of the resonant inductor, the primary side of the transformer, and the secondary side of the transformer, respectively. mos.con R is the on-state resistance of the MOSFET. Lr R p.w R s.w These are the resonant inductance, the equivalent series resistance of the primary side of the transformer, and the secondary side of the transformer, respectively, V F I is the forward voltage drop of the body diode. r.rms I s.rms These are the effective values ​​of the resonant current and the secondary current, respectively. Resonant current effective value I r.rms It can be obtained from the following formula:

[0119]

[0120] Because i s =n(i Lr -i Lm i can be obtained. s Effective value I s.rms for:

[0121]

[0122] Primary-side switch drive loss P dr for:

[0123] P dr =8u gs Q g f s (29)

[0124] Among them, u gs Q is the gate-source voltage. g This represents the total gate charge.

[0125] Based on the above analysis, the total loss of the resonant circuit can be obtained as the sum of the aforementioned losses, which can be expressed as:

[0126] P sum =P Lm.core +P Lr.core +P con +P off +P dr (30)

[0127] Based on the total power loss model obtained from formula (30) and the main parameters of a specific example 15kW / 720V resonant circuit, the resonant circuit can be adjusted for different duty cycles D. opt The losses under different load conditions while maintaining rated input and output voltage are evaluated. This embodiment mainly analyzes the losses under light load conditions. In this invention, light load refers to a current load ≤ 50% of the rated load. When analyzing different loads, a 15kW / 720V LLC circuit prototype is used as an example to analyze the total power loss as a function of duty cycle under different load conditions. The analysis results show that for the following load conditions: 10%, 20%, 30%, 40%, and 50% load, the corresponding total circuit losses reach their minimum values ​​at duty cycles of 0.1, 0.19, 0.28, 0.436, and 0.5, respectively. Considering that most light load conditions mainly refer to loads of 30% and below, the accompanying drawings only provide curves for 10%, 20%, and 30% loads regarding the total power loss as a function of duty cycle under different load conditions. The results are as follows... Figures 6a to 6c As shown.

[0128] To achieve zero-voltage turn-on (ZVS) of the switching transistor, the duty cycle should be greater than the critical duty cycle D required to achieve ZVS. ZVS :

[0129]

[0130] Based on the above analysis of the total power loss as a function of duty cycle under different load conditions, it was found that the optimal duty cycle is a piecewise linear function with load as the independent variable, such as... Figure 7 As shown, for a 15kW / 720V LLC circuit prototype, under a load of 10%-50%, the optimal duty cycle can be divided into three segments of a linear function. Therefore, the optimal duty cycle of the primary-side switch in the LLC circuit can be determined by the following formula:

[0131]

[0132] In the above formula, I n I is the rated output current of the LLC resonant circuit. o I is the actual output current of the LLC resonant circuit. o / I nThis indicates the load percentage.

[0133] For the AC gain G of the resonant circuit AC for:

[0134]

[0135] in:

[0136] According to equation (33), the AC gain is related to the switching frequency and the duty cycle. When the duty cycle is constant, the AC gain decreases as the switching frequency increases. When the switching frequency is constant, the AC gain decreases as the duty cycle decreases. Therefore, when the input and output voltages are constant, i.e., the AC gain is constant, decreasing the duty cycle will decrease the AC gain. To maintain a constant AC gain, the switching frequency needs to be decreased.

[0137] Therefore, based on the above analysis, and according to the optimal duty cycle and AC gain relationship determined under light load conditions, according to... Figure 9 The control system, as shown, consists of four main parts: an outer loop for DC voltage, an inner loop for resonant current, an optimal duty cycle loop, and a PWM pulse generator. The output current of the circuit determines whether the load is under light load. When the load is under light load, the outer loop for DC voltage and the inner loop for resonant current achieve dual-loop frequency conversion control to obtain the switching frequency and adjust the output voltage. The outer loop for DC voltage uses a proportional-integral (PI) controller, with the input load DC voltage reference value V. o_ref and load DC voltage V o Then, a primary-side resonant current reference value is generated. A PI-based current control loop is used to achieve error-free tracking control of the LLC resonant current to its reference target value, generating the switching frequency. Based on the preset optimal duty cycle and switching frequency of the load current, the input is fed into a PWM pulse generator to generate the drive signal for the primary-side switching transistor, completing the frequency conversion asymmetric control of the resonant circuit.

[0138] To verify the efficiency improvement of the method used in this invention under light load conditions, the efficiency of the method of this invention is compared with that of traditional frequency converter control methods under different load conditions, such as... Figure 8 As shown, it is clear that, especially under 30% load conditions, the method of the present invention has a very significant effect on improving efficiency.

[0139] Although embodiments of the invention have been shown and described, it will be understood by those skilled in the art that various changes, modifications, substitutions and alterations can be made to these embodiments without departing from the principles and spirit of the invention, the scope of which is defined by the appended claims and their equivalents.

Claims

1. A variable frequency asymmetric control method for a full-bridge LLC under light load, characterized in that, include: A full-bridge bidirectional LLC resonant converter topology is established, which includes a primary-side full-bridge inverter circuit composed of switches Q1 to Q4, a secondary-side full-bridge rectifier circuit composed of switches Q5 to Q8, an LLC resonant cavity connected to the output of the primary-side full-bridge inverter circuit, and a transformer connecting the primary and secondary circuits. A total power loss model for the LLC resonant circuit is established. The duty cycle of the upper bridge arm switch in the primary full-bridge inverter circuit is taken as the optimal duty cycle. The switches of the lower bridge arm in the primary full-bridge inverter circuit are complementary and conduct. The optimal duty cycle under different load conditions is determined based on the minimum total power loss. To determine the load condition, when the load is lightly loaded, the switching frequency is obtained based on the dual-loop modulation of the DC voltage outer loop and the resonant current inner loop. The obtained switching frequency and the optimal duty cycle corresponding to the load are input to the PWM pulse generator to generate the drive signal for the switching transistor in the primary-side full-bridge inverter circuit. The light load condition refers to a situation where the current load is ≤50% of the rated load; The optimal duty cycle calculation method under light load conditions is as follows: (1) Establish the total power loss model according to the following formula: In the above formula, P sum For total power loss, P Lm.core This refers to the power loss of the transformer core. P Lr.core The core power loss of the resonant inductor. P con For the conduction power loss of the resonant converter, P off For the turn-off power loss of the resonant converter, P dr This refers to the drive power loss of the switching transistors in the primary-side full-bridge inverter circuit. (2) Based on the total power loss model, calculate the total power loss of the resonant circuit under different duty cycles and different load conditions, and determine the optimal duty cycle based on the principle of minimizing the total power loss under light load conditions; The optimal duty cycle is determined by the following formula: In the above formula, I n This is the rated output current of the LLC resonant circuit. I o This is the actual output current of the LLC resonant circuit. I o / I n This indicates the load percentage.

2. The frequency converter asymmetric control method under light load for a full-bridge LLC according to claim 1, characterized in that: The core power loss of the transformer is as follows: In the above formula, W tFe.T For the mass of the transformer core iron, k , α , β The OSE coefficient is determined by the material properties. f eq For equivalent frequency, B Lm For transformer magnetic flux density, f s For switching frequency, n This refers to the turns ratio of the transformer. V O The output voltage of the resonant circuit. N p This refers to the number of turns in the primary winding of the transformer. A e This is the cross-sectional area of ​​the transformer core. D opt This represents the duty cycle of the upper bridge arm switching transistor in the primary-side full-bridge inverter circuit. The core power loss of the resonant inductor is as follows: In the above formula, W tFe.Lr For the mass of the resonant inductor core iron, v Lr For the resonant inductor voltage, ∆ B Lr The peak-to-peak magnetic flux density of the resonant inductor is... N r The number of turns of the resonant inductor. A r The cross-sectional area of ​​the resonant inductor; The power loss of the resonant converter during shutdown is as follows: In the above formula, P off.13 The upper bridge arm switching transistor in the primary-side full-bridge inverter circuit Q 1. Q 3's turn-off power loss P off.24 The lower bridge arm switching transistor in the primary-side full-bridge inverter circuit Q 2. Q 4. Turn-off power loss V in Input voltage, I off.13 For switching transistors Q 1. Q 3's turn-off current, I off.24 For switching transistors Q 2. Q 4's turn-off current, t off For switching transistors Q 1- Q 4. Shutdown time L m For the magnetizing inductance of the transformer, ω r The resonant angular frequency, T s For the switching cycle, θ The current lags behind the phase of the output voltage of the primary-side full-bridge inverter circuit; in, M The expression is: In the above formula, P O For output power, f r The resonant frequency; The power loss of the resonant converter is as follows: (Official 6) In the above formula, P mos.p.con , P ds.con These represent the power loss of the switching transistors in the primary-side full-bridge inverter circuit and the power loss of the body diodes in the secondary-side full-bridge rectifier circuit, respectively. P Lr.cu , P T.p.cu , P T.s.cu These are the copper losses of the resonant inductor, the primary side of the transformer, and the secondary side of the transformer, respectively. R mos.con The on-state resistance of the switching transistor is... R Lr , R p.w , R s.w These are the resonant inductance, the equivalent series resistance of the primary side of the transformer, and the secondary side of the transformer, respectively. V F The forward voltage drop of the body diode. I r.rms , I s.rms These are the effective values ​​of the resonant current and the secondary current, respectively. The power loss of the switching transistors in the primary-side full-bridge inverter circuit is as follows: In the above formula, u gs This is the gate-source voltage of the switching transistor. Q g This represents the total gate charge.

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