Feedback boost inverter and control method for suppressing torque ripple of brushless DC motor

Abstract

The present invention discloses a feedback boost inverter and control method for suppressing the torque pulsation of a brushless DC motor, and relates to the technical field of brushless DC motor control. The feedback boost inverter for suppressing the torque pulsation of a brushless DC motor includes providing a feedback boost inverter topology with adjustable size and capable of exceeding the power supply voltage. The feedback boost inverter topology includes an electrolytic capacitor, seven diodes, and seven insulated gate bipolar transistors. Compared with a traditional inverter, which only requires an additional insulated gate bipolar transistor, a diode, and an electrolytic capacitor, it has the advantages of simple structure and convenient control. In addition, the energy of the electrolytic capacitor used to raise the bus voltage in this topology comes entirely from the energy fed back by the motor, thereby improving the energy utilization rate of the motor. Based on this feedback boost inverter, a feedback boost inverter control method for suppressing the torque pulsation of a brushless DC motor is proposed, and this control method can achieve good torque pulsation suppression effect over a wide speed range.

Description

Technical Field

[0001] The present invention relates to the technical field of brushless DC motor control, in particular to a feedback boost inverter and a control method for suppressing torque pulsation of a brushless DC motor. Background Art

[0002] Brushless DC motors (BLDCMs) are widely used in aerospace, electric vehicles, medical devices, and household appliances due to their high power density, simple structure, compact size, and high efficiency. However, torque ripple (TRR) is a significant barrier to BLDCM applications in high-precision, high-stability applications. Torque ripple caused by current commutation can reach up to 50% of the average torque. Therefore, suppressing BLDCM commutation torque ripple is a hot topic in research.

[0003] Existing BLDCM commutation torque pulsation suppression methods can be roughly summarized into three categories: (1) Changing the modulation mode: By changing the modulation mode, the pulsation of the non-commutation phase current during commutation is suppressed, thereby suppressing the commutation torque pulsation; (2) Direct torque control: Ignoring the intermediate links and directly controlling the electromagnetic torque of the motor, it has good robustness to the influence of changes in motor parameters during operation; (3) Increasing the bus voltage during commutation: Raising the bus voltage during commutation to reduce the drop amplitude of the non-commutation phase current and thus suppressing the torque pulsation. Most of these methods increase the bus voltage by adding a front-stage converter.

[0004] Existing boost topologies used in BLDCM are mostly Cuk, SEPIC, Buck-Boost and other circuits, which require a large number of switching devices and additional inductors, inevitably significantly increasing the size and cost of the system. Summary of the Invention

[0005] The present invention aims to provide a regenerative boost inverter and control method for suppressing torque ripple in brushless DC motors (BLDCs). Compared to traditional inverters, the inverter requires only one additional insulated gate bipolar transistor (IGBT), one diode, and one electrolytic capacitor, resulting in a simpler structure and easier control. Based on this regenerative boost inverter, the present invention proposes a regenerative boost inverter control method for suppressing torque ripple in brushless DC motors (BLDCs). This method achieves excellent torque ripple suppression across a wide speed range for BLDCMs.

[0006] To solve the above technical problems, the technical solution adopted by the present invention is: a regenerative boost inverter for suppressing torque ripple of a brushless DC motor, including providing a regenerative boost inverter topology with adjustable size and capable of exceeding the power supply voltage. The regenerative boost inverter topology includes an electrolytic capacitor C0, diodes VD0 to VD6, an insulated gate bipolar transistor (IGBT) 0, IGBT 1 and IGBT 2 forming the A-phase bridge arm, and IGBT 3 and IGBT 4 forming the B-phase bridge arm. 4. The fifth insulated gate bipolar transistor VT5 and the sixth insulated gate bipolar transistor VT6 forming the C-phase bridge arm, the emitters of the first insulated gate bipolar transistor VT1, the third insulated gate bipolar transistor VT3 and the fifth insulated gate bipolar transistor VT5 are connected to the collectors of the second insulated gate bipolar transistor VT2, the fourth insulated gate bipolar transistor VT4 and the sixth insulated gate bipolar transistor VT6 respectively, the emitters of the second insulated gate bipolar transistor VT2, the fourth insulated gate bipolar transistor VT4 and the sixth insulated gate bipolar transistor VT6 are connected to the power supply U respectively. dc The negative electrodes are directly connected and anti-parallel connected to the second diode VD2, the fourth diode VD4 and the sixth diode VD6, respectively. The collectors of the first insulated gate bipolar transistor VT1, the third insulated gate bipolar transistor VT3 and the fifth insulated gate bipolar transistor VT5 are connected to the cathode of the zero diode VD0, and the anode of the zero diode VD0 is connected to the power supply U dc The positive electrode of the electrolytic capacitor C0 is connected to the collector of the zero insulated gate bipolar transistor VT0, the cathode of the first diode VD1, the third diode VD3 and the fifth diode VD5, and the negative electrode of the electrolytic capacitor C0 is connected to the power supply U dc The positive electrodes of the zero insulated gate bipolar transistor VT0 are connected, the emitter of the zero insulated gate bipolar transistor VT0 is connected to the cathode of the zero diode VD0, and the anodes of the diode No. 1 VD1, the diode No. 3 VD3 and the diode No. 5 VD5 are connected to the emitters of the first insulated gate bipolar transistor VT1, the third insulated gate bipolar transistor VT3 and the fifth insulated gate bipolar transistor VT5 respectively.

[0007] A further improvement of the technical solution of the present invention is that the capacitance C of the electrolytic capacitor C0 is * satisfy:

[0008]

[0009] Where, I N is the rated current of the brushless DC motor, T c is the switching period; U TH is the loop width of the hysteresis loop control; L is the equivalent inductance of the phase winding, V dcis the power supply voltage, J is the moment of inertia, K e is the opposite potential coefficient.

[0010] A further improvement of the technical solution of the present invention is that the control method of the feedback boost inverter for suppressing the torque pulsation of the brushless DC motor comprises the following steps:

[0011] Step S1: Obtain the rotor position θ of the brushless DC motor through the position sensor. The rotor position θ is passed through the speed calculation unit to obtain the actual mechanical angular velocity ω of the brushless DC motor. m ;

[0012] Step S2: Given mechanical angular velocity and the actual mechanical angular velocity ω m The non-commutation phase current reference value is obtained by making a difference and passing the speed PI controller ASR

[0013] Step S3: input the rotor position θ into the sector determination unit to obtain sector information S;

[0014] Step S4: Sector information S and three-phase current i A 、i B 、i C Input to the phase current selection unit to obtain the actual non-commutation phase current i n_com and the off-phase current i out ;

[0015] Step S5: Set the non-commutation phase current reference value The actual non-commutation phase current i n_com The current PI controller ACR is used to obtain the current-related duty cycle D1, which is added to the back-EMF-related duty cycle D2 to obtain the duty cycle D during the non-commutation period.

[0016] Step S6: The actual mechanical angular velocity ω m Input to the speed high and low state judgment unit to obtain the speed high and low state S ω , S ω 1 means the motor is running in high speed range, S ω 0 means the motor is running in the low speed range;

[0017] Step S7: The electrolytic capacitor voltage U obtained by the voltage sensor is C0 and the speed high and low state S ω Input to the electrolytic capacitor charge and discharge state judgment unit to obtain the electrolytic capacitor charge and discharge state S C , S C 1 means the electrolytic capacitor needs to be charged, S C 0 means the electrolytic capacitor does not need to be charged;

[0018] Step S8: turn off the phase current i out Input to the commutation signal judgment unit to obtain the commutation signal S com , when the phase current i out When it approaches 0, S com If it is 0, it means that the brushless DC motor is in the non-commutation period. Otherwise, S com When it is 1, it indicates that the brushless DC motor is in the commutation period;

[0019] Step S9: Set the duty cycle D during non-commutation period and the speed high and low state S ω , commutation signal S com , electrolytic capacitor charge and discharge status S C Input to the duty cycle conversion unit to obtain the actual duty cycle D required by the insulated gate bipolar transistor state query table * ;

[0020] Step S10: The actual duty cycle D * , electrolytic capacitor charge and discharge status S C , speed high and low state S ω , sector information S, commutation signal S com The actual switching signals of the zero-number insulated gate bipolar transistor VT0 to the sixth-number insulated gate bipolar transistor VT6 are input into the insulated gate bipolar transistor state query table to control the feedback boost inverter to drive the brushless DC motor to operate.

[0021] A further improvement of the technical solution of the present invention is that the calculation formula of the back EMF-related duty cycle D2 in step S5 is as follows:

[0022]

[0023] A further improvement of the technical solution of the present invention is that the speed high / low state judgment unit in step S6 is:

[0024]

[0025] In the formula, k represents the current moment, U C0L For S ω When it is 0, it is the lower limit of the electrolytic capacitor voltage.

[0026] A further improvement of the technical solution of the present invention is that the electrolytic capacitor charge and discharge state judgment unit in step S7 is:

[0027] When S ω When it is 0,

[0028]

[0029] When S ω When 1,

[0030]

[0031] A further improvement of the technical solution of the present invention is that the actual duty cycle D in step S9 is * The calculation formula is:

[0032] When S ω =0, S com =1, D * =2D;

[0033] When S ω =1, S com =0, S C =1,

[0034] Other states, D * =D.

[0035] A further improvement of the technical solution of the present invention is that the insulated gate bipolar transistor state query table in step S10 includes the switching states of the insulated gate bipolar transistor No. 0 VT0 to the insulated gate bipolar transistor No. 6 VT6 in the 6 sectors during the non-commutation period and the 6 sectors during the commutation period when the motor is operating in the high-speed range or the low-speed range and whether the electrolytic capacitor needs to be charged.

[0036] Due to the adoption of the above technical solution, the technical advancements achieved by the present invention are:

[0037] 1. Compared with the boost topology currently used in BLDCM, the regenerative boost inverter topology proposed in this invention has the advantages of simple structure and convenient control. In this topology, the energy of the electrolytic capacitor used to boost the bus voltage comes entirely from the energy fed back by the motor, eliminating the need for an external power supply and improving the energy utilization of the motor. In addition, this control method can effectively suppress torque ripple over a wide speed range.

[0038] 2. The present invention describes a method for suppressing torque pulsation over a wide speed range of a brushless DC motor based on a feedback boost inverter. When the motor is running at low speed, since the bus voltage required for commutation is less than the power supply voltage, the commutation torque pulsation can be suppressed by pulse width modulation. When running at high speed, in order to quickly commutate and maintain the stability of the non-commutation phase current during commutation, a bus voltage higher than the power supply voltage needs to be provided. Therefore, an electrolytic capacitor can be connected in series to the DC power supply side during commutation to raise the bus voltage and thus suppress the commutation torque pulsation. The energy of the electrolytic capacitor used to raise the bus voltage comes entirely from motor feedback, and no additional power supply is required. BRIEF DESCRIPTION OF THE DRAWINGS

[0039] Figure 1 is a schematic diagram of a feedback boost inverter circuit according to an embodiment of the present invention;

[0040] Figure 2 This is a control block diagram of a feedback boost inverter control method for suppressing torque ripple of a brushless DC motor according to an embodiment of the present invention;

[0041] Figure 3 During the non-commutation period in the embodiment of the present invention, when S ω =0, S C = 0, the flow path diagram of the AB phase conduction current, where (a) and (b) are the flow paths of the current when VT1 is turned on and off respectively;

[0042] Figure 4 During the non-commutation period in the embodiment of the present invention, when S ω =0, S C =1, the flow path diagram of the AB phase conduction current, where (a) and (b) are the flow paths of the current when VT1 is turned on and off respectively;

[0043] Figure 5 During the non-commutation period in the embodiment of the present invention, when S ω =1, S C = 0, the flow path diagram of the AB phase conduction current, where (a) and (b) are the flow paths of the current when VT1 is turned on and off respectively;

[0044] Figure 6 During the non-commutation period in the embodiment of the present invention, when S ω =1, S C =1, the flow path diagram of the AB phase conduction current, where (a) and (b) are the flow paths of the current when VT4 is turned on and off respectively;

[0045] Figure 7 During the commutation period in the embodiment of the present invention, when S ω =0, S C = 0, the current flow path diagram of the switch from CB phase conduction to AB phase conduction, where (a) and (b) are the current flow paths when VT1 is turned on and off respectively;

[0046] Figure 8 During the commutation period in the embodiment of the present invention, when S ω =0, S C =1, the schematic diagram of the current flow path when the CB phase is turned on and the AB phase is turned on, where (a) and (b) are the schematic diagrams of the current flow path when VT1 is turned on and off respectively;

[0047] Figure 9 During the commutation period in the embodiment of the present invention, when S ω =1, schematic diagram of the current flow path when the CB phase conducts and the AB phase conducts. DETAILED DESCRIPTION

[0048] In order to enable those skilled in the art to better understand the solutions of the present invention, the technical solutions in the embodiments of the present invention will be clearly and completely described below in conjunction with the drawings in the embodiments of the present invention. Obviously, the embodiments described are only part of the embodiments of the present invention, not all of the embodiments. Based on the embodiments of the present invention, all other embodiments obtained by ordinary technicians in this field without making creative efforts should fall within the scope of protection of the present invention.

[0049] It should be noted that the terms "number one", "number two", etc. in the description and claims of the present invention and the above-mentioned drawings are used to distinguish similar objects and are not necessarily used to describe a specific order or sequence. It should be understood that the numbers used in this way can be interchanged where appropriate, so that the embodiments of the present invention described herein can be implemented in an order other than those illustrated or described herein. In addition, the terms "including" and "having" and any variations thereof are intended to cover non-exclusive inclusions. For example, a process, method, system, product or device that includes a series of steps or units is not necessarily limited to those steps or units explicitly listed, but may include other steps or units that are not explicitly listed or inherent to these processes, methods, products or devices.

[0050] See also Figure 1 A feedback boost inverter for suppressing torque ripple of a brushless DC motor includes a feedback boost inverter topology with adjustable size and capable of exceeding the power supply voltage. The feedback boost inverter topology includes an electrolytic capacitor C0, diodes VD0 to VD6, an insulated gate bipolar transistor VT0, an insulated gate bipolar transistor VT1 and a second insulated gate bipolar transistor VT2 constituting an A-phase bridge arm, an insulated gate bipolar transistor VT3 and a fourth insulated gate bipolar transistor VT4 constituting a B-phase bridge arm, and a fifth insulated gate bipolar transistor VT1 and a second insulated gate bipolar transistor VT2 constituting a C-phase bridge arm. The emitters of the first insulated gate bipolar transistor VT1, the third insulated gate bipolar transistor VT3 and the fifth insulated gate bipolar transistor VT5 are connected to the collectors of the second insulated gate bipolar transistor VT2, the fourth insulated gate bipolar transistor VT4 and the sixth insulated gate bipolar transistor VT6 respectively, and the emitters of the second insulated gate bipolar transistor VT2, the fourth insulated gate bipolar transistor VT4 and the sixth insulated gate bipolar transistor VT6 are connected to the power supply U dcThe negative electrodes are directly connected and anti-parallel connected to the second diode VD2, the fourth diode VD4 and the sixth diode VD6, respectively. The collectors of the first insulated gate bipolar transistor VT1, the third insulated gate bipolar transistor VT3 and the fifth insulated gate bipolar transistor VT5 are connected to the cathode of the zero diode VD0, and the anode of the zero diode VD0 is connected to the power supply U dc The positive electrode of the electrolytic capacitor C0 is connected to the collector of the zero insulated gate bipolar transistor VT0, the cathode of the first diode VD1, the third diode VD3 and the fifth diode VD5, and the negative electrode of the electrolytic capacitor C0 is connected to the power supply U dc The positive electrodes of the zero insulated gate bipolar transistor VT0 are connected, the emitter of the zero insulated gate bipolar transistor VT0 is connected to the cathode of the zero diode VD0, and the anodes of the diode No. 1 VD1, the diode No. 3 VD3 and the diode No. 5 VD5 are connected to the emitters of the first insulated gate bipolar transistor VT1, the third insulated gate bipolar transistor VT3 and the fifth insulated gate bipolar transistor VT5 respectively.

[0051] Among them, the capacitance of electrolytic capacitor C0 is C * satisfy:

[0052]

[0053] Where, I N is the rated current of the brushless DC motor, T c is the switching period; U TH is the loop width of the hysteresis loop control; L is the equivalent inductance of the phase winding, V dc is the power supply voltage, J is the moment of inertia, K e is the opposite potential coefficient.

[0054] See also Figure 2 , which shows a control block diagram of a regenerative boost inverter control method for suppressing torque ripple of a brushless DC motor according to an embodiment of the present invention. The process of suppressing torque ripple involves a speed calculation unit, a speed PI controller ASR, a sector determination unit, a phase current selection unit, a current PI controller ACR, a speed high and low state determination unit, an electrolytic capacitor charge and discharge state determination unit, a commutation signal determination unit, a duty cycle conversion unit, an insulated gate bipolar transistor state lookup table, a regenerative boost inverter, and a BLDCM. The method includes:

[0055] Step S1: Obtain the rotor position θ of the brushless DC motor through the position sensor. The rotor position θ is passed through the speed calculation unit to obtain the actual mechanical angular velocity ω of the brushless DC motor. m ;

[0056] Step S2: Given mechanical angular velocity and the actual mechanical angular velocity ω mThe non-commutation phase current reference value is obtained by making a difference and passing the speed PI controller ASR

[0057] Step S3: Input the rotor position θ into the sector determination unit to obtain sector information S. The drive of the brushless DC motor requires detecting the rotor position and then determining the commutation signal, thereby dividing it into six sectors;

[0058] Step S4: Sector information S and three-phase current i A 、i B 、i C Input to the phase current selection unit to obtain the actual non-commutation phase current i n_com and the off-phase current i out ;

[0059] Step S5: Set the non-commutation phase current reference value The actual non-commutation phase current i n_com The current-related duty cycle D1 is obtained by making a difference and passing it through the current PI controller ACR. The duty cycle D1 is added to the back-EMF-related duty cycle D2 to obtain the duty cycle D during the non-commutation period. The back-EMF-related duty cycle D2 is calculated by the following formula:

[0060]

[0061] Step S6: The actual mechanical angular velocity ω m Input to the speed high and low state judgment unit to obtain the speed high and low state S ω , S ω 1 means the motor is running in high speed range, S ω 0 means the motor is running in the low speed range; the speed high and low state judgment unit is:

[0062]

[0063] Step S7: The electrolytic capacitor voltage U obtained by the voltage sensor is C0 and the speed high and low state S ω Input to the electrolytic capacitor charge and discharge state judgment unit to obtain the electrolytic capacitor charge and discharge state S C , S C 1 means the electrolytic capacitor needs to be charged, S C 0 means the electrolytic capacitor does not need to be charged;

[0064] The electrolytic capacitor charge and discharge status judgment unit is:

[0065] When S ω When it is 0,

[0066]

[0067] In the formula, k represents the current moment, U C0L For S ω When it is 0, it is the lower limit of the electrolytic capacitor voltage;

[0068] When S ω When 1,

[0069]

[0070] Step S8: turn off the phase current i out Input to the commutation signal judgment unit to obtain the commutation signal S com , commutation signal S during commutation com is 1, commutation ends S com Here, the phase current i of the off phase is detected to determine whether the commutation is completed. out When it is close to 0, the commutation is completed. out When it approaches 0, S com If it is 0, it means that the brushless DC motor is in the non-commutation period. Otherwise, S com When it is 1, it indicates that the brushless DC motor is in the commutation period;

[0071] Step S9: Set the duty cycle D during non-commutation period and the speed high and low state S ω , commutation signal S com , electrolytic capacitor charge and discharge status S C Input to the duty cycle conversion unit to obtain the actual duty cycle D required by the insulated gate bipolar transistor state query table * , which satisfies:

[0072] When S ω =0, S com =1,

[0073] D * =2D;

[0074] When S ω =1, S com =0, S C =1,

[0075]

[0076] Other states,

[0077] D * =D;

[0078] Step S10: The actual duty cycle D * , electrolytic capacitor charge and discharge status S C , speed high and low state S ω , sector information S, commutation signal S comThe actual switching signals of the zero-number insulated gate bipolar transistor VT0 to the sixth-number insulated gate bipolar transistor VT6 are input into the insulated gate bipolar transistor state query table to control the feedback boost inverter to drive the brushless DC motor to operate.

[0079] The insulated gate bipolar transistor state query table includes the switching states of the insulated gate bipolar transistor No. 0 VT0 to the insulated gate bipolar transistor No. 6 VT6 in the six sectors during the non-commutation period and the six sectors during the commutation period, when the motor is operating in the high-speed range or the low-speed range and whether the electrolytic capacitor needs to be charged.

[0080] The IGBT state query table is divided into the IGBT state query table during non-commutation and commutation, as shown below:

[0081] Table 1 Insulated gate bipolar transistor state query table during non-commutation period

[0082]

[0083] Table 2 Insulated gate bipolar transistor state query table during commutation

[0084]

[0085] Among them, Ⅰ represents sector Ⅰ, VI→Ⅰ represents sector VI switching to sector Ⅰ, and other similar units have similar diagrams; the seven bits in each sector represent the switching states of the zero-number insulated gate bipolar transistor to the sixth-number insulated gate bipolar transistor in turn, 1 represents on, 0 represents off, D * Indicates the actual duty cycle D * Perform chopping; speed high and low state S ω 0 means the current moment is low speed operation, 1 means high speed operation; electrolytic capacitor charge and discharge state S C 0 indicates that charging is not required at the current moment, 1 indicates that charging is required, and / indicates that the selection of the insulated gate bipolar transistor state query table is not affected.

[0086] The conduction states of the insulated gate bipolar transistors under different circuit modes in the insulated gate bipolar transistor state query table are analyzed. For the insulated gate bipolar transistor state query table during non-commutation, the rotor is located in the first sector, that is, AB conduction, as an example for explanation, and other sectors can be analyzed by analogy.

[0087] When the speed is high or low, S ωWhen it is 0, it means that the current motor is running at low speed, and the PWM-ON modulation mode with the smallest commutation torque pulsation is adopted. Therefore, AB is turned on, that is, the upper tube of phase A is chopped and the lower tube of phase B is constantly turned on. At this time, if the zero-number insulated gate bipolar transistor is turned on, the electrolytic capacitor is discharged. If the zero-number insulated gate bipolar transistor is turned off, the electrolytic capacitor is neither discharged nor charged. Therefore, when the electrolytic capacitor is charged and discharged in the state S C When it is 1, that is, when the electrolytic capacitor needs to be charged, the insulated gate bipolar transistor is in the on state 0|D * 0|01|00; when the electrolytic capacitor is in charge and discharge state S C When it is 0, that is, when the electrolytic capacitor does not need to be charged, the insulated gate bipolar transistor is in the on state of 1|D * 0|01|00; when the speed is high or low, S ω When it is 1, it means that the motor is currently running at high speed. At this time, the motor needs to be charged, that is, the electrolytic capacitor is charged and discharged. C When it is 1, in order to make all the feedback energy of the motor be used to charge the electrolytic capacitor, the H_ON-L_PWM modulation method is adopted. Therefore, the conduction state of the insulated gate bipolar transistor at this time is 0|10|0D * |00; while the motor does not need to be charged, that is, the electrolytic capacitor is in the charge and discharge state S C When it is 0, in order to make the electrolytic capacitor discharge completely during the commutation period, the H_PWM-L_ON modulation method is adopted. Therefore, the on-state of the insulated gate bipolar transistor at this time is 0|D * 0|01|00.

[0088] The insulated gate bipolar transistor state query table during commutation is explained by taking the commutation from sector VI to sector I as an example, and other sectors can be analyzed by analogy.

[0089] When the speed is high or low, S ω When it is 0, it means that the motor is currently running at low speed. At this time, there is no need to increase the bus voltage. It is only necessary to increase the duty cycle during the non-commutation period to 2 times the original duty cycle. Its switching state is consistent with sector I. When the speed is high or low, S ω When it is 1, it means that the motor is currently running at high speed. At this time, the bus voltage needs to be increased and the zero-number insulated gate bipolar transistor needs to be turned on. Therefore, the conduction state of the insulated gate bipolar transistor at this time is 1|10|01|00.

[0090] Furthermore, the duty cycle D * The calculation method is as follows:

[0091] During non-commutation period, when S ω =0, S C = 0, the flow path diagram of the AB phase conduction current is as follows Figure 3As shown, (a) and (b) are schematic diagrams of the current flow path when VT1 is turned on and off, respectively. VT0 is always on, and VT1 is at an actual duty cycle D * Chopping, VT4 is always on, the voltages of the stator winding terminals of phases A and B relative to the negative pole of the power supply are D * (V dc +U C0 ), 0, in S ω = 0 when U C0 <<V dc , so D * (V dc +U C0 ) can be approximated as D * V dc , at this time the phase current i A =-i B =I N , opposite electromotive force e A =-e B =E=K e ω m , the voltage equation of the winding is:

[0092]

[0093] From formula (1), we can see that in order to keep the non-commutation phase current stable, the duty cycle D * satisfy:

[0094]

[0095] During non-commutation period, when S ω =0, S C =1, the flow path diagram of the AB phase conduction current is as follows Figure 4 As shown in the figure, (a) and (b) are schematic diagrams of the current flow path when VT1 is turned on and off, respectively. VT0 is turned off and VT1 is at the actual duty cycle D * Chopping, VT4 is always on, the voltages of the stator winding terminals of phases A and B relative to the negative pole of the power supply are D * V dc , 0, phase current i A =-i B =I N , opposite electromotive force e A =-e B =E, the voltage equation of the winding is:

[0096]

[0097] From formula (3), we can see that in order to maintain the stability of the non-commutation phase current, the actual duty cycle D at this time is * satisfy:

[0098]

[0099] During non-commutation period, when S ω =1, S C = 0, the flow path diagram of the AB phase conduction current is as follows Figure 5 As shown in the figure, (a) and (b) are schematic diagrams of the current flow path when VT1 is turned on and off, respectively. VT0 is turned off and VT1 is at the actual duty cycle D * Chopping, VT4 is always on, the voltages of the stator winding terminals of phases A and B relative to the negative pole of the power supply are D * V dc , 0, phase current i A =-i B =I N , opposite electromotive force e A =-e B =E, the voltage equation of the winding is:

[0100]

[0101] From formula (5), we can see that in order to keep the non-commutation phase current stable, the duty cycle D * satisfy:

[0102]

[0103] During non-commutation period, when S ω =1, S C =1, the flow path diagram of the AB phase conduction current is as follows Figure 6 As shown, (a) and (b) are schematic diagrams of the current flow path when VT4 is turned on and off, respectively. VT0 is turned off, VT1 is constantly on, and VT4 is at an actual duty cycle D * Chopping, the voltages of the A and B phase stator winding terminals relative to the negative pole of the power supply are V dc 、(1-D * )(V dc +U C0 ), phase current i A =-i B =I N , opposite electromotive force e A =-e B =E, the voltage equation of the winding is:

[0104]

[0105] From formula (7), we can see that in order to keep the non-commutation phase current stable, the duty cycle D * satisfy:

[0106]

[0107] During commutation, when S ω =0, S C = 0, the current flow path diagram of the AB phase current flow is as follows: Figure 7 As shown, (a) and (b) are schematic diagrams of the current flow path when VT1 is turned on and off, respectively. VT0 is always on, and VT1 is at an actual duty cycle D * Chopping, VT4 is always on, the voltages of the stator winding terminals of phases A, B, and C relative to the negative pole of the power supply are D * (V dc +U C0 ), 0, 0, in S ω = 0 when U C0 < <V dc , so D * (V dc +U C0 ) can be approximated as D * V dc , opposite electromotive force e A =-e B =e C =E, the voltage equation of the winding is:

[0108]

[0109] From formula (9), we can see that in order to maintain the stability of the non-commutation phase current, the actual duty cycle D at this time is * satisfy:

[0110]

[0111] During commutation, when S ω =0, S C =1, the current flow path diagram of the AB phase current flow is as follows: Figure 8 As shown in the figure, (a) and (b) are schematic diagrams of the current flow path when VT1 is turned on and off, respectively. VT0 is turned off and VT1 is at the actual duty cycle D * Chopping, VT4 is always on, the voltages of the stator winding terminals of phases A, B, and C relative to the negative pole of the power supply are D * V dc , 0, 0, reverse electromotive force e A =-e B =e C =E, the voltage equation of the winding is:

[0112]

[0113] From formula (11), we can see that in order to maintain the stability of the non-commutation phase current, the actual duty cycle D at this time is * satisfy:

[0114]

[0115] During commutation, when S ω =1, the current flow path diagram of the AB phase current flow is as follows: Figure 9 As shown, VT0 is always on, VT1 is always on, VT4 is always on, and the voltages of the stator winding terminals of phases A, B, and C relative to the negative pole of the power supply are V dc +U C0 , 0, 0, reverse electromotive force e A =-e B =e C =E, the voltage equation of the winding is:

[0116]

[0117] Furthermore, the electrolytic capacitor capacitance C * The selection is explained as follows:

[0118] When S ω = 0, the voltage rise and voltage drop of the electrolytic capacitor C0 caused by the charge and discharge of the electrolytic capacitor C0 in a single switching cycle under rated load should be much smaller than the loop width U of the hysteresis control. TH Based on the above charging and discharging equivalent circuit during low-speed operation, the maximum value of the electrolytic capacitor C0 voltage boost ΔU C0L_up , the maximum value of the voltage drop ΔU C0L_down , as shown in formula (14):

[0119]

[0120] During commutation, when S ω =1, the maximum voltage drop ΔU caused by the discharge of electrolytic capacitor C0 under rated load C0H_down Should be smaller than the loop width U of the hysteresis control TH , as shown in formula (15):

[0121]

[0122] When S ω =1 and the motor speed changes, the charge and discharge rate of the electrolytic capacitor C0 should be greater than the rate of change of the expected value of the electrolytic capacitor C0 voltage, as shown in formula (16):

[0123]

[0124] From equations (14) to (16), we can get the electrolytic capacitor capacitance C * The selection method is shown in formula (17):

[0125]

[0126] Where, I N is the rated current of the brushless DC motor, T c is the switching period; U TH is the loop width of the hysteresis loop control; L is the equivalent inductance of the phase winding, V dc is the power supply voltage, J is the moment of inertia, K e is the opposite potential coefficient.

[0127] The feedback boost inverter topology circuit used in the embodiment of the present invention utilizes the motor's own feedback to charge the electrolytic capacitor C0, thereby improving the energy utilization rate of the motor without the need for an external power supply and achieving good torque pulsation suppression effect over a wide speed range.

Claims

1. A feedback boost inverter for suppressing torque ripple of a brushless DC motor, characterized by: The invention provides a feedback boost inverter topology with adjustable size and higher than the power supply voltage, wherein the feedback boost inverter topology includes an electrolytic capacitor C0, diodes VD0 to VD6, an insulated gate bipolar transistor VT0, an insulated gate bipolar transistor VT1 and a second insulated gate bipolar transistor VT2 constituting the A-phase bridge arm, an insulated gate bipolar transistor VT3 and a fourth insulated gate bipolar transistor VT4 constituting the B-phase bridge arm, and an insulated gate bipolar transistor VT5 constituting the C-phase bridge arm. 5 and No. 6 insulated gate bipolar transistor VT6, the emitters of the No. 1 insulated gate bipolar transistor VT1, the No. 3 insulated gate bipolar transistor VT3 and the No. 5 insulated gate bipolar transistor VT5 are respectively connected to the collectors of the No. 2 insulated gate bipolar transistor VT2, the No. 4 insulated gate bipolar transistor VT4 and the No. 6 insulated gate bipolar transistor VT6, and the emitters of the No. 2 insulated gate bipolar transistor VT2, the No. 4 insulated gate bipolar transistor VT4 and the No. 6 insulated gate bipolar transistor VT6 are respectively connected to the power supply U dc The negative electrodes are directly connected and anti-parallel connected to the second diode VD2, the fourth diode VD4 and the sixth diode VD6, respectively. The collectors of the first insulated gate bipolar transistor VT1, the third insulated gate bipolar transistor VT3 and the fifth insulated gate bipolar transistor VT5 are connected to the cathode of the zero diode VD0, and the anode of the zero diode VD0 is connected to the power supply U dc The positive electrode of the electrolytic capacitor C0 is connected to the collector of the zero insulated gate bipolar transistor VT0, the cathode of the first diode VD1, the third diode VD3 and the fifth diode VD5, and the negative electrode of the electrolytic capacitor C0 is connected to the power supply U dc The positive electrodes of the zero insulated gate bipolar transistor VT0 are connected, the emitter of the zero insulated gate bipolar transistor VT0 is connected to the cathode of the zero diode VD0, and the anodes of the diode No. 1 VD1, the diode No. 3 VD3 and the diode No. 5 VD5 are connected to the emitters of the first insulated gate bipolar transistor VT1, the third insulated gate bipolar transistor VT3 and the fifth insulated gate bipolar transistor VT5 respectively.

2. The feedback boost inverter for suppressing torque ripple of a brushless DC motor according to claim 1, characterized in that: The capacitance value of the electrolytic capacitor C0 is C * satisfy: Where, I N is the rated current of the brushless DC motor, T c is the switching period; U TH is the loop width of the hysteresis loop control; L is the equivalent inductance of the phase winding, V dc is the power supply voltage, J is the moment of inertia, K e is the opposite potential coefficient.

3. A control method for a feedback boost inverter for suppressing torque ripple of a brushless DC motor, characterized by: The steps include: Step S1: Obtain the rotor position θ of the brushless DC motor through the position sensor. The rotor position θ is passed through the speed calculation unit to obtain the actual mechanical angular velocity ω of the brushless DC motor. m ; Step S2: Given mechanical angular velocity and the actual mechanical angular velocity ω m The non-commutation phase current reference value is obtained by making a difference and passing the speed PI controller ASR Step S3: input the rotor position θ into the sector determination unit to obtain sector information S; Step S4: Sector information S and three-phase current i A 、i B 、i C Input to the phase current selection unit to obtain the actual non-commutation phase current i n_com and the off-phase current i out ; Step S5: Set the non-commutation phase current reference value The actual non-commutation phase current i n_com The current PI controller ACR is used to obtain the current-related duty cycle D1, which is added to the back-EMF-related duty cycle D2 to obtain the duty cycle D during the non-commutation period. Step S6: The actual mechanical angular velocity ω m Input to the speed high and low state judgment unit to obtain the speed high and low state S ω , S ω 1 means the motor is running in high speed range, S ω 0 means the motor is running in the low speed range; Step S7: The electrolytic capacitor voltage U obtained by the voltage sensor is C0 and the speed high and low state S ω Input to the electrolytic capacitor charge and discharge state judgment unit to obtain the electrolytic capacitor charge and discharge state S C , S C 1 means the electrolytic capacitor needs to be charged, S C 0 means the electrolytic capacitor does not need to be charged; Step S8: turn off the phase current i out Input to the commutation signal judgment unit to obtain the commutation signal S com , when the phase current i out When S is 0, com If it is 0, it means that the brushless DC motor is in the non-commutation period. Otherwise, S com When it is 1, it indicates that the brushless DC motor is in the commutation period; Step S9: Set the duty cycle D during non-commutation period and the speed high and low state S ω , commutation signal S com , electrolytic capacitor charge and discharge status S C Input to the duty cycle conversion unit to obtain the actual duty cycle D required by the insulated gate bipolar transistor state query table * ; Step S10: The actual duty cycle D * , electrolytic capacitor charge and discharge status S C , speed high and low state S ω , sector information S, commutation signal S com The actual switching signals of the zero-number insulated gate bipolar transistor VT0 to the sixth-number insulated gate bipolar transistor VT6 are input into the insulated gate bipolar transistor state query table to control the feedback boost inverter to drive the brushless DC motor to operate.

4. The method for controlling a feedback boost inverter for suppressing torque ripple of a brushless DC motor according to claim 3, wherein: The calculation formula of the back EMF-related duty cycle D2 in step S5 is as follows:

5. The method for controlling a feedback boost inverter for suppressing torque ripple of a brushless DC motor according to claim 3, wherein: The speed high / low state judgment unit in step S6 is: In the formula, k represents the current moment, U C0L For S ω When it is 0, it is the lower limit of the electrolytic capacitor voltage.

6. The method for controlling a feedback boost inverter for suppressing torque ripple of a brushless DC motor according to claim 3, wherein: The electrolytic capacitor charge and discharge state judgment unit in step S7 is: When S ω When it is 0, When S ω When 1, 7. The method for controlling a feedback boost inverter for suppressing torque ripple of a brushless DC motor according to claim 3, wherein: The actual duty cycle D in step S9 * The calculation formula is: When S ω =0, S com =1, D * =2D; When S ω =1, S com =0, S C =1, Other states, D * =D.

8. The method for controlling a feedback boost inverter for suppressing torque ripple of a brushless DC motor according to claim 3, wherein: The insulated gate bipolar transistor state query table in step S10 includes the switching states of the insulated gate bipolar transistors VT0 to VT6 in the six sectors during the non-commutation period and the six sectors during the commutation period when the motor is operating in the high-speed range or the low-speed range and whether the electrolytic capacitor needs to be charged.

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