A feedback boost circuit based on a three-branch current-mode reference voltage source

Abstract

The present invention provides a feedback-boosting circuit based on a three-branch current-mode reference voltage source, comprising a current-replicating circuit and a feedback-boosting circuit. The current-replicating circuit includes three branches that use current mirrors to replicate current, thereby generating proportional currents. The feedback-boosting circuit is connected to the three branches of the current-replicating circuit and is configured to increase the feedback coefficient, thereby reducing the low-frequency noise and offset at the input of an operational amplifier to the reciprocal multiple of the feedback coefficient when equivalently transferred to the output of the operational amplifier. This reduces current disturbances in the first and second branches, thereby weakening the impact of the operational amplifier's offset and low-frequency noise on the reference source output. The present invention alters the circuit structure of a traditional current-mode reference voltage source, increases the feedback coefficient of the feedback network, and attenuates the input offset and low-frequency noise of the operational amplifier when equivalently transferred to its output, thereby improving the output accuracy of the reference source.

Description

Technical Field

[0001] The invention belongs to the technical field of microelectronics, and in particular relates to a feedback boost circuit based on a three-branch current mode reference voltage source. Background Art

[0002] Voltage reference sources are widely used in many analog and mixed-signal circuits. There are two common types of voltage reference sources: voltage-mode reference sources (BGRs) and current-mode reference sources (BGRs).

[0003] refer to Figure 1 , Figure 1 This is a voltage-mode BGR. The advantage of a voltage-mode BGR is its strong drive capability, allowing it to directly drive a load. However, its output is fixed, typically 1.2V or 2.4V. Therefore, its power supply voltage is typically higher than 1.2V. In low-voltage, low-frequency noise applications, the system power supply voltage cannot meet the requirements of 1.2V or above, and a reference voltage below 1V is required.

[0004] refer to Figure 2 As shown, Figure 2 The current mode BGR is used. With the current mode reference, the supply voltage can be reduced to below 1.2V, and an adjustable V below 1V can be obtained. REF However, in current-mode voltage references, the offset and low-frequency noise of the op amp can degrade the reference's output accuracy. To address this issue, auto-zeroing has been proposed, which reduces noise and offset by sampling and neutralizing the signal within two cycles. However, the effects of channel charge injection and clock feedthrough in auto-zeroing can degrade the voltage reference's accuracy. Others have used chopping techniques to reduce noise and offset, but this requires additional oscillators and filters, increasing chip area. Summary of the Invention

[0005] In order to solve the above problems existing in the prior art, the present invention provides a feedback boost circuit based on a three-branch current mode reference voltage source. The technical problem to be solved by the present invention is achieved through the following technical solutions:

[0006] The present invention provides a feedback boost circuit based on a three-branch current mode reference voltage source, comprising: a current replication circuit and a feedback boost circuit;

[0007] The current replication circuit includes three branches, the first branch is a PTAT current generation branch, the second branch is a CTAT current generation branch, and the third branch is a reference voltage source output branch; the three branches use current mirrors to replicate currents to generate proportional currents;

[0008] The feedback boost circuit is connected to the three branches of the current replication circuit and is used to increase the feedback coefficient so that the low-frequency noise and offset at the input end of the operational amplifier are reduced to the inverse multiple of the feedback coefficient when equivalent to the output end of the operational amplifier, thereby reducing the current disturbance of the first and second branches, thereby weakening the impact of the offset and low-frequency noise of the operational amplifier on the output of the reference source.

[0009] Beneficial effects of the present invention:

[0010] The present invention provides a feedback-boosting circuit based on a three-branch current-mode reference voltage source, comprising a current replication circuit and a feedback-boosting circuit. The current replication circuit includes three branches, one for generating PTAT current, one for generating CTAT current, and one for outputting the reference voltage source. The feedback-boosting circuit is configured to increase the feedback coefficient, thereby reducing the low-frequency noise and offset at the input of an operational amplifier to the reciprocal multiple of the feedback coefficient when the output is transferred to the operational amplifier. This reduces current disturbances in the first and second branches, thereby weakening the impact of the operational amplifier's offset and low-frequency noise on the reference source output. The present invention alters the circuit structure of a conventional current-mode reference voltage source, increases the feedback coefficient of the feedback network, and attenuates the input offset and low-frequency noise of the operational amplifier when transferred to its output, thereby improving the output accuracy of the reference source.

[0011] The present invention will be further described in detail below with reference to the accompanying drawings and embodiments. BRIEF DESCRIPTION OF THE DRAWINGS

[0012] Figure 1 Schematic diagram of traditional voltage-mode BGR;

[0013] Figure 2 Schematic diagram of traditional current mode BGR;

[0014] Figure 3 It is the feedback model of the traditional current mode BGR;

[0015] Figure 4 This is a structural block diagram of a feedback boost circuit based on a three-branch current mode reference voltage source of the present invention;

[0016] Figure 5 A detailed structural diagram of a feedback boost circuit based on a three-branch current mode reference voltage source according to the present invention;

[0017] Figure 6 This is a structural diagram of a first feedback boost circuit based on a three-branch current mode reference voltage source of the present invention;

[0018] Figure 7 This is a structural diagram of a second feedback boost circuit based on a three-branch current mode reference voltage source of the present invention;

[0019] Figure 8 For attachment Figure 6 Feedback model of the feedback boost circuit shown;

[0020] Figure 9 For attachment Figure 7 Feedback model of the feedback boost circuit shown;

[0021] Figure 10 For attachment Figure 6 The noise and offset suppression simulation results of the feedback boost circuit shown;

[0022] Figure 11 For attachment Figure 7 The noise and offset rejection simulation results of the feedback boost circuit are shown. DETAILED DESCRIPTION

[0023] The present invention will be further described in detail below with reference to specific examples, but the embodiments of the present invention are not limited thereto.

[0024] To demonstrate the effectiveness of the solution described in the present invention in improving the output accuracy of the reference source, the impact of the offset and low-frequency noise of the operational amplifier in a traditional current-mode voltage reference on the output accuracy of the reference source is quantified and analyzed as follows:

[0025] Will Figure 2 The conventional current mode BGR shown is equivalent to Figure 3 The feedback model shown. Figure 3 The feedback model consists of the operational amplifier OP, the feedback network and the M3, R5, 1 / g mQ2 The output branch consists of three parts. Among them, 1 / g mQ1 represents the equivalent transconductance of Q1 and Q2, 1 / g mQ2 Represents the equivalent transconductance of Q3, and the sizes of PMOS tubes M1 to M3 are equal. OS,OP and V n,OP They represent the input offset voltage and low-frequency noise of the operational amplifier, respectively. When they are transmitted to the output V O When the voltage is 1 / F, it is amplified by 1 / F times, where F represents the feedback coefficient of the feedback network. Taking the offset voltage as an example, it is:

[0026]

[0027] V O Small signal disturbance ΔV at O The disturbance ΔV caused by points A and B A , ΔV B They are:

[0028]

[0029]

[0030] In formulas (2) and (3), g mp represents the transconductance of the current mirrors M1 to M3. The feedback coefficient F of the feedback network is shown in the following formula (4):

[0031]

[0032] The operational amplifier's input offset voltage V OS,OP The reference output voltage error caused by V OS,REF yes:

[0033]

[0034] Similarly, the low-frequency noise V n,OP The reference output voltage error caused by V n,REF yes:

[0035]

[0036] As can be seen from equations (5) to (6), the input offset voltage and low-frequency noise of the operational amplifier are amplified by R5 / R2 when transmitted to the reference source output. This value is usually set to 10, that is, the input offset voltage of the operational amplifier is amplified by 10 times when transmitted to the output. Therefore, in this traditional current-mode BGR, the reference output voltage error caused by the input offset and low-frequency noise of the operational amplifier has a significant impact on the output accuracy. Therefore, the present invention provides a feedback boost circuit based on a three-branch current-mode reference voltage source for suppressing input offset and low-frequency noise. The technical solution of the present invention is described in detail below.

[0037] like Figure 4 As shown, the present invention provides a feedback boost circuit based on a three-branch current mode reference voltage source, comprising: a current replication circuit and a feedback boost circuit;

[0038] The current replication circuit includes three branches, the first branch is a PTAT current generation branch, the second branch is a CTAT current generation branch, and the third branch is a reference voltage source output branch; the three branches use current mirrors to replicate currents to generate proportional currents;

[0039] refer to Figure 4 The PTAT current generating branch of the present invention includes a PMOS tube M1, the CTAT current generating circuit includes a PMOS tube M2, and the reference voltage source output branch includes a PMOS tube M3 and a resistor R5;

[0040] Among them, the gate of the PMOS tube M1, the gate of the PMOS tube M2, and the gate of the PMOS tube M3 are connected, and the source of the PMOS tube M1, the source of the PMOS tube M2, and the source of the PMOS tube M3 are all connected to the power supply V DD The drain of the PMOS tube M1 is connected to the first connection terminal of the feedback boost circuit, the drain of the PMOS tube M2 is connected to the second connection terminal of the feedback boost circuit, the drain of the PMOS tube M3 is connected to the high potential end of the resistor R5, and the other end of the resistor R5 is connected to the third connection terminal of the feedback boost circuit; the high potential end of the resistor R5 is the output end of the reference voltage source output branch, which is used to output the reference voltage.

[0041] The feedback boost circuit is connected to the three branches of the current replication circuit and is used to increase the feedback coefficient so that the low-frequency noise and offset at the input end of the operational amplifier are reduced to the inverse multiple of the feedback coefficient when equivalent to the output end of the operational amplifier, thereby reducing the current disturbance of the first and second branches, thereby weakening the impact of the offset and low-frequency noise of the operational amplifier on the output of the reference source.

[0042] refer to Figure 5 The feedback boost circuit includes: an NPN transistor Q1, an NPN transistor Q2, a resistor R1, a resistor R2, a resistor R3, a resistor R4 and an operational amplifier OP;

[0043] Among them, the collector of the NPN tube Q1 is connected to the low potential end of the resistor R1, the high potential end of the resistor R4 is connected to the high potential end of the resistor R1, the base of the NPN tube Q1 and the low potential end of the resistor R2 are connected to the collector of the NPN tube Q2, the base of the NPN tube Q2, the high potential end of the resistor R3 and the high potential end of the resistor R2 are connected, and the emitter of the NPN tube Q1, the emitter of the NPN tube Q2, the low potential end of the resistor R3, and the low potential end of the resistor R4 are all grounded; the output end of the operational amplifier OP is connected between the gate of the PMOS tube M1 and the gate of the PMOS tube M2, and the two input ends of the operational amplifier OP are connected to the resistor R2 and one end of the resistor R1 in two connection modes.

[0044] There are two ways to connect the operational amplifier OP:

[0045] Solution ①: Reference Figure 6 The non-inverting input terminal of the operational amplifier OP is connected to the collector of the NPN transistor Q2, the inverting input terminal of the operational amplifier OP is connected to the collector of the NPN transistor Q1, the output terminal of the operational amplifier OP is connected to the gate of the PMOS transistor M1, the gate of the PMOS transistor M2, and the gate of the PMOS transistor M3 in the current replication circuit, and the high potential end of the resistor R1 is connected to the high potential end of the resistor R2;

[0046] Solution ②: Reference Figure 7The non-inverting input terminal of the operational amplifier OP is connected to the high potential end of the resistor R2, the inverting input terminal of the operational amplifier OP is connected to the high potential end of the resistor R1, and the output terminal of the operational amplifier OP is connected to the gates of the PMOS tube M1, the gates of the PMOS tube M2, and the gates of the PMOS tube M3 in the current replication circuit.

[0047] The first connection mode increases the feedback coefficient of the feedback boost circuit to Therefore, the offset and low-frequency noise of the op amp are attenuated to their original values ​​when they are equivalent to the output of the reference voltage source. where n represents a positive integer less than 8. The optimal value of n is 6. When it is 6, the output accuracy is improved by about 17 times compared with the traditional structure.

[0048] The second connection mode increases the feedback coefficient of the feedback boost circuit to g mp ×R2×g mQ ×(r op ||r Q ), which causes the offset and low-frequency noise of the op amp to be equivalent to the output of the reference voltage source being attenuated by g mQ (r op ||r Q ) times.

[0049] The two connection methods result in different internal connections of the feedback boost circuit, thus forming two schemes for the feedback boost circuit based on the three-branch current mode reference voltage source of the present invention. The following analyzes the principles of suppressing offset and low-frequency noise of the two schemes.

[0050] For solution ①, take low-frequency noise as an example to analyze the effect of this structure on weakening low-frequency noise and misalignment. Figure 8 The feedback model of solution ① is shown, and the low-frequency noise V n,OP V O The disturbance V n,O for:

[0051]

[0052] Among them, α and η are V O to V Z and V Z to V A,B The gain of α can be expressed as

[0053] α=2g mp ×R EQ (8)

[0054] Among them, R EQ It represents the impedance looking down from the drain of M2. Assuming V Z There is a disturbance ΔVZ , this disturbance produces a current disturbance ΔI Z .So,

[0055] ΔI A =g mQ ×ΔV z (9)

[0056] Among them, ΔI A is the current disturbance flowing through R2. Since the DC currents flowing through Q1 and Q2 are almost the same, their transconductance g mQ equal:

[0057]

[0058] Then the current disturbance flowing through R2 can be expressed as:

[0059]

[0060] V A The voltage disturbance can be expressed as:

[0061] ΔV A =ΔV Z -ΔI A ×R2=(1-lnn)×ΔV Z (12)

[0062] The current disturbance ΔI flowing through R1 B It can be expressed as:

[0063] ΔI B =g mQ ×ΔV A =(lnn / R2)×(1-lnn)×ΔV Z (13)

[0064] Since the values ​​of R1 and R2 are equal, V B The voltage disturbance can be expressed as:

[0065] ΔV B =ΔV Z -ΔI B ×R1=(1-lnn+ln 2 n)×ΔV Z (14)

[0066] Then we can find R EQ :

[0067]

[0068] According to (12) and (14), the gain η is obtained:

[0069]

[0070] Combining (7), (8), (15) and (16), V n,OP V O The disturbance ΔV n,O for:

[0071]

[0072] V n,OP V REF The disturbance is:

[0073]

[0074] Among them, 2-lnn must be greater than 0, that is, n must be less than 8, otherwise the loop will become positive feedback, resulting in unstable reference. In 1 to 7, the larger n is, the greater V n，OP V REF However, when n = 7, 2-ln7 is 0.05, which is a value close to a negative number. This may cause the loop to become a positive feedback loop when the temperature, process, etc. change. Assuming n = 6, (18) becomes:

[0075]

[0076] Similarly, the op amp's V OS，OP V REF The disturbance is:

[0077]

[0078] Comparing equations (19) and (20) with equations (5) and (6), it is found that after adopting the current-mode offset and low-frequency noise suppression technology proposed in this work, the offset and low-frequency noise of the op amp equivalent to the reference voltage output end are reduced to 1 / 17 of the original value.

[0079] For solution ②, take mismatch as an example to analyze the effect of this structure on reducing low-frequency noise and misalignment. Figure 9 The feedback model of solution ② is shown, V O Small signal disturbance ΔV at O The disturbance ΔV at point A A yes:

[0080]

[0081] In formula (21), g mp Represents the transconductance of the current mirror M1 to M3. 1 / g mQ Represents the equivalent transconductance of Q1 and Q2. V O Small signal disturbance ΔV at O The disturbance ΔV at point BB There are two paths that can be solved using the linear superposition principle.

[0082] The first path is V O Directly via M2 to B:

[0083] ΔV B1 =-g mp ×(r Q +R1)||r op ×ΔV O (twenty two)

[0084] In formula (22), r Q is the small signal impedance of Q1 and Q2, r op is the small signal impedance of the current mirror M1~M3. Q , (22) can be simplified as:

[0085] ΔV B1 =-g mp ×r Q ||r op ×ΔV O (twenty three)

[0086] Another path and the disturbance ΔV at C C Regarding, ΔV C for:

[0087] ΔV C =ΔV A -ΔV A g mQ R2(24)

[0088] V O Small signal disturbance ΔV at O Another path of disturbance to B is that V O Through M1 to A, the signal is transferred to C, and then reaches B through Q1:

[0089] ΔV B2 =-g mQ ×(r Q +R1)||r op ×ΔV C

[0090] ≈-g mQ ×r Q ||r op ×ΔV C (25)

[0091] =g mp ×r Q ||r op ×(1-gmQ R2)ΔV o

[0092] In formula (25), since R1 is much smaller than r Q , which is simplified here. The feedback coefficient F of the feedback network is shown in formula (26):

[0093]

[0094] The operational amplifier's input offset voltage V OS,OP The reference output voltage error caused by V OS,REF yes:

[0095]

[0096] Similarly, the low-frequency noise V n,OP The reference output voltage error caused by V n,REF yes:

[0097]

[0098] Due to the use of offset and noise suppression technology, the feedback coefficient F of the feedback network has changed. Comparing equations (27) and (28) with (5) and (6), the offset and noise of the operational amplifier are equivalent to being attenuated at the output by g. mQ (r op ||r Q ) times, that is, the offset and noise of the operational amplifier are suppressed. In order to verify the principle of reducing the low-frequency noise and offset of the reference source proposed in the present invention, this paper simulates and verifies the offset suppression characteristics of the reference source. After the startup is completed, the BGR in the present invention can provide a reference voltage value of 600mV. A 10mV offset is added to the op amp input of the BGR in scheme one and scheme two respectively to simulate random offset, simulate the change of the reference output voltage, and obtain the noise and offset suppression simulation results of the reference source in the two schemes.

[0099] For solution ①, if Figure 10 As shown, the reference voltage is 603.638mV without the offset. After the 10mV offset is introduced, the reference voltage is 600.9612mV, a change of approximately 2.68mV. This indicates that the reference source designed in this invention has a suppression ratio of approximately 3.7x for the internal offset and low-frequency noise of the clamping loop. In a traditional BGR, if a 10mV offset is added to the BGR op amp input, this value is amplified by approximately 10 times. Therefore, offset and noise suppression techniques can mitigate this phenomenon.

[0100] For solution ②, if Figure 11As shown in FIG. 1 , a 10mV offset only causes a 3mV change in the BGR output, meaning the offset at the op amp input is reduced. The effect of the BGR designed in the present invention on low-frequency noise is similar to that of the offset.

[0101] Although the present application is described herein with reference to various embodiments, those skilled in the art will be able to understand and implement other variations of the disclosed embodiments in practicing the claimed application by reviewing the drawings, the disclosure, and the appended claims. In the claims, the word "comprising" does not exclude other components or steps, and "a" or "an" does not exclude a plurality.

[0102] The above is a further detailed description of the present invention in conjunction with specific preferred embodiments, and the specific implementation of the present invention should not be considered to be limited to these descriptions. For those skilled in the art of the present invention, without departing from the concept of the present invention, several simple deductions or substitutions can be made, which should be considered to fall within the scope of protection of the present invention.

Claims

1. A feedback boost circuit based on a three-branch current mode reference voltage source, characterized in that: include: Current replication circuit and feedback boost circuit; The current replication circuit includes three branches, the first branch is a PTAT current generation branch, the second branch is a CTAT current generation branch, and the third branch is a reference voltage source output branch; the three branches use current mirrors to replicate currents to generate proportional currents; The feedback boost circuit is connected to the three branches of the current replication circuit and is used to increase the feedback coefficient so that the low-frequency noise and offset at the input of the operational amplifier are reduced to the reciprocal multiple of the feedback coefficient when equivalent to the output of the operational amplifier, thereby reducing the current disturbance of the first and second branches, thereby weakening the impact of the offset and low-frequency noise of the operational amplifier on the output of the reference source; The PTAT current generating branch includes a PMOS transistor M1, the CTAT current generating branch includes a PMOS transistor M2, and the reference voltage source output branch includes a PMOS transistor M3 and a resistor R5; Among them, the gate of the PMOS tube M1, the gate of the PMOS tube M2, and the gate of the PMOS tube M3 are connected, and the source of the PMOS tube M1, the source of the PMOS tube M2, and the source of the PMOS tube M3 are all connected to the power supply V DD The drain of the PMOS transistor M1 is connected to the first connection terminal of the feedback boost circuit, the drain of the PMOS transistor M2 is connected to the second connection terminal of the feedback boost circuit, the drain of the PMOS transistor M3 is connected to the high potential end of the resistor R5, and the other end of the resistor R5 is connected to the third connection terminal of the feedback boost circuit; the high potential end of the resistor R5 is the output end of the reference voltage source output branch, which is used to output the reference voltage; The feedback boost circuit includes: an NPN transistor Q1, an NPN transistor Q2, a resistor R1, a resistor R2, a resistor R3, a resistor R4 and an operational amplifier OP; Among them, the collector of the NPN tube Q1 is connected to the low potential end of the resistor R1, the high potential end of the resistor R4 is connected to the high potential end of the resistor R1, the base of the NPN tube Q1 and the low potential end of the resistor R2 are connected to the collector of the NPN tube Q2, the base of the NPN tube Q2, the high potential end of the resistor R3 and the high potential end of the resistor R2 are connected, and the emitter of the NPN tube Q1, the emitter of the NPN tube Q2, the low potential end of the resistor R3, and the low potential end of the resistor R4 are all grounded; the output end of the operational amplifier OP is connected between the gate of the PMOS tube M1 and the gate of the PMOS tube M2, and the two input ends of the operational amplifier OP are connected to the resistor R2 and one end of the resistor R1 in two connection modes.

2. A feedback boost circuit based on a three-branch current mode reference voltage source according to claim 1, characterized in that: In the first connection mode, the two input terminals of the operational amplifier OP are connected with the inverting input terminal of the operational amplifier OP connected to the low potential terminal of the resistor R2 , and the non-inverting input terminal of the operational amplifier OP connected to the low potential terminal of the resistor R1 .

3. The feedback boost circuit based on a three-branch current mode reference voltage source according to claim 2, characterized in that: The first connection mode makes the feedback coefficient of the feedback boost circuit increased to , which causes the offset and low-frequency noise of the op amp to be equivalent to the output of the reference voltage source output branch and is attenuated to the original value. ; in, Indicates the transconductance of PMOS tubes M1 to M3, Indicates resistance The resistance value, Represents a positive integer less than 8.

4. The feedback boost circuit based on a three-branch current mode reference voltage source according to claim 1, characterized in that: In the second connection mode, the two input terminals of the operational amplifier OP are connected with the inverting input terminal of the operational amplifier OP connected to the high potential terminal of the resistor R2 , and the non-inverting input terminal of the operational amplifier OP connected to the high potential terminal of the resistor R1 .

5. The feedback boost circuit based on a three-branch current mode reference voltage source according to claim 4, characterized in that: The second connection mode makes the feedback coefficient of the feedback boost circuit increased to , which causes the offset and low-frequency noise of the op amp to be equivalent to the output of the reference voltage source output branch being attenuated by g mQ (r op ||r Q ) times; Among them, g mp Indicates the transconductance of PMOS tubes M1 to M3, r Q is the small signal impedance of Q1 and Q2, 1 / g mQ represents the equivalent transconductance of Q1 and Q2, r op It is the small signal impedance of PMOS tubes M1 to M3, and the symbol || indicates parallel connection.

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