A method, system and medium for suppressing back electromotive force harmonic disturbance of a permanent magnet synchronous motor

Abstract

The application discloses a kind of permanent magnet synchronous motor back electromotive force harmonic disturbance suppression method, system and medium, the permanent magnet synchronous motor back electromotive force harmonic disturbance suppression method of the present application includes based on the working voltage and current of permanent magnet synchronous motor using sliding mode algorithm to obtain the estimated value of back electromotive force, wherein the sliding mode algorithm uses preset sliding mode switching function to reduce the high-frequency buffeting of back electromotive force estimation;The estimated value of back electromotive force is filtered to obtain the fundamental component of back electromotive force;Based on the fundamental component of back electromotive force, the estimation of rotor position angle and speed is carried out by phase-locked loop.The present application aims at inhibiting the harmonic interference of back electromotive force estimation, improves the estimation accuracy of rotor position angle and speed, and does not depend on specific rotor position estimation method, and can be mutually coordinated with existing rotor position estimation method.

Description

Technical Field

[0001] This invention relates to the field of motor control technology, specifically to a method, system, and medium for suppressing back electromotive force harmonic disturbances in a permanent magnet synchronous motor. Background Technology

[0002] Permanent magnet synchronous motors (PMSMs) possess advantages such as high power density, high efficiency, fast torque response, and small size. Furthermore, advancements in permanent magnet materials, high-performance processors, and power electronic components have led to the widespread application of high-speed PMSMs in industries such as industrial equipment manufacturing, aerospace, automotive, and processing. In PMSM control systems, obtaining the rotor position angle and speed is essential. Installing traditional mechanical sensors presents significant drawbacks, and in some special applications, their installation is not permitted. Therefore, by sampling the electrical signals (such as voltage and current) during motor operation and applying them through a specific mathematical model, the rotor's position and speed information can be estimated. This method is collectively known as sensorless technology for PMSMs.

[0003] In position estimation methods based on permanent magnet synchronous motor (PMSM) models, sliding mode algorithms are commonly used to estimate the back electromotive force (EMF) of PMSMs due to their simplicity, robustness to disturbances, and fast dynamic response. Rotor position angle and velocity information are then extracted from the back EMF via a phase-locked loop (PLL). However, due to inverter nonlinearity and interference from spatial magnetic field harmonics, the back EMF estimated by sliding mode contains significant 5th and 7th harmonic components in addition to the fundamental frequency, introducing 6th harmonic pulsations into the estimated rotor position. These 6th harmonic pulsations in position estimation lead to inaccurate decoupling of the motor control magnetic field, increase additional energy losses in the motor, and degrade the performance of sensorless control. Since the direct source of the rotor position harmonic error is harmonic interference from the back EMF, it is necessary to filter the back EMF. Commonly used back EMF filtering methods include multilayer bandpass filters, such as multilayer generalized second-order integrators and multilayer proportional resonant filters. These methods employ multilayer parallel structures, where parameters in each layer influence each other, making parameter tuning difficult and discretization challenging. Other literature has proposed filtering structures based on adaptive learning mechanisms, but these increase computational complexity to some extent. Summary of the Invention

[0004] The technical problem to be solved by the present invention is to provide a method, system and medium for suppressing back electromotive force harmonic disturbances of a permanent magnet synchronous motor, which addresses the above-mentioned problems of the prior art. The present invention aims to suppress harmonic interference of back electromotive force, improve the estimation accuracy of rotor position angle and speed, and does not depend on a specific rotor position estimation method, but can be used in conjunction with existing rotor position estimation methods.

[0005] To solve the above-mentioned technical problems, the technical solution adopted by the present invention is as follows:

[0006] A method for suppressing back electromotive force harmonic disturbances in a permanent magnet synchronous motor includes:

[0007] S101, based on the working voltage and current of the permanent magnet synchronous motor, a sliding mode algorithm is used to obtain the estimated value of the back electromotive force. The sliding mode algorithm uses a preset sliding mode switching function to reduce the high-frequency chattering of the back electromotive force estimation.

[0008] S102, Filter the estimated value of back electromotive force to obtain the fundamental component of back electromotive force;

[0009] S103, the rotor position angle and speed are estimated based on the fundamental component of the back electromotive force through a phase-locked loop.

[0010] Optionally, step S101 includes:

[0011] S201 obtains the three-phase current of the permanent magnet synchronous motor by sampling through the current sensor and transforms it into two-phase current in the stationary two-phase coordinate system;

[0012] S202 uses the two-phase current and voltage in the stationary two-phase coordinate system as input quantities, constructs a sliding mode observer based on the back electromotive force equation, and selects a preset sliding mode switching function. The selected sliding mode surface S is the difference between the two-phase current and its observed value in the stationary two-phase coordinate system. When the sliding mode converges, the estimated value of the back electromotive force in the stationary two-phase coordinate system is obtained.

[0013] Optionally, the preset sliding mode switching function is a hyperbolic tangent function.

[0014] Optionally, the functional expression of the back electromotive force equation in step S202 is:

[0015]

[0016]

[0017] In the above formula, u α and u β R is the operating voltage of the permanent magnet synchronous motor. s L is the resistance of the permanent magnet synchronous motor. d and L q These are the d-axis and q-axis inductances of a permanent magnet synchronous motor, respectively, where p is the differential operator and ω is the inductance. e i represents the rotor speed of the permanent magnet synchronous motor. α and i β Let e ​​be the two-phase current in a stationary two-phase coordinate system. α and e β Let ψ be the back electromotive force in a stationary two-phase coordinate system. f For the permanent magnet flux linkage of a permanent magnet synchronous motor, θ e This represents the rotor angle.

[0018] Optionally, filtering the estimated back EMF in step S102 means filtering out the 5th and 7th harmonic components in the estimated back EMF to obtain the fundamental component of the back EMF.

[0019] Optionally, in step S102, the filter used to filter the estimated back electromotive force is a frequency adaptive comb filter. The center frequency of the frequency adaptive comb filter is the same as the rotor speed of the permanent magnet synchronous motor. The rotor speed of the permanent magnet synchronous motor is fed back to the frequency adaptive comb filter in real time through a phase-locked loop to achieve frequency adaptation.

[0020] Optionally, the transfer function G(s) of the frequency adaptive comb filter is expressed as follows:

[0021]

[0022] In the above formula, Let z be the fundamental component of the back electromotive force in a stationary two-phase coordinate system. αβ Let be the estimated value of the back electromotive force in a stationary two-phase coordinate system, s be the integral operator, e be a constant, and j be the imaginary unit. Let be an estimate of the rotor speed period, and we have: in The rotor speed is fed back to the frequency adaptive comb filter in real time by the phase-locked loop.

[0023] Optionally, the frequency-adaptive comb filter includes:

[0024] The discrete delay module is used to process the estimated value z of the input α-axis back electromotive force. α And the estimated value of the β-axis back electromotive force z β Delay The estimated value of the α-axis back electromotive force z after the delay is obtained by using a linear interpolation method for discrete time-delay sampling. αd And the estimated value of the β-axis back electromotive force z βd ;

[0025] The angle rotation module is used to estimate the delayed α-axis back electromotive force z. αd And the estimated value of the β-axis back electromotive force z βd Multiplying by a Park inverse transform constant coefficient matrix to perform the Park inverse transform and obtain the back electromotive force estimate. and

[0026] The summation module is used to sum the estimated value z of the input α-axis back electromotive force. α and the back electromotive force estimate obtained by the angle rotation module. Add them together, then multiply by 1 / 2 to obtain the fundamental back electromotive force of the α-axis. The estimated value z of the input β-axis back electromotive force β and the back electromotive force estimate obtained by the angle rotation module. Add them together, then multiply by 1 / 2 to obtain the fundamental back electromotive force of the β axis.

[0027] Optionally, the function expression for discrete delay sampling using a linear interpolation method is:

[0028] z αd [kN]=f r ·z α [kN c ]+(1-f r )·z α [kN f ]

[0029] z βd [kN]=f r ·z β [kN c ]+(1-f r )·z β [kN f ]

[0030] In the above formula, z ad [kN]、z βd [kN] is the estimated value of the back electromotive force z. α and back electromotive force estimate z β The value z after linear interpolation at the kNth sampling time. α [kN c ]、z β [kN c [z] is the estimated back electromotive force along the α axis. α The back electromotive force estimate z along the β axis β At the kN c The value at each sampling time, z α [kN f ]、z β [kN f [z] is the estimated back electromotive force along the α axis. α The back electromotive force estimate z along the β axis β At the kN f The value at each sampling time point, where N is the discrete sampling length (which can be a decimal), f r =NN f N f =floor(N), N c =ceil(N)=N f+1; where:

[0031]

[0032] In the above formula, N is the discrete sampling length. T is an estimated value of the rotor speed period. s The sampling period is The rotor speed is fed back to the frequency adaptive comb filter in real time by the phase-locked loop.

[0033] Optionally, step S103 includes:

[0034] S301, the fundamental component of the back electromotive force in the stationary two-phase coordinate system is normalized to obtain the per-unit value of the fundamental component of the α-axis back electromotive force and the per-unit value of the fundamental component of the β-axis back electromotive force, respectively.

[0035] S302, multiply the per-unit value of the fundamental component of the α-axis back EMF and the cosine value of the current output rotor position angle as the first product term, multiply the per-unit value of the fundamental component of the β-axis back EMF and the sine value of the current output rotor position angle as the second product term, multiply the two product terms by -1 respectively, and then add them together to obtain the rotor position angle error;

[0036] S303, the rotor position angle error is passed through the PI controller to obtain the rotor speed estimate, and the rotor speed estimate is integrated to obtain the estimated rotor position angle.

[0037] Furthermore, the present invention also provides a system for suppressing back EMF harmonic disturbances in a permanent magnet synchronous motor, comprising a microprocessor and a memory interconnected thereto, wherein the microprocessor is programmed or configured to perform the method for suppressing back EMF harmonic disturbances in the permanent magnet synchronous motor.

[0038] Furthermore, the present invention also provides a computer-readable storage medium storing a computer program for being programmed or configured by a microprocessor to perform a method for suppressing back EMF harmonic disturbances of the permanent magnet synchronous motor.

[0039] Compared with the prior art, the present invention has the following main advantages:

[0040] 1. This invention includes using a sliding mode algorithm to obtain an estimate of the back electromotive force (EMF) based on the operating voltage and current of a permanent magnet synchronous motor. The sliding mode algorithm employs a preset sliding mode switching function to reduce high-frequency chattering in the back EMF estimation. The estimated back EMF is filtered to obtain the fundamental component of the back EMF. Based on the fundamental component of the back EMF, the rotor position angle and speed are estimated through a phase-locked loop. This method can eliminate high-frequency chattering and harmonic components of the back EMF, and effectively improve the estimation accuracy of the rotor position angle and speed.

[0041] 2. The present invention includes the estimation of rotor position angle and velocity based on the fundamental component of back electromotive force through a phase-locked loop. It does not depend on a specific rotor position estimation method and can be used in conjunction with existing rotor position estimation methods. Attached Figure Description

[0042] Figure 1 This is a schematic diagram of the basic process of the method in an embodiment of the present invention.

[0043] Figure 2 This is a schematic diagram illustrating the control principle of the method in an embodiment of the present invention.

[0044] Figure 3 This is a schematic diagram of the frequency adaptive comb filter proposed in an embodiment of the present invention.

[0045] Figure 4 This is a discrete schematic diagram of the discrete delay module of the frequency adaptive comb filter in an embodiment of the present invention.

[0046] Figure 5 This is the Bode plot corresponding to the transfer function of the frequency adaptive comb filter proposed in the embodiments of the present invention. Detailed Implementation

[0047] like Figure 1 and Figure 2 As shown, the method for suppressing back EMF harmonic disturbances in a permanent magnet synchronous motor in this embodiment includes:

[0048] S101, based on the working voltage and current of the permanent magnet synchronous motor, a sliding mode algorithm is used to obtain the estimated value of the back electromotive force. The sliding mode algorithm uses a preset sliding mode switching function to reduce the high-frequency chattering of the back electromotive force estimation.

[0049] S102, Filter the estimated value of back electromotive force to obtain the fundamental component of back electromotive force;

[0050] S103, the rotor position angle and speed are estimated based on the fundamental component of the back electromotive force through a phase-locked loop.

[0051] like Figure 1 and Figure 2 As shown, step S101 in this embodiment includes:

[0052] S201 obtains the three-phase current of the permanent magnet synchronous motor by sampling through the current sensor and transforms it into two-phase current in the stationary two-phase coordinate system;

[0053] S202 uses the two-phase current and voltage in the stationary two-phase coordinate system as input quantities, constructs a sliding mode observer based on the back electromotive force equation, and selects a preset sliding mode switching function. The selected sliding mode surface S is the difference between the two-phase current and its observed value in the stationary two-phase coordinate system. When the sliding mode converges, the estimated value of the back electromotive force in the stationary two-phase coordinate system is obtained.

[0054] The Clark transformation, which transforms three-phase currents into two-phase currents in a stationary two-phase coordinate system, is called the Clark transformation. The functional expression of the Clark transformation matrix used in the Clark transformation is as follows:

[0055]

[0056] In the above formula, T 3s / 2s This is the Clark transformation matrix.

[0057] In this embodiment, the preset sliding mode switching function is the hyperbolic tangent function, and its function expression is:

[0058]

[0059] In the above formula, F(x) represents the sliding mode switching function, tanh(x) represents the hyperbolic tangent function, m is the coefficient for adjusting the boundary layer of the hyperbolic tangent function, x is the independent variable, and e is a constant.

[0060] In this embodiment, the functional expression of the back electromotive force equation in step S202 is:

[0061]

[0062]

[0063] In the above formula, u α and u β R is the operating voltage of the permanent magnet synchronous motor. s L is the resistance of the permanent magnet synchronous motor. d and L q These are the d-axis and q-axis inductances of a permanent magnet synchronous motor, respectively, where p is the differential operator and ω is the inductance. e i represents the rotor speed of the permanent magnet synchronous motor. α and i β Let e ​​be the two-phase current in a stationary two-phase coordinate system. α and e β Let ψ be the back electromotive force in a stationary two-phase coordinate system. f For the permanent magnet flux linkage of a permanent magnet synchronous motor, θ e This represents the rotor angle.

[0064] In this embodiment, filtering the estimated back electromotive force (EMF) in step S102 refers to filtering out the 5th and 7th harmonic components from the estimated back EMF to obtain the fundamental component of the back EMF. Considering the traditional filterless method, the back EMF z observed by sliding mode... α z β The equation containing the fundamental component and the 5th and 7th harmonics is as follows:

[0065]

[0066] In the above formula, e1 is the amplitude of the fundamental component of the back electromotive force, e5 and e7 are the amplitudes of the 5th and 7th harmonic components of the back electromotive force, respectively, and ω e Let θ0 be the rotor speed of the permanent magnet synchronous motor, θ1 be the initial rotor position angle of the fundamental component, and θ5 and θ7 be the initial position angles of the 5th and 7th harmonic components, respectively. Considering the traditional method of directly transmitting the sliding mode observation back EMF to the phase-locked loop for phase estimation, the equation is as follows:

[0067]

[0068] In the above formula, ε is the rotor position angle error, and θ e θ is the actual rotor position angle. e =ω e t+θ1, For the estimated rotor position angle (the currently output rotor position angle), The initial rotor position angle is the estimated fundamental component. When the system reaches steady state, the phase-locked loop (PLL) estimates the rotational speed, which is then matched with the actual rotational speed. The rotor position error can then be simplified as:

[0069]

[0070] As shown in the above equation, when the 5th and 7th harmonics in the sliding mode estimation of the back EMF are directly used by the phase-locked loop (PLL) to estimate the rotor position without filtering, a 6th harmonic will be introduced into the rotor position error. Since the PI controller in the PLL has low suppression capability for the 6th AC harmonic, when the rotor position error containing the 6th harmonic is used as the input to the PLL, the extracted estimated position will also contain the 6th harmonic. Therefore, the proposed frequency adaptive comb filter suppresses the 5th and 7th harmonics of the back EMF, thereby suppressing the 6th harmonic pulsation of the rotor position angle estimated by the PLL, thus improving the accuracy of rotor position estimation.

[0071] like Figure 1 and Figure 2As shown, in step S102 of this embodiment, the filter used to filter the estimated back electromotive force is a frequency-adaptive comb filter. The center frequency of the frequency-adaptive comb filter is the same as the rotor speed of the permanent magnet synchronous motor. The rotor speed of the permanent magnet synchronous motor is fed back to the frequency-adaptive comb filter in real time through a phase-locked loop to achieve frequency adaptation. In this embodiment, the transfer function G(s) of the frequency-adaptive comb filter is expressed as follows:

[0072]

[0073] In the above formula, Let z be the fundamental component of the back electromotive force in a stationary two-phase coordinate system. αβ Let be the estimated value of the back electromotive force in a stationary two-phase coordinate system, s be the integral operator, e be a constant, and j be the imaginary unit. Let be an estimate of the rotor speed period, and we have: in The rotor speed is fed back to the frequency-adaptive comb filter in real time by the phase-locked loop. Based on the functional expression of the transfer function G(s) of the frequency-adaptive comb filter, the implementation structure of the frequency-adaptive comb filter proposed in this embodiment is as follows: Figure 3 As shown, the frequency adaptive comb filter includes:

[0074] The discrete delay module is used to process the estimated value z of the input α-axis back electromotive force. α And the estimated value of the β-axis back electromotive force z β Delay The estimated value of the α-axis back electromotive force z after the delay is obtained by using a linear interpolation method for discrete time-delay sampling. αd And the estimated value of the β-axis back electromotive force z βd ;

[0075] The angle rotation module is used to estimate the delayed α-axis back electromotive force z. ad And the estimated value of the β-axis back electromotive force z βd Multiplying by a Park inverse transform constant coefficient matrix to perform the Park inverse transform and obtain the back electromotive force estimate. and

[0076] The summation module is used to sum the estimated value z of the input α-axis back electromotive force. α and the back electromotive force estimate obtained by the angle rotation module. Add them together, then multiply by 1 / 2 to obtain the fundamental back electromotive force of the α-axis. The estimated value z of the input β-axis back electromotive force β and the back electromotive force estimate obtained by the angle rotation module. Add them together, then multiply by 1 / 2 to obtain the fundamental back electromotive force of the β axis.

[0077] Existing position harmonic suppression methods are relatively complex, generally using multiple layers of identical structures connected in parallel to filter out the 5th and 7th harmonics of the back electromotive force, with multiple adjustable parameters; while the frequency adaptive comb filter used in this embodiment is a single-layer structure with all filter parameters being fixed constants, making it simple in structure and easy to apply in practice.

[0078] In this embodiment, the function expression for discrete delay sampling using a linear interpolation method is as follows:

[0079] z αd [kN]=f r ·z α [kN c ]+(1-f r )·z α [kN f ]

[0080] z βd [kN]=f r ·z β [kN c ]+(1-f r )·z β [kN f ]

[0081] In the above formula, z αd [kN]、z βd [kN] is the estimated value of the back electromotive force z. α and back electromotive force estimate z β The value z after linear interpolation at the kNth sampling time. α [kN c ]、z β [kN c [z] is the estimated back electromotive force along the α axis. α The back electromotive force estimate z along the β axis β At the kN c The value at each sampling time, z α [kN f ]、z β [kN f [z] is the estimated back electromotive force along the α axis. α The back electromotive force estimate z along the β axis β At the kN f The value at each sampling time point, where N is the discrete sampling length (which can be a decimal), f r =NN f N f =floor(N), Nc =ceil(N)=N f +1; where:

[0082]

[0083] In the above formula, N is the discrete sampling length. T is an estimated value of the rotor speed period. s The sampling period is The rotor speed is fed back to the frequency adaptive comb filter in real time by the phase-locked loop.

[0084] In existing vector control methods, all control algorithms are implemented through discretization. Traditional filtering methods require precise discretization of their high-order frequency domain transfer functions, making their digital implementation complex and increasing computational burden. However, the frequency-adaptive comb filter used in this embodiment, through a delay module and linear interpolation calculations, is easily digitally discretized, resulting in a simple and efficient implementation.

[0085] The functional expression for the constant coefficient matrix of the inverse Park transform is:

[0086]

[0087] In the above formula, P -1 (π / 6) represents the constant coefficient matrix of the inverse Park transformation.

[0088] Figure 4 This is a schematic diagram illustrating the discrete implementation principle of the discrete delay module in this embodiment. The discretization of the delay is equivalent to storing the original sampled signal. The formula for calculating the discrete sampling length of the delay is:

[0089]

[0090] In the above formula, N is the discrete sampling length. T is an estimated value of the rotor speed period. s The sampling period is The rotor speed is fed back to the frequency-adaptive comb filter in real time by the phase-locked loop. Since the discrete sampling length must be an integer, Lagrange linear interpolation is used to process the discrete sampling length N to improve the delay sampling accuracy. The specific calculation formula is as follows:

[0091] z αd [kN]=f r ·z α [kN c ]+(1-f r )·z α [kN f ]

[0092] z βd [kN]=f r ·z β [kN c ]+(1-f r )·z β [kN f ]

[0093] In the above formula, z αd [kN]、z βd [kN] is the estimated value of the back electromotive force z. α and back electromotive force estimate z β The value z after linear interpolation at the kNth sampling time. α [kN c ]、z β [kN c [z] is the estimated back electromotive force along the α axis. α The back electromotive force estimate z along the β axis β At the kN c The value at each sampling time, z α [kN f ]、z β [kN f [z] is the estimated back electromotive force along the α axis. α The back electromotive force estimate z along the β axis β At the kN f The value at each sampling time, where N is the discrete sampling length, f r =NN f N f =floor(N), N c =ceil(N)=N f +1. Figure 5 The Bode plot is the transfer function of the frequency adaptive comb filter in this embodiment. As can be seen from the figure, the frequency adaptive comb filter has no amplitude and phase attenuation at the fundamental frequency, and has notch characteristics at the 5th and 7th harmonic frequencies, which can effectively filter out the harmonic components in the back electromotive force.

[0094] It should be noted that in step S103, the estimation of rotor position angle and velocity based on the fundamental component of the back electromotive force via a phase-locked loop can employ various rotor position angle and velocity estimation methods as needed. For example, as an optional implementation, such as... Figure 1 and Figure 2 As shown, step S103 in this embodiment includes:

[0095] S301, the fundamental component of the back electromotive force in the stationary two-phase coordinate system is normalized to obtain... αThe per-unit values ​​of the fundamental component of the axial back electromotive force and the per-unit values ​​of the fundamental component of the β-axis back electromotive force are expressed as follows:

[0096]

[0097]

[0098] In the above formula, Let be the per-unit value of the fundamental component of the α-axis back electromotive force. Let be the fundamental component of the α-axis back electromotive force. Let be the per-unit value of the fundamental component of the β-axis back electromotive force. This is the fundamental component of the β-axis back electromotive force;

[0099] S302, the first product term is obtained by multiplying the per-unit value of the fundamental component of the α-axis back EMF with the cosine value of the currently output rotor position angle, and the second product term is obtained by multiplying the per-unit value of the fundamental component of the β-axis back EMF with the sine value of the currently output rotor position angle. Then, both product terms are multiplied by -1 and added together to obtain the rotor position angle error, whose function expression is:

[0100]

[0101] In the above formula, ε is the rotor position angle error, and θ e This is the actual rotor position angle. The estimated rotor position angle (the currently output rotor position angle);

[0102] S303, the rotor position angle error is processed by a PI controller to obtain the rotor speed estimate, and its function expression is:

[0103]

[0104] In the above formula, k is the estimated speed of the rotor. p k is the proportional gain of the PI controller. i Let t be the integral control gain of the PI controller, ε be time, and ε be the rotor position angle error. The estimated rotor position angle is obtained by integrating the estimated rotor speed value, and its functional expression is:

[0105]

[0106] In the above formula, This is the estimated rotor position angle.

[0107] In summary, the method for suppressing back EMF harmonic disturbances in a permanent magnet synchronous motor (PMSM) in this embodiment obtains equivalent back EMF information through a sliding mode observer, then uses a frequency-adaptive comb filter to filter out the 5th and 7th harmonics to obtain the fundamental wave of the equivalent back EMF. Finally, a phase-locked loop (PLL) is used to obtain the estimated rotor position and speed, thereby improving the accuracy of rotor position and speed while maintaining good dynamic performance. Compared with existing technologies, this method for suppressing back EMF harmonic disturbances in a PMSM uses a comb filter to filter out back EMF harmonics, which has a simple structure and is easy to discretize. The filtered fundamental wave of the back EMF is used to estimate the rotor position and speed, greatly improving the accuracy of rotor position and speed observation in sensorless control methods for PMSMs. Compared with traditional methods, this method for suppressing back EMF harmonic disturbances in a PMSM is simpler to implement and easier to discretize. By suppressing rotor position harmonics, it further improves the control efficiency and performance of the PMSM.

[0108] Furthermore, this embodiment also provides a system for a method of suppressing back EMF harmonic disturbances in a permanent magnet synchronous motor, including a microprocessor and a memory interconnected thereto, wherein the microprocessor is programmed or configured to execute the method for suppressing back EMF harmonic disturbances in the permanent magnet synchronous motor.

[0109] Furthermore, this embodiment also provides a computer-readable storage medium storing a computer program, which is used to be programmed or configured by a microprocessor to perform the method for suppressing back EMF harmonic disturbances of the permanent magnet synchronous motor.

[0110] Those skilled in the art will understand that embodiments of this application can be provided as methods, systems, or computer program products. Therefore, this application can take the form of a completely hardware embodiment, a completely software embodiment, or an embodiment combining software and hardware aspects. Furthermore, this application can take the form of a computer program product embodied on one or more computer-readable storage media (including, but not limited to, disk storage, CD-ROM, optical storage, etc.) containing computer-usable program code. This application is described with reference to flowchart illustrations and / or block diagrams of methods, apparatus (systems), and computer program products according to embodiments of this application. It will be understood that each block of the flowchart illustrations and / or block diagrams, and combinations of blocks in the flowchart illustrations and / or block diagrams, can be implemented by computer program instructions. These computer program instructions can be provided to a processor of a general-purpose computer, special-purpose computer, embedded processor, or other programmable data processing apparatus to produce a machine, such that the instructions, which execute via the processor of the computer or other programmable data processing apparatus, create a machine for implementing the process. Figure 1 One or more processes and / or boxes Figure 1The computer program instructions may also be stored in a computer-readable storage medium that can direct a computer or other programmable data processing device to operate in a particular manner, such that the instructions stored in the computer-readable storage medium produce an article of manufacture including instruction means, which are implemented in a process Figure 1 One or more processes and / or boxes Figure 1 The functions specified in one or more boxes. These computer program instructions may also be loaded onto a computer or other programmable data processing apparatus to cause a series of operational steps to be performed on the computer or other programmable apparatus to produce a computer-implemented process, thereby providing instructions that execute on the computer or other programmable apparatus for implementing the process. Figure 1 One or more processes and / or boxes Figure 1 The steps of the function specified in one or more boxes.

[0111] The above description is merely a preferred embodiment of the present invention. The scope of protection of the present invention is not limited to the above embodiments. All technical solutions falling within the scope of the present invention's concept are within the scope of protection of the present invention. It should be noted that for those skilled in the art, any improvements and modifications made without departing from the principles of the present invention should also be considered within the scope of protection of the present invention.

Claims

1. A method for suppressing back electromotive force harmonic disturbances in a permanent magnet synchronous motor, characterized in that, include: S101, a sliding mode algorithm is used to obtain the estimated value of the back electromotive force based on the operating voltage and current of the permanent magnet synchronous motor. The sliding mode algorithm employs a preset sliding mode switching function to reduce high-frequency chattering in the back electromotive force estimation. The preset sliding mode switching function is a hyperbolic tangent function, and its function expression is as follows: In the above formula, This represents the sliding mode switching function. Represents the hyperbolic tangent function. m To adjust the coefficients of the hyperbolic tangent function boundary layer, x As the independent variable, e It is a constant; S102, the estimated back EMF is filtered to obtain the fundamental component of the back EMF; the filter used for filtering the estimated back EMF is a frequency-adaptive comb filter, the center frequency of which is the same as the rotor speed of the permanent magnet synchronous motor. The rotor speed of the permanent magnet synchronous motor is fed back to the frequency-adaptive comb filter in real time through a phase-locked loop to achieve frequency adaptation. The transfer function of the frequency-adaptive comb filter is... G ( s The function expression for ) is: In the above formula, Let be the fundamental component of the back electromotive force in a stationary two-phase coordinate system. This is an estimate of the back electromotive force in a stationary two-phase coordinate system. s Here, e is an integral operator, and e is a constant. j The imaginary unit, Let be an estimate of the rotor speed period, and we have: ,in The rotor speed is fed back to the frequency adaptive comb filter in real time by the phase-locked loop; S103, the rotor position angle and speed are estimated based on the fundamental component of the back electromotive force through a phase-locked loop.

2. The method for suppressing back electromotive force harmonic disturbances of a permanent magnet synchronous motor according to claim 1, characterized in that, Step S101 includes: S201 obtains the three-phase current of the permanent magnet synchronous motor by sampling through the current sensor and transforms it into two-phase current in the stationary two-phase coordinate system; S202 uses the two-phase current and voltage in the stationary two-phase coordinate system as input quantities, constructs a sliding mode observer based on the back electromotive force equation, and selects a preset sliding mode switching function. The selected sliding mode surface S is the difference between the two-phase current and its observed value in the stationary two-phase coordinate system. When the sliding mode converges, the estimated value of the back electromotive force in the stationary two-phase coordinate system is obtained.

3. The method for suppressing back electromotive force harmonic disturbances of a permanent magnet synchronous motor according to claim 1, characterized in that, In step S102, filtering the estimated value of the back electromotive force means filtering out the 5th and 7th harmonic components in the estimated value of the back electromotive force to obtain the fundamental component of the back electromotive force.

4. The method for suppressing back electromotive force harmonic disturbances of a permanent magnet synchronous motor according to claim 1, characterized in that, The frequency adaptive comb filter includes: The discrete delay module is used to process the estimated values ​​of the input α-axis back electromotive force separately. and the estimated value of the β-axis back electromotive force Delay The estimated value of the delayed α-axis back electromotive force was obtained by using a linear interpolation method for discrete time-delay sampling. and the estimated value of the β-axis back electromotive force ; An angle rotation module is used to estimate the delayed α-axis back electromotive force. and the estimated value of the β-axis back electromotive force Multiplying by a constant coefficient matrix of the inverse Park transform and performing the inverse Park transform yields the back electromotive force estimate. and ; The summation module is used to sum the estimated value of the input α-axis back electromotive force. and the back electromotive force estimate obtained by the angle rotation module Add them together, then multiply by 1 / 2 to obtain the fundamental back electromotive force of the α-axis. The estimated value of the input β-axis back electromotive force and the back electromotive force estimate obtained by the angle rotation module Add them together, then multiply by 1 / 2 to obtain the fundamental back electromotive force of the β axis. .

5. The method for suppressing back electromotive force harmonic disturbances of a permanent magnet synchronous motor according to claim 4, characterized in that, The function expression for discrete delay sampling using the linear interpolation method is as follows: In the above formula, , back electromotive force estimate and back electromotive force estimate In the The value after linear interpolation at each sampling time point , The back electromotive force estimate along the α axis back electromotive force estimates along the β axis In the The value at each sampling time. , The back electromotive force estimate along the α axis back electromotive force estimates along the β axis In the The value at each sampling time. N The discrete sampling length can be a decimal. , , ;in: In the above formula, N For discrete sampling length, This is an estimate of the rotor speed cycle. The sampling period is The rotor speed is fed back to the frequency adaptive comb filter in real time by the phase-locked loop.

6. The method for suppressing back electromotive force harmonic disturbances of a permanent magnet synchronous motor according to claim 1, characterized in that, Step S103 includes: S301, the fundamental component of the back electromotive force in the stationary two-phase coordinate system is normalized to obtain the per-unit value of the fundamental component of the α-axis back electromotive force and the per-unit value of the fundamental component of the β-axis back electromotive force, respectively. S302, multiply the per-unit value of the fundamental component of the α-axis back EMF and the cosine value of the current output rotor position angle as the first product term, multiply the per-unit value of the fundamental component of the β-axis back EMF and the sine value of the current output rotor position angle as the second product term, multiply the two product terms by -1 respectively, and then add them together to obtain the rotor position angle error; S303, the rotor position angle error is passed through the PI controller to obtain the rotor speed estimate, and the rotor speed estimate is integrated to obtain the estimated rotor position angle.

7. A system for suppressing back electromotive force harmonic disturbances in a permanent magnet synchronous motor, comprising a microprocessor and a memory interconnected thereon, characterized in that, The microprocessor is programmed or configured to perform the method for suppressing back EMF harmonic disturbances of a permanent magnet synchronous motor according to any one of claims 1 to 6.

8. A computer-readable storage medium storing a computer program, characterized in that, The computer program is used to be programmed or configured by a microprocessor to perform the method for suppressing back EMF harmonic disturbances of a permanent magnet synchronous motor according to any one of claims 1 to 6.

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