A control method, device and switching power supply

Abstract

This invention discloses a control method, device, and switching power supply. The control method includes: a timing configuration method for switching transistors Q1 and Q2, comprising configuring Q1 and Q2 to operate in a non-complementary driving sequence, and turning on Q2 only during the period from the first moment after Q1 is turned off in the current operating cycle to the time before Q1 is turned on in the next operating cycle. The first moment is when the primary-side magnetizing inductor current demagnetizes to zero, and the primary-side circuit of the switching power supply begins to resonate from the first moment, with the first voltage at the midpoint of the half-bridge starting to rise and forming an inflection point; and a method for feedback control of the output voltage of the switching power supply, comprising sampling the first voltage at the midpoint of the half-bridge and the second voltage across the resonant capacitor Cr at the inflection point, and performing closed-loop control of the switching power supply based on the first and second voltages. This invention can reduce the cost and design complexity of asymmetric half-bridge flyback resonant converter topology switching power supply, while realizing soft switching of the switching power supply.

Description

TECHNICAL FIELD

[0001] The present application relates to a switching converter, in particular to a control method, device and switching power supply. BACKGROUND

[0002] In recent years, in the application of small power supply, flyback power supply is the mainstream, especially in the current consumer electronics, usually need to be equipped with an AC / DC charger, because flyback power supply is very suitable for small power segment, while naturally provides the isolation effect, is a better choice. However, with the development of technology, the power supply of various trades put forward higher requirements, high efficiency, high power density, miniaturization and so on become the main research topic of power supply industry. However, the hard switching topology of flyback has no advantage in realizing high efficiency and high power density, while the asymmetric half-bridge flyback topology can realize soft switching, that is, zero voltage switching (Zeo-Voltage-Switching), abbreviated as ZVS; zero current switching (Zero-Current-Switching), abbreviated as ZCS. Soft switching technology mainly uses the principle of resonance to make the current (or voltage) in the switching device of the switching converter change according to the sine or quasi-sine law, and when the switching device current naturally goes to zero, the switching device turns off, or when the switching device voltage is zero, the switching device turns on, so that the switching device loss is zero. It can greatly improve the switching frequency, and provides the possibility for high efficiency, high power density and miniaturization of switching power supply.

[0003] At the same time, if you want to detect the output condition, usually use the secondary side feedback (commonly used optical coupling + TL431 feedback, control method), that is, output feedback, which increases the cost of the power supply, the product design is difficult, the isolation withstand voltage is low, and the life of the optical coupling itself will become the bottleneck of the power supply, the disadvantages are obvious. And the primary side feedback has been widely used because it does not need optical coupling and secondary side control.

[0004] In the Chinese patent application document with publication number CN 105375783B, a feedback control method of asymmetric half-bridge flyback converter is proposed based on asymmetric half-bridge flyback topology, which can ensure high output voltage accuracy, linear regulation rate and load regulation rate of the converter; at the same time, it can realize zero voltage switching (ZVS) of the primary side switch, improve the working frequency and efficiency of the converter, and further reduce the size of the whole machine; and the fewer turns of the primary side winding can reduce the coupling capacitance between the primary and secondary sides of the transformer, and improve the isolation withstand voltage between the primary and secondary sides of the converter. The principle diagram is shown in Figure 1 The primary side feedback (PSR) is adopted, the output voltage is fed back through the addition of an additional auxiliary winding Np2, and the output voltage is stabilized. However, this patent increases the auxiliary coil, which increases the cost and design complexity.

[0005] And at present, the most common way to supply power for the control IC is to use an auxiliary winding to provide operating voltage for the IC, such as Figure 2 As shown in the figure, an auxiliary winding Na is additionally added to obtain a suitable control IC supply voltage through the topology and turn ratio relationship, but the supply of the control IC without the auxiliary winding will become a problem.

[0006] Therefore, it is necessary to propose a control method without an auxiliary winding to achieve feedback, so as to simplify the peripheral circuit and further reduce the cost, and to solve the problem of the supply of the control IC after the auxiliary winding is removed. SUMMARY

[0007] Therefore, the technical problem to be solved by the present application is to propose a control method and device and a switching power supply to realize a switching power supply without an auxiliary winding using an asymmetric half-bridge flyback resonant converter topology, reduce the cost and design complexity, ensure high output voltage accuracy, realize zero-voltage turn-on (ZVS) of the primary side switch tube, zero-current turn-off (ZCS) of the secondary side switch tube, improve the working frequency and efficiency of the converter, and reduce the size of the whole machine.

[0008] As a first aspect of the present application, the implementation of the control method is as follows:

[0009] A control method for controlling the conduction and turn-off of a switching tube in a switching power supply, the switching power supply using an asymmetric half-bridge flyback resonant converter topology, including a switching tube Q1 located in the upper bridge arm, a switching tube Q2 located in the lower bridge arm, a resonant inductor Lr and a resonant capacitor Cr, the connection point of the switching tube Q1 and the switching tube Q2 being a half-bridge midpoint, wherein the control method comprises:

[0010] The driving timing configuration method of the switching tube Q1 and the switching tube Q2 includes configuring the switching tube Q1 and the switching tube Q2 to work in a non-complementary driving timing, and starting to turn on the switching tube Q2 only in the time period from the first time point after the switching tube Q1 is turned off in the current working period to the time before the switching tube Q1 is turned on in the next working period, the first time point being the time point when the primary excitation inductance current of the switching power supply demagnetizes to zero, the primary inductor is released, and the switching power supply is not clamped by the output end of the switching power supply, the primary circuit of the switching power supply starts to resonate from the first time point, and the first voltage of the half-bridge midpoint starts to rise to form an inflection point;

[0011] And the method for feedback control of the output voltage of the switching power supply includes taking the inflection point time as the sampling time, sampling the first voltage of the half-bridge midpoint and the second voltage across the resonant capacitor Cr at this time, and performing closed-loop control on the switching power supply according to the first voltage and the second voltage to realize stable output of the switching power supply.

[0012] Further, the driving module starts to turn on the switch tube Q2 when the first voltage of the half-bridge midpoint reaches the bottom of the oscillation.

[0013] As a second aspect of the present application, the control device is implemented as follows:

[0014] A control device for controlling the turn-on and turn-off of switch tubes in a switching power supply, the switching power supply adopting an asymmetric half-bridge flyback resonant converter topology, comprising a switch tube Q1 in the upper bridge arm, a switch tube Q2 in the lower bridge arm, a resonant inductor Lr and a resonant capacitor Cr, the connection point of the switch tube Q1 and the switch tube Q2 being a half-bridge midpoint, wherein the control device comprises:

[0015] a driving module for configuring the driving timing of the switch tube Q1 and the switch tube Q2, including configuring the switch tube Q1 and the switch tube Q2 to work in a non-complementary driving timing, and starting to turn on the switch tube Q2 in a time period from a first time point after the switch tube Q1 is turned off in the current working period to before the switch tube Q1 is turned on in the next working period, the first time point being when the primary excitation inductor current of the switching power supply demagnetizes to zero, the primary inductor is released, and the switching power supply is not clamped by the output end of the switching power supply, at which time the primary circuit of the switching power supply starts to resonate, and the first voltage of the half-bridge midpoint starts to rise to form an inflection point;

[0016] and a feedback module for feedback control of the output voltage of the switching power supply, including taking the inflection point time as a sampling time, sampling the first voltage of the half-bridge midpoint and the second voltage across the resonant capacitor Cr at this time, and performing closed-loop control of the switching power supply according to the first voltage and the second voltage to achieve stable output of the switching power supply.

[0017] Further, the driving module starts to turn on the switch tube Q2 when the first voltage of the half-bridge midpoint reaches the bottom of the oscillation.

[0018] Further, the control device further comprises a voltage conversion circuit for converting the voltage provided by the resonant capacitor Cr as a voltage source into a working voltage required by active devices in the switching power supply.

[0019] Further, the control device is partially or wholly integrated into an IC.

[0020] As a third aspect of the present application, the switching power supply is implemented as follows:

[0021] The application discloses a switching power supply which adopts an asymmetric half-bridge flyback resonant converter topology, and comprises a switching tube Q1 located at an upper bridge arm, a switching tube Q2 located at a lower bridge arm, a resonant inductor Lr and a resonant capacitor Cr, and a connection point of the switching tube Q1 and the switching tube Q2 is a half-bridge midpoint, wherein the switching power supply further comprises the control device according to any one of the second aspect.

[0022] Further, a capacitance Cr of the resonant capacitor Cr, a time ΔT of the switching tube Q2 turned on in a period, an inductance Lr of the resonant inductor Lr and a ratio M comply with the following equation:

[0023]

[0024] Wherein, M is a conduction time ΔT of the switching tube Q2 and a resonant period of the resonant capacitor Cr and the resonant inductor from a first time after the switching tube Q1 is turned off to a time before the switching tube Q1 is turned on in a next working cycle.

[0025] Further, the ratio M is greater than 0.45+ a first set value and less than 0.45- a second set value.

[0026] Preferably, the switching tube Q1 and / or the switching tube Q2 are MOS tubes or GaN devices.

[0027] Compared with the prior art, the application has the following beneficial effects:

[0028] (1) The first voltage of the half-bridge midpoint and the second voltage between the resonant capacitor Cr are sampled for feedback control of the switching power supply, which is a primary side feedback control method without an auxiliary winding, can reduce the cost and design complexity of the switching power supply, and realizes zero-voltage turn-on (ZVS) of the primary side switching tube of the asymmetric half-bridge flyback resonant converter and zero-current turn-off (ZCS) of the secondary side switching tube, improves the frequency and efficiency and reduces the volume.

[0029] (2) The resonant capacitor Cr is further used as a power supply voltage source to provide a working voltage for active devices in the switching power supply, so that the whole control device is simplified and integrated. DETAILED DESCRIPTION

[0030] Figure 1 Figure 1 is a circuit diagram of a switching power supply according to a first embodiment of the application; Figure 5 ;

[0031] Figure 2 Figure 2 is a circuit diagram of a common auxiliary winding power supply circuit;

[0032] Figure 3 Figure 3 is a first circuit diagram of a switching power supply according to a third embodiment of the application;

[0033] Figure 4 For Figure 3 Equivalent circuit diagram of the switching power supply circuit;

[0034] Figure 5 For Figure 3 Voltage and current waveform diagram of the switching power supply circuit related node;

[0035] Figure 6 Second equivalent circuit diagram of the switching power supply of the third embodiment of the present application;

[0036] Figure 7 Third equivalent circuit diagram of the switching power supply of the third embodiment of the present application;

[0037] Figure 8 Fourth equivalent circuit diagram of the switching power supply of the third embodiment of the present application. DETAILED DESCRIPTION

[0038] In order to make the objects, technical solutions and advantages of the present application clearer, the present application will be further described in detail below with reference to the drawings and embodiments. It should be understood that the specific embodiments described herein are only used to explain the present application and not intended to limit the present application.

[0039] It should be noted that the terms "first", "second" and the like in the description and claims of the present application and the above-mentioned drawings are used to distinguish similar objects, and do not necessarily indicate a specific order or sequence. It should be understood that the data thus used can be interchanged under appropriate circumstances, so as to describe the embodiments of the present application. In addition, the terms "comprise" and "have" and any variations thereof are intended to cover non-exclusive inclusion, for example, a process, method, system, product or device comprising a series of steps or units does not necessarily limit to those steps or units clearly listed, but can include other steps or units not clearly listed or inherent to these processes, methods, products or devices.

[0040] In addition, the embodiments in the present application and the features in the embodiments can be combined with each other without conflict.

[0041] It should be understood that in the specification, claims and drawings, when describing a step succeeding to another step, the step can directly succeed to the other step, or succeed to the other step through a third step; when describing an element / unit "connected" to another element / unit, the element / unit can be "directly connected" to the other element / unit, or "connected" to the other element / unit through a third element / unit.

[0042] In addition, the drawings of the present disclosure are only schematic and are not necessarily drawn to scale. Identical or similar components show identical or similar designations throughout the various figures, which are thus omitted from a repeated description. Some of the blocks in the drawings are functional entities that may be implemented using software, or in one or more hardware modules or integrated circuits, or using different networks and / or processor devices and / or microcontroller devices.

[0043] First embodiment

[0044] The present embodiment provides a control method for controlling the turn-on and turn-off of a switching tube in a switching power supply, the switching power supply adopting an asymmetric half-bridge flyback resonant converter topology, including a switching tube Q1 located in the upper bridge arm, a switching tube Q2 located in the lower bridge arm, a resonant inductor Lr and a resonant capacitor Cr, the connection point of the switching tube Q1 and the switching tube Q2 being a half-bridge midpoint SW, the control method comprising:

[0045] The driving timing configuration method of the switching tube Q1 and the switching tube Q2 includes configuring the switching tube Q1 and the switching tube Q2 to work in a non-complementary driving timing, and starting to turn on the switching tube Q2 only in the time period from the first time point after the switching tube Q1 is turned off in the current working period to before the switching tube Q1 is turned on in the next working period, the first time point being the time point when the primary excitation inductor current of the switching power supply demagnetizes to zero, the primary inductor is released, and the switching power supply is not clamped by the output end of the switching power supply, the primary circuit of the switching power supply starts to resonate from the first time point, and the first voltage Vsw of the half-bridge midpoint SW starts to rise to form an inflection point;

[0046] And the method for feedback control of the output voltage of the switching power supply includes taking the inflection point time as a sampling time, sampling the first voltage Vsw of the half-bridge midpoint SW and the second voltage Vcr across the resonant capacitor Cr at this time, and performing closed-loop control on the switching power supply according to the first voltage Vsw and the second voltage Cr to realize the stable voltage output of the switching power supply.

[0047] The resonant inductor Lr can be the leakage inductance of the primary winding of the transformer of the switching power supply, or a specially designed inductor, the resonant capacitor Cr can be arranged in the circuit between the half-bridge midpoint and one end of the primary winding of the transformer of the switching power supply, or arranged in the circuit between the other end of the primary winding of the transformer of the switching power supply and the input ground of the switching power supply, and the values of the resonant inductor Lr and the resonant capacitor Cr can be designed by the person skilled in the art according to the circuit parameters.

[0048] The switch Q2 is turned on only in the time period from the moment when the primary excitation inductor current of the switch power supply is demagnetized to zero and the primary inductor is released to the moment before the switch Q1 is turned on in the next working cycle, so as to realize zero current switching (ZCS) of the secondary rectifier of the switch power supply and zero voltage switching (ZVS) of the primary switch Q1. The parameter design will be described in detail in the third embodiment.

[0049] In the control method, the first voltage of the half-bridge midpoint and the second voltage between the resonant capacitor Cr are sampled for feedback control of the switch power supply, which is a primary feedback control method without auxiliary winding, can reduce the cost and design complexity of the switch power supply, and realizes zero voltage switching (ZVS) of the primary switch of the asymmetric half-bridge flyback resonant converter and zero current switching (ZCS) of the secondary switch, improves the frequency and efficiency, and reduces the volume.

[0050] Further, the switch Q2 is turned on only when the first voltage Vsw of the half-bridge midpoint SW is oscillated to the bottom, so as to avoid hard switching loss caused by too high voltage and loss caused by resonant oscillation.

[0051] Second embodiment

[0052] The control device is used for controlling the turn-on and turn-off of the switch of the switch power supply, and the switch power supply adopts an asymmetric half-bridge flyback resonant converter topology, which includes a switch Q1 in the upper bridge arm, a switch Q2 in the lower bridge arm, a resonant inductor Lr and a resonant capacitor Cr, and the connection point of the switch Q1 and the switch Q2 is a half-bridge midpoint SW, wherein the control device comprises:

[0053] The driving module is used for configuring the driving time sequence of the switch Q1 and the switch Q2, including configuring the switch Q1 and the switch Q2 to work in a non-complementary driving time sequence, and turning on the switch Q2 only in the time period from the first moment after the switch Q1 is turned off in the current working cycle to the moment before the switch Q1 is turned on in the next working cycle, the first moment being the moment when the primary excitation inductor current of the switch power supply is demagnetized to zero and the primary inductor is released, and the switch power supply starts to resonate from the first moment, and the first voltage Vsw of the half-bridge midpoint SW starts to rise to form an inflection point.

[0054] The feedback module is used for feedback control of the output voltage of the switch power supply, including taking the inflection point moment as a sampling moment, sampling the first voltage Vsw of the half-bridge midpoint SW and the second voltage Vcr between the resonant capacitor Cr at this moment, and performing closed-loop control on the switch power supply according to the first voltage Vsw and the second voltage Vcr to realize stable output of the switch power supply.

[0055] Further, the driving module starts to turn on the switch tube Q2 only when the first voltage Vsw of the half-bridge midpoint SW is at the bottom of the oscillation, which can avoid the hard turn-on loss caused by too high voltage and the loss caused by the resonance oscillation.

[0056] Further, the control device further comprises a voltage conversion circuit, which is configured to convert the voltage provided by the resonance capacitor Cr as a voltage source into a working voltage required by active devices in the switching power supply, so as to simplify the whole control device and facilitate integration.

[0057] The voltage conversion circuit is a circuit with voltage conversion function, such as LDO (low dropout regulator) or switching converter.

[0058] Further, the specific implementation of any of the above control devices is partially or entirely integrated into an IC.

[0059] Third embodiment

[0060] The embodiment provides a switching power supply, which adopts an asymmetric half-bridge flyback resonant converter topology, and comprises a switch tube Q1 located in an upper bridge arm, a switch tube Q2 located in a lower bridge arm, a resonant inductor Lr and a resonant capacitor Cr. The connection point of the switch tube Q1 and the switch tube Q2 is a half-bridge midpoint. The switching power supply further comprises the specific implementation of any of the control devices in the second embodiment.

[0061] Further, the capacitance Cr of the resonant capacitor Cr, the turn-on time ΔT of the switch tube Q2 in a cycle, the inductance Lr of the resonant inductor Lr and the ratio M comply with the following equation:

[0062]

[0063] The ratio M is the turn-on time ΔT of the switch tube Q2 and the resonance period of the resonant capacitor Cr and the resonant inductor from the first time after the switch tube Q1 is turned off to the time before the switch tube Q1 is turned on in the next working cycle.

[0064] Further, the ratio M is greater than 0.45+the first set value, and less than 0.45-the second set value.

[0065] It should be noted that the value of the ratio M is debugged and verified to be 0.45, and during the operation of the circuit, the turn-on time ΔT of the switch tube Q2 and the resonance period of the resonance inductance Lr and the resonance capacitance Cr will fluctuate within an acceptable accuracy range due to interference, therefore, the value of M cannot be fixed at 0.45 in practice, but will fluctuate around 0.45, so the value of M is greater than 0.45+the first set value; and less than 0.45-the second set value, and the first set value and the second set value are desirably as small as possible.

[0066] Further, the switch tube Q1 and / or the switch tube Q2 is a MOS tube or a GaN device.

[0067] It should be noted that when the switch tube Q1 and the switch tube Q2 are MOS tubes, diodes D1 and D2 are connected in parallel across the two ends of the MOS tubes, and the diodes D1 and D2 can be parasitic diodes of the MOS tubes or independent diodes connected in parallel externally, since the diodes D1 and D2 are parasitic diodes of the MOS tubes, oscillation will occur due to the influence of reverse recovery of the diodes, and effective control of the MOS tubes cannot be achieved, therefore, independent diodes D1 and D2 are generally connected in parallel across the two ends of the MOS tubes; when the switch tube Q1 and the switch tube Q2 are GaN devices, since GaN has no reverse recovery problem, independent diodes D1 and D2 can be connected in parallel.

[0068] Figure 3 The first circuit diagram of the switching power supply of the third embodiment of the present application is as follows, Figure 4 The equivalent circuit diagram of Figure 3 , wherein the resonance inductance Lr is the leakage inductance of the primary winding of the switching power supply transformer, and the resonance capacitance Cr is arranged in the circuit between the other end of the primary winding of the switching power supply transformer and the input ground of the switching power supply, Figure 3 The control device inIs not drawn, and can be understood through The waveform diagram of Figure 5 , The meanings of the waveforms in Are as follows: Figure 5

[0069] Vsw, is the first voltage of the half-bridge midpoint SW;

[0070] Vds1, is the voltage across the drain and source of the switch tube Q1;

[0071] Vcr, is the second voltage across the resonance capacitance Cr;

[0072] Im / Ir, is the current of the primary excitation inductance Lm of the switching power supply / the current of the leakage inductance lr of the transformer;

[0073] Drv1, is the drive signal of the switch tube Q1, high level turn-on, low level turn-off;

[0074] Drv2, the driving signal of the switch Q2, high level is on, low level is off;

[0075] In combination Figure 5 the waveform diagram, Figure 3 The working process of the switching power supply is as follows:

[0076] In the steady state working phase of the asymmetric half-bridge flyback converter, assuming that t0 is the starting time, then a complete working cycle can be divided into the following phases through t0-t9. In turn, they are:

[0077] t0-t1 phase, which corresponds to Figure 5 In the driving signal Drv1 in the high level, that is, the switch Q1 is turned on, at this time the energy of the input end of the switching power supply is given to the transformer primary excitation through the switch Q1, the transformer leakage inductance Lr, the primary excitation inductance Lm and the resonant capacitor Cr. This stage can be regarded as the resonant capacitor Cr resonates with the transformer leakage inductance Lr and the primary excitation inductance Lm. Since the resonant period is large, the excitation current can be approximately regarded as linearly increasing. From the t0 time, the excitation current Im linearly increases until the t1 time, and the switch Q1 is turned off.

[0078] t1-t2 phase, the switch Q1 and the switch Q2 are both in the off state. In this phase, the leakage inductance Lr and the excitation inductance Lm are both in the freewheeling state. The leakage inductance Lr, the excitation inductance Lm, the junction capacitances C1 and C2 between the drain and source of the switch Q1 and the switch Q2 and the resonant capacitor Cr resonate, so that the junction capacitance C1 of the switch Q1 is charged, the voltage between the drain and source of the switch Q1 rises, the junction capacitance C2 of the switch Q2 is discharged, and the first voltage Vsw of the half-bridge midpoint SW decreases. When Vcr-Vsw>N*Vo, the secondary diode D3 is turned on, and the energy is transmitted to the secondary side. Until the junction capacitance C2 of the switch Q2 is discharged, the first voltage Vsw of the half-bridge midpoint SW is clamped by the body diode D2 of the switch Q2, and the junction capacitance C1 of the switch Q1 is charged to the maximum.

[0079] t2-t3 phase, the secondary diode D3 is turned on, the energy stored in the primary side of the transformer is released to the secondary side, the excitation current Im linearly decreases, the primary side leakage inductance Lr and the resonant capacitor Cr resonate, and the primary side resonant current Ir flows through the body diode D2 of the switch Q2 to continue to flow. At the t3 time, the resonant current Ir decreases to 0. Since the body diode D2 of the switch Q2 exists, the current cannot be reversed, the resonant circuit is blocked, and the primary side resonant current Ir becomes zero. At this time, the second voltage Vcr across the resonant capacitor Cr is charged to the maximum value.

[0080] During the t3-t4 stage, the transformer continues to supply energy to the secondary side, the excitation current Im continues to decrease linearly, the first voltage Vsw at the midpoint SW of the half-bridge remains unchanged at 0, the secondary diode D3 continues to conduct, at time t4, the excitation current Im decreases to 0, and the current Id of the secondary diode D3 naturally decreases to zero; during this stage, the Vcr voltage remains unchanged.

[0081] During the t4-t5 stage, the transformer magnetizing inductance Lm is not clamped, so the leakage inductance Lr, the transformer primary magnetizing inductance Lm, and the drain-source junction capacitances C1 and C2 of switching transistors Q1 and Q2 resonate (since the resonant capacitance Cr is much larger than the junction capacitances C1 and C2, it can be considered that the resonant capacitance Cr does not participate in the resonance, and its voltage remains basically unchanged). At resonance, the sum of the voltages across junction capacitances C1 and C2 is equal to the input voltage Vin of the switching power supply. The Vsw voltage decreases, and the voltage drop across transistor Q1 increases. The initial value of the voltage across junction capacitance C1 is Vin, and the initial value of the voltage across junction capacitance C2 is... When the voltage is 0 and the magnetizing inductor current Im is 0, after resonance begins, junction capacitance C1 starts discharging and junction capacitance C2 starts charging. The first voltage Vsw at the midpoint SW of the half-bridge starts to rise, reaching an inflection point. This continues until junction capacitance C1 has completely discharged, at which point the first voltage Vsw at the midpoint SW reaches its maximum value. Then, junction capacitance C2 starts discharging and junction capacitance C1 starts charging. When the voltage of junction capacitance C1 reaches its maximum value, the first voltage Vsw at the midpoint SW of the half-bridge resonates to its minimum value, i.e., the valley. One resonant cycle is completed at time t5. Q2 can be controlled to turn on at this valley or at a subsequent resonant valley. Figure 5 The goal is to ensure that the circuit opens at the first trough. If it opens at a subsequent resonance trough, it will cycle through and form resonance according to the working process of this stage.

[0082] The t5-t6 stage, this stage Figure 5 At time t5, the first voltage oscillation at the midpoint SW of the half-bridge is detected to have reached its lowest point. At this time, the drive signal Drv2 is at a high level, the switch Q2 is turned on, the secondary diode D3 is turned on, the resonant capacitor Cr begins to discharge, and the magnetizing inductor Lm is reverse-excited, that is, the magnetizing current Im rises linearly in the negative direction. At the same time, the leakage inductance Lr and the resonant capacitor Cr resonate, releasing the energy stored in the resonant capacitor Cr to the secondary side. The primary resonant current Ir resonates along an approximately sinusoidal trajectory until the resonant current Ir and the magnetizing inductor current Im are equal. At this time, the current flowing through the secondary diode D3 naturally crosses zero, and this stage ends.

[0083] The t6-t7 stage, this stage Figure 5 The driving signal Drv2 remains high, the switching transistor Q2 remains on, and the current in the secondary diode D3 remains zero. At this time, the primary inductance of the transformer is released and not clamped, and the primary magnetizing inductance Lm and the resonant capacitor Cr resonate with a relatively large resonant period, approximating a straight line.

[0084] In the t7-t8 stage, the switch tube Q1 and the switch tube Q2 are in the off state, the leakage inductance Lr and the excitation inductance Lm are to be freewheeling, the leakage inductance Lr, the excitation inductance Lm and the junction capacitances C1 and C2 between the drain and source of the switch tube Q1 and the switch tube Q2 and the resonance capacitor Cr resonate, the junction capacitance C1 of the switch tube Q1 is discharged, the voltage of the switch tube Q1 decreases, the junction capacitance C2 of the switch tube Q2 is charged, the first voltage Vsw of the half-bridge midpoint SW rises, when Vcr-Vsw<N*Vo, the secondary diode D3 is turned off, until the first voltage Vsw of the half-bridge midpoint SW rises to the input voltage Vin of the switching power supply, at this time, the voltage across the primary winding of the transformer reverses, the voltage of the switch tube Q1 decreases to 0V.

[0085] In the t8-t9 stage, the leakage inductance Lr and the excitation inductance Lm and the resonance capacitor Cr resonate. Figure 5 In the t8-t9 stage, the leakage inductance Lr and the excitation inductance Lm and the resonance capacitor Cr resonate.

[0086] Thus, one cycle is completed, and then the same working process is repeated.

[0087] The first time after the switch tube Q1 is turned off to the time before the switch tube Q1 is turned on in the next working cycle is the t4-t5 stage in the above formula. Figure 5 In the t4-t5 stage, the first voltage Vsw of the half-bridge midpoint SW starts to rise, forming an inflection point (as shown in the Vsw voltage waveform), and the time when the inflection point appears is taken as the sampling time, the first voltage Vsw of the half-bridge midpoint SW and the second voltage Vcr across the resonance capacitor Cr are sampled, and the relationship of the output voltage can be obtained according to the Kirchhoff's voltage law (KVL) and the coupling relationship between the primary and secondary sides. Figure 5

[0088]

[0089] Wherein, N is the turns ratio of the transformer, and K is the proportion of the leakage inductance of the primary winding to the excitation inductance of the primary winding.

[0090] ​Thus, by the above expression, the first voltage Vsw of the half-bridge midpoint SW and the second voltage Vcr of the resonant capacitor Cr at the inflection point are sampled, and the information of the output voltage is obtained, so that the control of the secondary side output can be realized in the primary side.

[0091] Meanwhile, in order to improve the efficiency and reduce the loss, at the stage of t4-t5, when the resonance starts, the junction capacitor C1 starts to discharge, the junction capacitor C2 starts to charge, the first voltage of the half-bridge midpoint SW starts to rise, the voltage of the switch tube Q1 decreases, and the sum of the voltages of the junction capacitors C1 and C2 is the input voltage Vin of the switching power supply. Until the junction capacitor C1 is discharged completely, the first voltage Vsw of the half-bridge midpoint SW reaches the maximum value, then the junction capacitor C2 starts to discharge, and the junction capacitor C1 charges. When the voltage across the junction capacitor C1 reaches the maximum value, the first voltage Vsw of the half-bridge midpoint SW resonates to the minimum value, i.e., the bottom of the valley, and the resonance is formed in turn according to the above. At this time, the turn-on time of the control switch tube Q2 is the first resonance valley bottom, which can avoid the hard turn-on loss caused by too high voltage turn-on and the loss caused by the resonance oscillation.

[0092] Meanwhile, in order to realize the zero current turn-off (ZCS) of the secondary side and the zero voltage turn-on (ZVS) of the switch tube Q1, the turn-on time of the switch tube Q2 and the resonant parameters need to be designed. Specifically, the minimum negative current I N :

[0093]

[0094] Further, according to the set negative current I N , the turn-on time ΔT of the switch tube Q2 is obtained:

[0095]

[0096] The ratio of the turn-on time ΔT of the switch tube Q2 to the resonant period Tr (resonance of the resonant capacitor Cr and the leakage inductance Lr) at the stage of t5-t6 is set as M, so that:

[0097]

[0098] Then, the value of the resonant capacitor Cr is obtained:

[0099]

[0100] In this way, by setting the appropriate minimum negative current I N , the turn-on time ΔT of the switch tube Q2, and the capacitance of the resonant capacitor Cr, the zero current turn-off (ZCS) of the secondary side and the zero voltage turn-on (ZVS) of the switch tube Q1 can be realized.

[0101] Figure 6 The second equivalent circuit diagram of the switching power supply of the third embodiment of the present application, Figure 6 The difference between Figure 4 lies in the connection mode of the resonance capacitor Cr, Figure 6 The resonance capacitor Cr is under in the Figure 4 The resonance capacitor Cr is on in the, although the connection mode is different, the working mechanism of the switching power supply does not change, in the t4-t5 stage, the relationship of the output voltage can be obtained by the Kirchhoff voltage law (KVL) and the coupling relationship of the primary and secondary sides as follows:

[0102]

[0103] Wherein, N is the turns ratio of the transformer, K is the proportion of the leakage inductance of the primary side of the transformer to the inductance of the primary side.

[0104] Figure 6 The specific working principle details and the realization of the secondary side ZCS and the primary side ZVS will not be repeated here, and those skilled in the art can deduce them according to the Figure 4 analysis content.

[0105] Figure 7 The third equivalent circuit diagram of the switching power supply of the third embodiment of the present application, Figure 6 The difference between Figure 4 lies in that the power supply device is drawn, and the voltage conversion circuit is additionally added to the power supply device, and the other implementation circuits of the power supply device are integrated into the control IC outside the voltage conversion circuit.

[0106] Figure 7 The resonance capacitor Cr is taken as a voltage source, and the voltage across the resonance capacitor Cr is converted into the working voltage required by the control IC through the voltage conversion circuit. In combination with the waveforms shown in Figure 5 , it can be seen that the second voltage Vcr across the resonance capacitor Cr fluctuates around N*Vo under the control of the control device in the switching power supply of the present embodiment, the second voltage Vcr reaches the maximum at t3, and the second voltage Vcr reaches the minimum at t9. Therefore, the second voltage Vcr can be converted into the voltage required by the control IC by taking the resonance capacitor Cr as a voltage source through the voltage conversion circuit. The working current of the control IC is usually about 2mA, and the required energy is small, which has little effect on the working of the switching power supply of the present embodiment. The specific working process is basically the same as that of Figure 4 circuit, and those skilled in the art can deduce it by themselves.

[0107] Therefore, Figure 7 The switching power supply can realize that the resonance capacitor Cr additionally supplies power to the control IC by the above operation, so that the switching power supply of the present embodiment does not need an auxiliary winding to realize primary side control, which is conducive to reducing the size of the switching power supply.

[0108] Figure 8 The fourth equivalent circuit diagram of the switching power supply of the third embodiment of the present application is shown in Fig. 6, which is different from the third equivalent circuit diagram of the switching power supply of the second embodiment of the present application in that the power supply circuit is integrated into the control IC, so that the design and production of the switching power supply are more convenient. Figure 7

[0109] The above is only the preferred embodiment of the present application, it should be noted that the above preferred embodiment should not be considered as a limitation of the present application, for those skilled in the art, within the spirit and scope of the present application, several improvements and refinements can be made without departing from the spirit and scope of the present application, these improvements and refinements should also be considered as the protection scope of the present application, hereinafter, the protection scope of the present application should be limited by the scope of the claims.​

Claims

1. A control method for controlling the turn-on and turn-off of a switching tube in a switching power supply, the switching power supply adopting an asymmetric half-bridge flyback resonant converter topology, comprising a switching tube Q1 located in the upper bridge arm, a switching tube Q2 located in the lower bridge arm, a resonant inductor Lr and a resonant capacitor Cr, the connection point of the switching tube Q1 and the switching tube Q2 being a half-bridge midpoint, characterized in that, The control method comprises: The driving timing configuration method of the switch tube Q1 and the switch tube Q2 comprises configuring the switch tube Q1 and the switch tube Q2 to work in non-complementary driving timing, and starting to turn on the switch tube Q2 only in the time period from the first time point after the switch tube Q1 is turned off in the current working cycle to before the switch tube Q1 is turned on in the next working cycle, the first time point being the time point when the original side excitation inductance current of the switch power supply is demagnetized to zero, the original side inductance is released, and is not clamped by the output end of the switch power supply, the original side circuit of the switch power supply starts to resonate from the first time point, and the first voltage of the half-bridge midpoint starts to rise to form an inflection point; and the feedback control method of the output voltage of the switch power supply comprises taking the inflection point time as a sampling time point, sampling the first voltage of the half-bridge midpoint and the second voltage across the resonance capacitor Cr at the time point, obtaining the output voltage according to the first voltage and the second voltage based on Kirchhoff's voltage law and the primary-secondary side coupling relationship, and performing closed-loop control on the switch power supply by using the obtained output voltage to realize the stable output of the switch power supply.

2. The control method according to claim 1, characterized in that: The switch tube Q2 is started to be turned on only when the first voltage of the half-bridge midpoint oscillates to the bottom.

3. A control device for controlling the turn-on and turn-off of a switching tube in a switching power supply, the switching power supply adopting an asymmetric half-bridge flyback resonant converter topology, comprising a switching tube Q1 located in an upper bridge arm, a switching tube Q2 located in a lower bridge arm, a resonant inductor Lr and a resonant capacitor Cr, a connection point of the switching tube Q1 and the switching tube Q2 being a half-bridge midpoint, characterized in that, The control device comprises: The driving module is configured to configure the driving timing of the switch tube Q1 and the switch tube Q2, comprising configuring the switch tube Q1 and the switch tube Q2 to work in non-complementary driving timing, and starting to turn on the switch tube Q2 only in the time period from the first time point after the switch tube Q1 is turned off in the current working cycle to before the switch tube Q1 is turned on in the next working cycle, the first time point being the time point when the original side excitation inductance current of the switch power supply is demagnetized to zero, the original side inductance is released, and is not clamped by the output end of the switch power supply, the original side circuit of the switch power supply starts to resonate from the first time point, and the first voltage of the half-bridge midpoint starts to rise to form an inflection point; and the feedback module is configured to perform feedback control on the output voltage of the switch power supply, comprising taking the inflection point time as a sampling time point, sampling the first voltage of the half-bridge midpoint and the second voltage across the resonance capacitor Cr at the time point, obtaining the output voltage according to the first voltage and the second voltage based on Kirchhoff's voltage law and the primary-secondary side coupling relationship, and performing closed-loop control on the switch power supply by using the obtained output voltage to realize the stable output of the switch power supply.

4. The control device of claim 3, wherein: The switch tube Q2 is started to be turned on only when the first voltage of the half-bridge midpoint oscillates to the bottom.

5. The control device of claim 3, wherein The control device further comprises a voltage conversion circuit configured to convert the voltage provided by the resonance capacitor Cr as a voltage source into a working voltage required by active devices in the switch power supply.

6. The control device according to any one of claims 3 to 5, characterized by: The control device is partially or wholly integrated into an IC.

7. A switching power supply employing an asymmetric half-bridge flyback resonant converter topology, comprising a switch Q1 located in the upper bridge arm, a switch Q2 located in the lower bridge arm, a resonant inductor Lr and a resonant capacitor Cr, the connection point of the switch Q1 and the switch Q2 being the half-bridge midpoint, characterized in that, The switch power supply further comprises the control device according to any one of claims 3 to 6.

8. The switching power supply of claim 7, wherein: The resonant capacitance Cr, the time of the switch tube Q2 turned on in a cycle The resonant inductance Lr and the ratio M follow the following equation: Wherein, M is the on time of the switch tube Q2 The resonance period of the resonance capacitor Cr and the resonance inductor in the time period from the first time when the switch tube Q1 is off to the time before the switch tube Q1 is on in the next working cycle.

9. The switching power supply of claim 8, wherein: The ratio M is greater than 0.45+the first set value, and less than 0.45-the second set value.

10. The switching power supply of claim 7, wherein: The switch tube Q1 and / or the switch tube Q2 is a MOS tube or a GaN device. The switch tube Q1 and / or the switch tube Q2 is a MOS tube or a GaN device.

Images