Power conversion device

Abstract

An object of the present invention is to obtain a power conversion device that can prevent an increase in ripples caused by a change in the switching frequency. The power conversion device (100) includes: semiconductor switching elements (Q1, Q2) connected in series between a DC voltage source (101) and the output side; an inductor (105); a control unit (103) that controls the switching frequency of the semiconductor switching elements (Q1, Q2); and a voltage detector (109), a voltage detector (111), and a current detector (110) that respectively detect the voltage value of the input voltage, the voltage value of the output voltage, and the current value of the inductor current. At a timing synchronized with the carrier wave, the voltage value of the input voltage, the voltage value of the output voltage, the current value of the inductor current, and the change in the switching frequency are detected.

Description

Technical Field

[0001] The present application relates to a power conversion device. Background Art

[0002] In the case of interconnecting power generated by solar power generation to an AC system through an inverter, a power conversion device including a chopper circuit that performs DC-DC conversion is used between the inverter and the solar power generator. As a method for improving the efficiency of the chopper circuit, a method of controlling the switching frequency according to the operating state to reduce the loss of the chopper circuit can be cited. Here, the loss of the chopper circuit includes the loss of the reactor and the loss of the semiconductor switching element, etc. Conventionally, there has been a technique of obtaining the loss of the chopper circuit as a function of the switching frequency and selecting the switching frequency with the minimum loss (for example, refer to Patent Document 1).

[0003] Prior Art Documents

[0004] Patent Documents

[0005] Patent Document 1: Japanese Patent Laid-Open No. 2011-101554 Summary of the Invention

[0006] Technical Problem to be Solved by the Invention

[0007] However, in the case of using the technique described in Patent Document 1, if the timing of changing the switching frequency is not appropriately set, the ripple may increase as the switching frequency changes.

[0008] The present application discloses a technique for solving the above problems, and an object thereof is to obtain a power conversion device that can prevent an increase in ripple caused by a change in the switching frequency.

[0009] Technical Means for Solving the Technical Problem

[0010] The power conversion device disclosed in the present application converts an input voltage from a DC voltage source and outputs it as an output voltage. The power conversion device includes: a plurality of semiconductor switching elements connected in series between the DC voltage source and the output side; an inductor connected between the DC voltage source and the output side; a control unit that controls the switching frequency of the plurality of semiconductor switching elements; and a detection unit that detects the voltage value of the input voltage, the voltage value of the output voltage, and the current value of the inductor current flowing through the inductor. The control unit includes: a voltage control unit that calculates an output voltage command value based on the voltage value of the input voltage and the voltage value of the output voltage obtained through detection; a frequency control unit that calculates a frequency command value based on the voltage value of the input voltage, the output voltage command value, and the switching frequency; and a carrier generation unit that generates a carrier based on the frequency command value, and at a timing synchronized with the carrier, performs detection based on the detection unit and change of the switching frequency based on the control unit.

[0011] Advantages of the Invention

[0012] According to the power conversion device disclosed in the present application, an increase in ripple caused by a change in the switching frequency can be prevented. BRIEF DESCRIPTION OF THE DRAWINGS

[0013] Figure 1 It is a schematic structural diagram showing the power conversion device in Embodiment 1.

[0014] Figure 2 It is a block diagram showing the frequency control unit related to Embodiment 1.

[0015] Figure 3 It is a block diagram showing the loss calculation unit related to Embodiment 1.

[0016] Figure 4A It is a circuit diagram of the boost chopper related to Embodiment 1.

[0017] Figure 4B It is a diagram showing the generation mode of the switching loss of the power conversion device in Embodiment 1.

[0018] Figure 5 It is a diagram showing the current path before the semiconductor switching element Q1 is turned off in the power conversion device in Embodiment 1.

[0019] Figure 6 It is a diagram showing the current path before the semiconductor switching element Q2 is turned on in the power conversion device in Embodiment 1.

[0020] Figure 7It is a diagram showing the current path before the semiconductor switch element Q2 is turned off in the power conversion device in Embodiment 1.

[0021] Figure 8 It is a diagram showing the current path before the semiconductor switch element Q1 is turned on in the power conversion device in Embodiment 1.

[0022] Figure 9 It is a diagram showing the timing for updating the switching frequency in the power conversion device in Embodiment 1.

[0023] Figure 10 It is a diagram showing the switching frequency control unit related to Embodiment 2 and is a diagram showing the outline of frequency control using the hill-climbing method.

[0024] Figure 11 It is a diagram for explaining the frequency control using the hill-climbing method.

[0025] Figure 12 It is a flowchart showing the operation of the frequency determination unit related to Embodiment 2.

[0026] Figure 13 It is a circuit diagram of the power conversion device in Embodiment 3 and is a circuit diagram when the power conversion unit is a buck chopper.

[0027] Figure 14 It is a circuit diagram of the power conversion device in Embodiment 4 and is a circuit diagram when the power conversion unit is a buck-boost chopper.

[0028] Figure 15A It is a circuit diagram of the buck-boost chopper related to Embodiment 4.

[0029] Figure 15B It is a diagram showing the generation mode of switching losses in the power conversion device in Embodiment 4.

[0030] Figure 16 It is a diagram showing the current path before the semiconductor switch element Q1 is turned off in the power conversion device in Embodiment 4.

[0031] Figure 17 It is a diagram showing the current path before the semiconductor switch element Q2 is turned on in the power conversion device in Embodiment 4.

[0032] Figure 18 It is a diagram showing the current path before the semiconductor switch element Q2 is turned off in the power conversion device in Embodiment 4.

[0033] Figure 19 It is a diagram showing the current path before the semiconductor switch element Q1 is turned on in the power conversion device in Embodiment 4.

[0034] Figure 20A This is the circuit diagram of the power conversion device in Embodiment 5, which is the circuit diagram when an IGBT is used as the switching element of the boost chopper.

[0035] Figure 20B This is a diagram showing the generation mode of the switching loss of the power conversion device in Embodiment 5.

[0036] Figure 21A This is the circuit diagram of the power conversion device in Embodiment 6, which is the circuit diagram when an IGBT is used as the switching element of the buck - boost chopper.

[0037] Figure 21B This is a diagram showing the generation mode of the switching loss of the power conversion device in Embodiment 6.

[0038] Figure 22 This is a diagram showing an example of the hardware structure of the control unit related to each embodiment. Detailed Embodiments

[0039] Embodiment 1.

[0040] Based on Figures 1 to 9 、 Figure 22 Embodiment 1 will be described. Figure 1 This is a schematic structural diagram of the power conversion device in Embodiment 1. The power conversion device 100 includes: a power conversion unit 102 connected to a DC voltage source 101 and converting the input voltage input from the DC voltage source 101 into an output voltage of a desired magnitude; and a control unit 103 for controlling the power conversion unit 102.

[0041] The power conversion unit 102 has a boost chopper circuit, which includes semiconductor switching elements Q1 and Q2 respectively composed of MOSFETs (Metal Oxide Semiconductor Field Effect Transistors) and connected in series with each other. Specifically, a smoothing capacitor 104 is connected in parallel with the DC voltage source 101. The connection point between the positive electrode side of the DC voltage source 101 and the positive electrode side of the smoothing capacitor 104 is connected to the connection point between the source terminal of the semiconductor switching element Q1 and the drain terminal of the semiconductor switching element Q2 via a reactor 105, that is, an inductor. The drain terminal of the semiconductor switching element Q1 is connected to the positive electrode side of the smoothing capacitor 108 on the output side. The source terminal of the semiconductor switching element Q2 is connected to the negative electrode side of the DC voltage source 101 and the negative electrode side of the smoothing capacitor 108. In addition, free - wheeling diodes D1 and D2 are respectively anti - parallel connected to the semiconductor switching elements Q1 and Q2.

[0042] The voltage across both ends of the smoothing capacitor 104 is detected by the voltage detector 109 as the input voltage. The voltage detector 109 outputs the detected voltage value to the control unit 103 as the input voltage detection value Vin_det. The voltage across both ends of the smoothing capacitor 108 is detected by the voltage detector 111 as the output voltage. The voltage detector 111 outputs the detected voltage value to the control unit 103 as the output voltage detection value Vout_det. A current detector 110 for detecting the inductor current is provided between the connection point on the positive electrode side of the DC voltage source 101 and the positive electrode side of the smoothing capacitor 104 and the reactor 105. The current detector 110 outputs the detected current value to the control unit 103 as the inductor current detection value IL_det.

[0043] The control unit 103 includes: a voltage control unit 112, which inputs the input voltage detection value Vin_det, the output voltage detection value Vout_det, the inductor current detection value IL_det, and a frequency command value fsw_ref described later, and uses these values to calculate a duty ratio command value Dref, that is, an output voltage command value; a frequency control unit 113, which inputs the input voltage detection value Vin_det, the output voltage detection value Vout_det, the inductor current detection value IL_det, and the duty ratio command value Dref, and uses these values to calculate the frequency command value fsw_ref; a carrier generation unit 114, which inputs the frequency command value fsw_ref and uses the frequency command value fsw_ref to generate a carrier car; and a gate signal generation unit 115, which inputs the duty ratio command value Dref and the carrier car, and uses the duty ratio command value Dref and the carrier car to generate gate signals GQ1 and GQ2 through PWM control. The gate signals GQ1 and GQ2 are gate signals for driving the semiconductor switch elements Q1 and Q2, respectively.

[0044] The voltage control unit 112 changes the control parameter according to the frequency command value fsw_ref. As an example of this control parameter, there is an integral time Ti of proportional-integral control. The integral time Ti is determined by the following formula (1).

[0045] [Mathematical formula 1]

[0046]

[0047] In formula (1), Ti0 is the designed value of the integral time. As shown in formula (1), the integral time Ti is changed according to the frequency command value fsw_ref, so that even when the switching frequency is changed, the operation can be performed with the designed integral time.

[0048] Figure 2 This is a block diagram showing the frequency control unit according to Embodiment 1. The frequency control unit 113 includes: a loss calculation unit 113a that calculates the loss generated by the operation power conversion unit 102; and a frequency determination unit 113b that determines a frequency command value fsw_ref to optimize the loss. The loss calculation unit 113a receives an input voltage detection value Vin_det, an output voltage detection value Vout_det, an inductor current detection value IL_det, and a duty ratio command value Dref, and uses these values to calculate a loss calculation value Ploss. The frequency determination unit 113b uses a loss map set in advance based on the input and output voltages and the inductor current to determine the frequency command value fsw_ref so that the loss generated by the power conversion unit 102 is minimized.

[0049] Figure 3 This is a block diagram showing the loss calculation unit according to Embodiment 1. The loss calculation unit 113a includes: a current change amount calculation unit 113a1 that receives an input voltage detection value Vin_det, an output voltage detection value Vout_det, and a duty ratio command value Dref, and uses these values to calculate a current change amount ΔI; a magnetic flux density calculation unit 113a2 that uses the current change amount ΔI to calculate a magnetic flux density change amount ΔB; an iron loss calculation unit 113a3 that uses the magnetic flux density change amount ΔB and the duty ratio command value Dref to calculate an iron loss Pir; a copper loss calculation unit 113a4 that uses the inductor current detection value IL_det to calculate a copper loss Pcop; a switching loss calculation unit 113a5 that uses the inductor current detection value IL_det and the current change amount ΔI to calculate a switching loss Psw; and a conduction loss calculation unit 113a6 that uses the inductor current detection value IL_det and the current change amount ΔI to calculate a conduction loss Pcon. In addition, the loss calculation unit 113a includes: four multipliers 113a7 that multiply the respectively calculated iron loss Pir, copper loss Pcop, switching loss Psw, and conduction loss Pcon by constants Kloss1, Kloss2, Kloss3, and Kloss4; and an adder 113a8 that adds the outputs of the respective multipliers 113a7 to calculate the loss calculation value Ploss.

[0050] The current change amount calculation unit 113a1 calculates the current change amount ΔI according to the following formula (2).

[0051] [Mathematical formula 2]

[0052]

[0053] In Equation (2), L is the inductance of the reactor 105, and fsw is the switching frequency of the semiconductor switching elements Q1 and Q2.

[0054] The magnetic flux density operation unit 113a2 calculates the magnetic flux density change amount ΔB according to the following Equation (3).

[0055] [Mathematical formula 3]

[0056]

[0057] In Equation (3), N is the number of turns of the reactor 105, and S is the cross-sectional area of the iron core of the reactor 105.

[0058] The iron loss operation unit 113a3 calculates the iron loss Pir according to the IGSE (improved Generalized Steinmetz Equation) shown in the following Equation (4).

[0059] [Mathematical formula 4]

[0060]

[0061] In Equation (4), VOLcore is the iron core volume of the reactor 105, and α, β*, and k*i are constants determined by the magnetic flux density change amount ΔB, the DC bias magnetic field, and the characteristics of the magnetic material of the reactor 105. For example, when ferrite is used as the magnetic material, the value of α is 0 to 2, the value of β* is 0 to 4, and the value of k*i is 0 to 12.

[0062] The copper loss operation unit 113a4 calculates the copper loss Pcop according to the following Equation (5).

[0063] [Mathematical formula 5]

[0064]

[0065] In Equation (5), Rw is the winding resistance of the reactor 105, and IL_det is the detected value of the inductor current.

[0066] The switching loss operation unit 113a5 calculates the switching loss Psw of the semiconductor switching elements Q1 and Q2 according to the following Equation (6).

[0067] [Mathematical formula 6]

[0068]

[0069] In Equation (6), Eon_Q1 and Eoff_Q1 are the conduction loss and turn-off loss of semiconductor switch element Q1, and Eon_Q2 and Eoff_Q2 are the conduction loss and turn-off loss of semiconductor switch element Q2. In addition, Erec is the recovery loss of freewheeling diodes D1 and D2.

[0070] Eon_Q1 is a function of the drain current ID_on_Q1 and the drain-source voltage VDS_on_Q1 when semiconductor switch element Q1 is conducting. Therefore, it is obtained by using a data table or the like that represents the relationship between the conduction loss, the drain current, and the drain-source voltage, based on the drain current and the drain-source voltage when calculating Eon_Q1. Eoff_Q1 is a function of the drain current ID_off_Q1 and the drain-source voltage VDS_off_Q1 when semiconductor switch element Q1 is turning off. Therefore, it is obtained by using a data table or the like that represents the relationship between the turn-off loss, the drain current, and the drain-source voltage, based on the drain current and the drain-source voltage when calculating Eoff_Q1. Such a data table can be a pre-made data table. The same applies to Eon_Q2 and Eoff_Q2. Erec is a function of the anode current IA_rec and the cathode-anode voltage VKA_rec during recovery. Therefore, it is obtained by using a data table or the like that represents the relationship between the recovery loss, the anode current, and the cathode-anode voltage, based on the anode current and the cathode-anode voltage when calculating Erec.

[0071] In addition, the current and voltage at the time of occurrence of each loss are described later.

[0072] In addition, in the calculation of Eon_Q1 and the like included in the switching loss Psw, the method of using a data table as described above is not limited, and an approximate formula composed of the above current and voltage can also be used. As an example, as shown in Equation (7) below, Eon_Q1 is approximately expressed as a power series of ID. In Equation (7), ID is the current value at the timing when the conduction loss occurs in semiconductor switch element Q1, and is a value calculated based on the inductor current detection value IL_det. an (n is a positive integer) is a coefficient obtained by generating an approximate formula from a data table.

[0073] [Equation 7]

[0074]

[0075] The same applies to the other losses included in Equation (6). Eoff_Q1, Eon_Q2, Eoff_Q2, and Erec can be approximately obtained by a power series composed of the current values at the timings when the turn-off losses occur in the semiconductor switching element Q1, the current values at the timings when the conduction losses occur in the semiconductor switching element Q2, the current values at the timings when the turn-off losses occur in the semiconductor switching element Q2, and the current values at the timings when the recovery losses occur in the freewheeling diodes D1 and D2.

[0076] The conduction loss calculation unit 113a6 calculates the conduction losses Pcon of the semiconductor switching elements Q1 and Q2 according to the following Equation (8).

[0077] [Mathematical formula 8]

[0078] P con =(V on_DS +V on_SD )I L_det …(8) In Equation (8), Von_DS is the conduction voltage between the drain and source of the semiconductor switching elements Q1 and Q2, Von_SD is the conduction voltage between the source and drain of the semiconductor switching elements Q1 and Q2, and Von_DS and Von_SD are values depending on the drain current and the source current, respectively.

[0079] The multiplier 113a7 multiplies the iron loss Pir, copper loss Pcop, switching loss Psw, and conduction loss Pcon calculated as above by constants Kloss1, Kloss2, Kloss3, and Kloss4, respectively. Here, the constants Kloss1, Kloss2, Kloss3, and Kloss4 are 0 or 1, respectively. The adder 113a8 sums up the respective losses multiplied by the constants Kloss1, Kloss2, Kloss3, and Kloss4 to calculate the loss calculation value Ploss.

[0080] Here, the hardware structure of each functional unit that implements the control unit 103 will be described. Figure 22This is a diagram showing an example of the hardware configuration of the control unit of the power conversion device in each embodiment. The control unit 103 mainly consists of a processor 81, a memory 82 serving as a main storage device, and an auxiliary storage device 83. The processor 81 is composed of, for example, a CPU (Central Processing Unit), an ASIC (Application Specific Integrated Circuit), a DSP (Digital Signal Processor), an FPGA (Field Programmable Gate Array), etc. The memory 82 is composed of a volatile storage device such as a random access memory, and the auxiliary storage device 83 is composed of a non-volatile storage device such as a flash memory or a hard disk. The auxiliary storage device 83 stores a prescribed program executed by the processor 81. The processor 81 appropriately reads this program for execution and performs various arithmetic processes. At this time, the above-mentioned prescribed program is temporarily saved from the auxiliary storage device 83 to the memory 82, and the processor 81 reads the program from the memory 82. Figure 2 , Figure 3 The arithmetic processes of the respective functional units shown above are realized by the processor 81 executing a prescribed program. The result of the arithmetic process obtained by the processor 81 is temporarily stored in the memory 82 and stored in the auxiliary storage device 83 according to the purpose of the executed arithmetic process.

[0081] In addition, the control unit 103 includes an input circuit 84 that accepts the input of the inductor current detection value IL_det, the input voltage detection value Vin_det, and the output voltage detection value Vout_det output from the power conversion unit 102; and an output circuit 85 that outputs the gate signals GQ1 and GQ2 to the power conversion unit 102.

[0082] Next, the switching loss generation mode will be described. Figure 4A This is a circuit diagram of the boost chopper according to Embodiment 1. Figure 4B This is a diagram showing the switching loss generation mode of the power conversion device in Embodiment 1. Figure 4A In this figure, I_Q1 is the current flowing through the semiconductor switching element Q1, and I_Q2 is the current flowing through the semiconductor switching element Q2. In addition, I_D1 is the current flowing through the freewheeling diode D1, and I_D2 is the current flowing through the freewheeling diode D2. Further, in the following description, the inductor current IL flowing through the reactor 105 is always set to be positive. In addition, the positive directions of the respective currents and voltages are Figure 4A as shown.

[0083] Figure 4BIn this case, the state at the moment when the inductor current IL is detected is set as the initial state. In the initial state, the gate signal GQ1 is on (the semiconductor switching element Q1 is on), and the gate signal GQ2 is off (the semiconductor switching element Q2 is off). When the control period is set as T, according to the duty ratio command value Dref, at the moment t1 (= T×(1 - Dref) / 2), the gate signal GQ1 is off (the semiconductor switching element Q1 is off). At the moment t2 (= t1 + td), the gate signal GQ2 is on (the semiconductor switching element Q2 is on). Additionally, td is the dead time. At the moment t3 (= t1 + T×Dref), the gate signal GQ2 is off (the semiconductor switching element Q2 is off). At the moment t4 (= t3 + td), the gate signal GQ1 is on again (the semiconductor switching element Q1 is on).

[0084] As Figure 4B shown, at the moment t1, the turn-off loss (Eoff_Q1) of the semiconductor switching element Q1 is generated, and at the moment t2, the turn-on loss (Eon_Q2) of the semiconductor switching element Q2 is generated. In addition, at the moment t3, the turn-off loss (Eoff_Q2) of the semiconductor switching element Q2 is generated, and at the moment t4, the turn-on loss (Eon_Q1) of the semiconductor switching element Q1 is generated. Also, at the moment t2 when the semiconductor switching element Q2 is on, the recovery loss (Erec) is generated in the freewheeling diode D1. Additionally, the inductor current IL is equal to the inductor current detection value IL_det in the initial state, but then varies with the period T. The variation amplitude during the vibration period is ΔIL.

[0085] Figure 5 is a diagram showing the current path before the semiconductor switching element Q1 is turned off in the power conversion device in Embodiment 1. Figure 5 In this case, Vin represents the input voltage and Vout represents the output voltage. As Figure 5 shown, the MOSFET of the semiconductor switching element Q1 is on, so the current flowing through the boost chopper circuit becomes equal to the inductor current IL.

[0086] According to Figure 4B and Figure 5 , when calculating the turn-off loss generated in the semiconductor switching element Q1 in the boost chopper, the following equations (9) and (10) are used to calculate the drain current ID_off_Q1 and the drain-source voltage VDS_off_Q1 when the semiconductor switching element Q1 is turned off.

[0087] [Equation 9]

[0088]

[0089] [Equation 10]

[0090] V DS_off_Q1 = V fwd_D1 …(10)

[0091] In Equation (10), Vfwd_D1 is the forward voltage of the freewheeling diode D1 and has a value that depends on the current (I_D1) flowing through the freewheeling diode D1.

[0092] Figure 6 FIG. is a diagram showing the current path before the semiconductor switching element Q2 is turned on in the power conversion device according to Embodiment 1, and shows the current path during the dead time after the semiconductor switching element Q1 is turned off. However, the dead time is sufficiently short with respect to the control period, and the change in the current during the dead time can be ignored. During the dead time after the semiconductor switching element Q1 is turned off, the current flowing through the freewheeling diode D1 is equal to the inductor current IL.

[0093] According to Figure 4B and Figure 6 , when calculating the conduction loss generated in the semiconductor switching element Q2 in the boost chopper, the following Equations (11) and (12) are used to calculate the drain current ID_on_Q2 and the drain-source voltage VDS_on_Q2 when the semiconductor switching element Q2 is turned on.

[0094] [Mathematical Equation 11]

[0095]

[0096] [Mathematical Equation 12]

[0097] V DS_on_Q2 = V out_det …(12)

[0098] In addition, when calculating the recovery loss generated in the freewheeling diode D1, the following Equations (13) and (14) are used to calculate the anode current IA_rec and the cathode-anode voltage VKA_rec when the freewheeling diode D1 recovers.

[0099] [Mathematical Equation 13]

[0100]

[0101] [Mathematical Equation 14]

[0102] V KA_rec = V out_det …(14)

[0103] Figure 7 FIG. is a diagram showing the current path before the semiconductor switching element Q2 is turned off in the power conversion device according to Embodiment 1. As Figure 7As shown, the MOSFET of the semiconductor switching element Q2 is turned on. Therefore, the current flowing through the boost chopper circuit becomes equal to the inductor current IL.

[0104] According to Figure 4B and Figure 7 , when calculating the turn-off loss generated in the semiconductor switching element Q2 in the boost chopper, the following equations (15) and (16) are used to calculate the drain current ID_off_Q2 and the drain-source voltage VDS_off_Q2 when the semiconductor switching element Q2 is turned off.

[0105] [Equation 15]

[0106]

[0107] [Equation 16]

[0108] V DS_off_Q2 =V out_det …(16)

[0109] Figure 8 FIG. is a diagram showing the current path before the semiconductor switching element Q1 is turned on in the power conversion device according to Embodiment 1, and shows the current path during the dead time after the semiconductor switching element Q2 is turned off. During the dead time after the semiconductor switching element Q1 is turned off, the current flowing through the freewheeling diode D1 is equal to the inductor current IL.

[0110] According to Figure 4B and Figure 8 , when calculating the conduction loss generated in the semiconductor switching element Q1 in the boost chopper, the following equations (17) and (18) are used to calculate the drain current ID_on_Q1 and the drain-source voltage VDS_on_Q1 when the semiconductor switching element Q1 is turned on.

[0111] [Equation 17]

[0112]

[0113] [Equation 18]

[0114] V DS_on_Q1 =V fwd_D1 …(18)

[0115] Figure 9 FIG. is a diagram showing the timing for updating the switching frequency in the power conversion device according to Embodiment 1. Figure 9In it, the time courses of the frequency command value fws_ref, the integration time Ti, the carrier car, the gate signals GQ1, GQ2, the output voltage Vout, the input voltage Vin, and the inductor current IL are shown. In addition, T1 and T2 respectively show the switching periods when the frequency command values are fsw_refl and fsw_ref2. Figure 9 As shown, A is the timing when the phase of the triangular-wave carrier car is 0 deg (the trough of the carrier), and the timing B is the timing when the phase of the triangular-wave carrier car is 180 deg (the peak of the carrier).

[0116] In Embodiment 1, the output voltage Vout, the input voltage Vin, and the inductor current IL are detected at the timing A. That is, the output voltage detection value Vout_det, the input voltage detection value Vin_det, and the inductor current detection value IL_det are obtained. As Figure 9 shown, the inductor current IL varies due to the influence of the switch, but by detecting the inductor current IL at the timing A when the triangular-wave carrier is at the trough, the influence of the switch on the detection value can be reduced, and the average value of the inductor current IL can be detected. In addition, the detection of the output voltage Vout, the input voltage Vin, and the inductor current IL can be performed at the timing B when the triangular-wave carrier is at the peak. In this case, similarly to the case of timing A, the average value of the inductor current IL can also be detected.

[0117] In addition, when the frequency command value fsw_ref is changed from fsw_ref1 to fsw_ref2, the frequency command value fsw_ref and the integration time Ti are changed at the timing A when the triangular-wave carrier is at the trough. Thereby, an increase in ripple due to the change in the switching frequency can be prevented. In addition, the same effect can be obtained by changing the frequency command value fsw_ref and the integration time Ti at the timing B when the triangular-wave carrier is at the peak.

[0118] In addition, in Embodiment 1, the case where the carrier is a triangular-wave carrier has been described, but the case where the carrier is a sawtooth-wave carrier can also be considered. When the carrier is a triangular-wave carrier, the input voltage, etc. are detected at the peak or trough of the carrier, so that the influence of the ripple caused by the switch can be reduced, and the average value of the inductor current can be detected. On the other hand, when the carrier is a sawtooth-wave carrier, the timing of the switch is the same as the timing of the peak and trough of the carrier. If the timing of the detection of the input voltage, etc. and the change of the switching frequency is matched with the peak or trough of the carrier, the influence of the ripple is generated. Therefore, the detection of the input voltage, etc. and the change of the switching frequency are staggered from the peak and trough of the carrier and are performed synchronously with the carrier. Thereby, the same effect as the above case of the triangular-wave carrier can be obtained.

[0119] According to Embodiment 1, an increase in ripple caused by a change in the switching frequency can be prevented. More specifically, at the timing synchronized with the carrier, detection of the input voltage, output voltage, and inductor current accompanying the change in the switching frequency, and the change in the switching frequency are performed. Thereby, the influence of the switching on the detected values can be reduced. In addition, by changing the switching frequency in a state where the influence of the switching on the detected values is reduced, an increase in ripple caused by the change in the switching frequency can be prevented.

[0120] Embodiment 2.

[0121] Next, based on Figures 10 to 12 Embodiment 2 will be described. In Embodiment 1, the loss map is used to determine the specific value of the frequency command value fsw_ref, but in Embodiment 2, the hill-climbing method is used to determine the frequency command value fsw_ref. Figure 10 is a diagram showing the frequency control unit according to Embodiment 2, and is a diagram showing an outline of the frequency control using the hill-climbing method. The frequency control unit 213 includes a loss calculation unit 113a and a frequency determination unit 213b. The loss calculation unit 113a is the same as that in Embodiment 1. Utilizing the case where the loss calculation value Ploss is a downwardly convex function with respect to the switching frequency, the frequency determination unit 213b determines the frequency command value fsw_ref so that the loss calculation value Ploss is minimized.

[0122] Figure 11 is a diagram for explaining the frequency control using the hill-climbing method, and is a diagram showing the operation principle of the frequency determination unit 213b. In the case where the loss calculation value Ploss has a downwardly convex relationship with respect to the switching frequency fsw, the hill-climbing method can be used to search for the frequency command value fsw_ref at which the loss calculation value Ploss is minimized. In a situation where the maximum value and minimum value of the frequency command value fsw_ref are set to fsw_max and fsw_min according to the maximum loss Ploss_max at the time of thermal design, the frequency determination unit 213b obtains the loss minimum frequency fsw_refm at which the loss calculation value Ploss is minimized. From Equation (2) and Equation (5), it can be seen that if the switching frequency fsw increases, the current change amount ΔI decreases, and the copper loss Pcop also decreases. In addition, if the current change amount ΔI decreases, the conduction loss Pcon also decreases. On the other hand, from Equation (6), it can be seen that if the switching frequency fsw increases, the switching loss Psw increases. The characteristics of the iron loss Pir with respect to the switching frequency fsw vary depending on the magnetic material, but in the case of a ferrite core used for an inductor (reactor) in a power conversion device, it generally increases with respect to the switching frequency fsw.

[0123] In the case where the loss minimum frequency fsw_refm is obtained by using the hill climbing method, first, the loss calculation unit 113a calculates a loss calculation value Ploss including at least one of the iron loss Pir, the copper loss Pcop, the switching loss Psw, and the conduction loss Pcon, and inputs the loss calculation value Ploss to the frequency determination unit 213b. When currently operating at a switching frequency fsw1 lower than the loss minimum frequency fsw_refm and the loss calculation value is Ploss1, the frequency determination unit 213b increases the switching frequency by Δfsw and sets the switching frequency to fsw2. As Figure 11 shown, if the loss calculation value Ploss2 when the switching frequency is fsw2 is smaller than the loss calculation value Ploss1, the frequency determination unit 213b further increases the switching frequency by Δfsw. When the loss calculation value Ploss after the increase in the switching frequency is larger than that before the increase in the switching frequency, the frequency determination unit 213b determines that the switching frequency fsw at which the loss is minimum has been passed, decreases the switching frequency by Δfsw and returns, and determines the returned value as the loss minimum frequency fsw_refm. The frequency determination unit 213b outputs the loss minimum frequency fsw_refm as the frequency command value fsw_ref.

[0124] Regarding the operation of the frequency determination unit 213b, it is described in detail using the Figure 12 flowchart shown. First, the value 1 is set for the variable N (step ST001). The variable N is used to grasp the magnitude relationship between the switching frequency fsw and the loss minimum frequency fsw_refm during the search for the loss minimum point based on the hill climbing method. In addition, the loss calculation value Ploss at the start of the search has already been calculated.

[0125] Next, the switching frequency fsw is increased by Δfsw (step ST002), the loss calculation value Ploss is recalculated using the increased switching frequency fsw, and the loss calculation value Ploss from the previous time (before the increase in the switching frequency) is compared with the loss calculation value Ploss from this time (after the increase in the switching frequency) (step ST003). If the loss calculation value Ploss from the previous time is larger than the loss calculation value Ploss from this time, proceed to step ST004; otherwise (the loss calculation value Ploss from the previous time is less than or equal to the loss calculation value Ploss from this time), proceed to step ST005.

[0126] In the case where the previous loss calculation value Ploss is greater than the current loss calculation value Ploss in step ST003, the value of the current variable N is set to 0 (step ST004). In this case, the frequency determination unit 213b determines that the previous switching frequency fsw is smaller than the minimum loss frequency fsw_refm, and returns to step ST002. Thus, in the case where the condition "the previous loss calculation value Ploss is greater than the current loss calculation value Ploss" continues, the switching frequency fsw is continuously increased.

[0127] In the case where the previous loss calculation value Ploss is less than or equal to the current loss calculation value Ploss in step ST003, it is determined whether the value of the variable N is 0 or 1 (step ST005). If the value of the variable N is 1, proceed to step ST006. If the value of the variable N is not 1 (is 0), proceed to step ST007. When the variable N is 0, the frequency determination unit 213b determines that as a result of increasing the switching frequency, the switching frequency fsw that minimizes the loss is passed, and the switching frequency fsw is decreased by Δfsw (step ST007). The frequency determination unit 213b determines the decreased switching frequency fsw as the minimum loss frequency fsw_refm and ends the search for the minimum loss frequency fsw_refm.

[0128] When the value of the variable N is 1 in step ST005, the switching frequency fsw is decreased by Δfsw (step ST006).

[0129] After step ST006, the loss calculation value Ploss is recalculated using the decreased switching frequency fsw, and the previous (before the switching frequency decrease) loss calculation value Ploss is compared with the current (after the switching frequency decrease) loss calculation value Ploss (step ST008). In the case where the previous loss calculation value Ploss is greater than the current loss calculation value Ploss, the frequency determination unit 213b determines that the previous switching frequency fsw is greater than the minimum loss frequency fsw_refm, and returns to step ST006. Thus, in the case where the condition "the previous loss calculation value Ploss is greater than the current loss calculation value Ploss" continues, the switching frequency fsw is continuously decreased.

[0130] In the case where the previous loss calculation value Ploss is less than or equal to the current loss calculation value Ploss, proceed to step ST009. In this case, the frequency determination unit 213b determines that as a result of decreasing the switching frequency, the switching frequency fsw that minimizes the loss is passed, and the switching frequency fsw is increased by Δfsw (step ST009). The frequency determination unit 213b determines the increased switching frequency fsw as the minimum loss frequency fsw_refm and ends the search for the minimum loss frequency fsw_refm.

[0131] According to Embodiment 2, the same results as those of Embodiment 1 can be obtained. In addition, the hill climbing method is used to determine the switching frequency command value with the minimum loss. Therefore, the power converter can operate at the switching frequency with the minimum loss without presetting the loss map.

[0132] Embodiment 3.

[0133] Next, based on Figure 13 Embodiment 3 will be described. In Embodiments 1 and 2, the case where the power conversion device is a boost chopper was described, but in Embodiment 3, the case of a buck chopper will be described. Figure 13 FIG. is a circuit diagram of the power conversion device in Embodiment 3, which is a circuit diagram when the power conversion device is a buck chopper. In addition, Figure 13 the description of the control unit is omitted. The power conversion unit 302 is connected to the DC voltage source 101 and is driven by a control unit 103 (not shown).

[0134] The power conversion unit 302 includes a buck chopper circuit, and the buck chopper circuit includes semiconductor switch elements Q1 and Q2 each formed of a MOSFET and connected in series with each other. Specifically, the smoothing capacitor 104 is connected in parallel with the DC voltage source 101, and the connection point between the positive electrode side of the DC voltage source 101 and the positive electrode side of the smoothing capacitor 104 is connected to the drain terminal of the semiconductor switch element Q1. The connection point between the source terminal of the semiconductor switch element Q1 and the drain terminal of the semiconductor switch element Q2 is connected to the positive electrode side of the smoothing capacitor 108 serving as the output side via the reactor 105. The source terminal of the semiconductor switch element Q2 is connected to the negative electrode side of the DC voltage source 101 and the negative electrode side of the smoothing capacitor 108. A current detector 110 for detecting the inductor current is provided between the connection point between the source terminal of the semiconductor switch element Q1 and the drain terminal of the semiconductor switch element Q2 and the reactor 105. In addition, freewheeling diodes D1 and D2 are respectively connected in anti-parallel to the semiconductor switch elements Q1 and Q2. Other configurations are the same as those in Embodiment 1. In addition, since the buck chopper only has the opposite input-output relationship to the boost chopper and the basic operation is the same as that of the boost chopper, the description is omitted.

[0135] As described above, in Embodiment 3, the basic operation is also the same as that in Embodiment 1. Therefore, similar to Embodiment 1, by detecting the input voltage, output voltage, and inductor current and changing the frequency command value synchronously with the carrier, the same effects as those in Embodiment 1 can also be obtained in Embodiment 3.

[0136] Embodiment 4.

[0137] Next, based onFigures 14 to 19 Embodiment 4 will be described. In Embodiment 4, the case where the power conversion device is a buck-boost chopper will be described. Figure 14 FIG. is a circuit diagram of the power conversion device in Embodiment 4, which is a circuit diagram when the power conversion device is a buck-boost chopper. In addition, Figure 14 the description of the control unit is omitted. The power conversion unit 402 is connected to the DC voltage source 101 and is driven by the control unit 103 (not shown).

[0138] The power conversion unit 402 includes a buck-boost chopper circuit, which includes semiconductor switch elements Q1 and Q2 each formed of a MOSFET and connected in series with each other. Specifically, a smoothing capacitor 104 and a reactor 105 are connected in parallel with the DC voltage source 101, and the connection point between the positive electrode side of the DC voltage source 101 and the positive electrode side of the smoothing capacitor 104 is connected to the drain terminal of the semiconductor switch element Q1. The connection point between the source terminal of the semiconductor switch element Q1 and the drain terminal of the semiconductor switch element Q2 is connected to one end of the reactor 105. The other end of the reactor 105 is connected to the negative electrode side of the smoothing capacitor 108 on the output side. The source terminal of the semiconductor switch element Q2 is connected to the positive electrode side of the smoothing capacitor 108. A current detector 110 for detecting the inductor current is provided between the connection point between the source terminal of the semiconductor switch element Q1 and the drain terminal of the semiconductor switch element Q2 and the reactor 105. In addition, freewheeling diodes D1 and D2 are respectively anti-parallel connected to the semiconductor switch elements Q1 and Q2. Other configurations are the same as those in Embodiment 1. In the buck-boost chopper, the basic operation is also the same as that of the boost chopper and the buck chopper. However, since the voltage value and the current value of the switching loss calculation value are different, the following description will be given.

[0139] Figure 15A FIG. is a circuit diagram of the buck-boost chopper according to Embodiment 4, Figure 15B FIG. is a diagram showing the generation mode of the switching loss of the power conversion device in Embodiment 4. Figure 15A In FIG., I_Q1 is the current flowing through the semiconductor switch element Q1, and I_Q2 is the current flowing through the semiconductor switch element Q2. In addition, I_D1 is the current flowing through the freewheeling diode D1, and I_D2 is the current flowing through the freewheeling diode D2. In addition, in the following description, the inductor current IL flowing through the reactor 105 is always set to be positive. In addition, the positive directions of the respective currents and voltages are Figure 15A as shown.

[0140] Figure 15BAmong them, the state at the moment when the inductor current IL is detected is set as the initial state. In the initial state, the gate signal GQ1 is on (the semiconductor switching element Q1 is on), and the gate signal GQ2 is off (the semiconductor switching element Q2 is off). When the control period is set as T, according to the duty ratio command value Dref, at the moment t11 (= T×Dref / 2), the gate signal GQ1 is off (the semiconductor switching element Q1 is off). At the moment t12 (= t11 + td), the gate signal GQ2 is on (the semiconductor switching element Q2 is on). Additionally, td is the dead time. At the moment t13 (= t11 + T×(1 - Dref)), the gate signal GQ2 is off (the semiconductor switching element Q2 is off). At the moment t14 (= t13 + td), the gate signal GQ1 is on again (the semiconductor switching element Q1 is on).

[0141] As Figure 15B shown, at the moment t11, the turn-off loss (Eoff_Q1) of the semiconductor switching element Q1 occurs, and at the moment t2, the turn-on loss (Eon_Q2) of the semiconductor switching element Q2 occurs. In addition, at the moment t13, the turn-off loss (Eoff_Q2) of the semiconductor switching element Q2 occurs, and at the moment t14, the turn-on loss (Eon_Q1) of the semiconductor switching element Q1 occurs. And, at the moment t14 when the semiconductor switching element Q1 is on, the recovery loss (Erec) occurs in the freewheeling diode D2. In addition, the inductor current IL is equal to the inductor current detection value IL_det in the initial state, but then varies with the period T. The variation amplitude during the vibration period is ΔIL.

[0142] Figure 16 is a diagram showing the current path before the semiconductor switching element Q1 is turned off in the power conversion device in Embodiment 4. Figure 16 Among them, Vin represents the input voltage, and Vout represents the output voltage. As Figure 16 shown, the MOSFET of the semiconductor switching element Q1 is on, so the current flowing through the buck-boost chopper circuit becomes equal to the inductor current IL.

[0143] According to Figure 15B and Figure 16 , when calculating the turn-off loss generated in the semiconductor switching element Q1 in the buck-boost chopper, the following equations (19) and (20) are used to calculate the drain current ID_off_Q1 and the drain-source voltage VDS_off_Q1 when the semiconductor switching element Q1 is turned off.

[0144] [Equation 19]

[0145]

[0146] [Equation 20]

[0147] V DS_off_Q1 = V in_det -V out_det

[0148] …(20)

[0149] Figure 17 FIG. is a diagram showing the current path before the semiconductor switch element Q2 is turned on in the power conversion device according to Embodiment 4, and shows the current path during the dead time after the semiconductor switch element Q1 is turned off. During the dead time after the semiconductor switch element Q1 is turned off, the current flowing through the freewheeling diode D2 is equal to the inductor current IL.

[0150] According to Figure 15B and Figure 17 , when calculating the conduction loss generated in the semiconductor switch element Q2 in the buck-boost chopper, the following equations (21) and (22) are used to calculate the drain current ID_on_Q2 and the drain-source voltage VDS_on_Q2 when the semiconductor switch element Q2 is turned on.

[0151] [Equation 21]

[0152]

[0153] [Equation 22]

[0154] V DS_on_Q2 = V fwd_D2 …(22)

[0155] Figure 18 FIG. is a diagram showing the current path before the semiconductor switch element Q2 is turned off in the power conversion device according to Embodiment 4. As Figure 18 shown, the MOSFET of the semiconductor switch element Q2 is turned on, and thus the current flowing through the buck-boost chopper circuit becomes equal to the inductor current IL.

[0156] According to Figure 15B and Figure 18 , when calculating the turn-off loss generated in the semiconductor switch element Q2 in the buck-boost chopper, the following equations (23) and (24) are used to calculate the drain current ID_off_Q2 and the drain-source voltage VDS_off_Q2 when the semiconductor switch element Q2 is turned off.

[0157] [Equation 23]

[0158]

[0159] [Equation 24]

[0160] VDS_oft_Q2 = V fwd_D2 …(24)

[0161] Figure 19 FIG. is a diagram showing a current path before the semiconductor switch element Q1 is turned on in the power conversion device according to Embodiment 4, and shows the current path during the dead time after the semiconductor switch element Q2 is turned off. During the dead time after the semiconductor switch element Q2 is turned off, the current flowing through the freewheeling diode D2 is equal to the inductor current IL.

[0162] According to Figure 15B and Figure 19 , when calculating the conduction loss generated in the semiconductor switch element Q1 in the buck-boost chopper, the following equations (25) and (26) are used to calculate the drain current ID_on_Q1 and the drain-source voltage VDS_on_Q1 when the semiconductor switch element Q1 is turned on.

[0163] [Equation 25]

[0164]

[0165] [Equation 26]

[0166] V DS_on_Q1 = V in_det - V out_det

[0167] …(26)

[0168] In addition, when calculating the recovery loss generated in the freewheeling diode D2, the following equations (27) and (28) are used to calculate the anode current IA_rec and the cathode-anode voltage VKA_rec when the freewheeling diode D1 recovers.

[0169] [Equation 27]

[0170]

[0171] [Equation 28]

[0172] V KA_rec = V in_det - V out_det

[0173] …(28)

[0174] As described above, in Embodiment 4, the current values and voltage values required to calculate the respective losses that make up the loss calculation value Ploss are different from those in Embodiment 1. On the other hand, the variation in the inductor current IL caused by the influence of the switch is the same as that in Embodiment 1. Therefore, as in Embodiment 1, the input voltage, output voltage, and inductor current are detected synchronously with the carrier wave, and the frequency command value is changed, so that the same effects as those in Embodiment 1 can also be obtained in Embodiment 4.

[0175] Embodiment 5.

[0176] Next, based on Figure 20A and Figure 20B Embodiment 5 will be described. The difference between Embodiment 5 and Embodiment 1 is that an IGBT (Insulated Gate Bipolar Transistor) is used as the semiconductor switching element. Figure 20A is a circuit diagram of the power conversion device in Embodiment 5, which is a circuit diagram when an IGBT is used as the switching element of the boost chopper. In addition, Figure 20A the description of the control unit is omitted. The power conversion unit 502 is connected to the DC voltage source 101 and is driven by the control unit 103 (not shown). The power conversion unit 502 has a boost chopper circuit, and this boost chopper circuit includes semiconductor switching elements Q1* and Q2* that are each composed of an IGBT and are connected in series with each other. Comparing Figure 20A with Figure 4A it can be seen that in Embodiment 5, only the semiconductor switching elements Q1 and Q2 in Embodiment 1 are replaced with semiconductor switching elements Q1* and Q2*, respectively. Therefore, the description of other structures is omitted.

[0177] Figure 20B is a diagram showing the generation mode of the switching loss of the power conversion device in Embodiment 5. Figure 20B In, the state at the time when the inductor current IL is detected is set as the initial state. When an IGBT is used as the semiconductor switching element of the boost chopper, the gate signal GQ1 is always set to off (the semiconductor switching element Q1* is always off). The gate signal GQ2 is off in the initial state (the semiconductor switching element Q2* is off). When the control period is set to T, according to the duty ratio command value Dref, at time t21 (= T×(1 - Dref) / 2), the gate signal GQ2 is on (the semiconductor switching element Q2* is on). At time t22 (= t21 + T×Dref), the gate signal GQ2 is off (the semiconductor switching element Q2* is off).

[0178] As Figure 20BAs shown, the conduction loss (Eon_Q2) of the semiconductor switching element Q2* and the recovery loss (Erec) of the freewheeling diode D1 are generated at time t21, and the turn-off loss (Eoff_Q2) of the semiconductor switching element Q2* is generated at time t22. In addition, the inductor current IL is equal to the inductor current detection value IL_det in the initial state, but then varies with a period T. The variation amplitude during the vibration period is ΔIL.

[0179] In the case of using an IGBT as the semiconductor switching element of the boost chopper, the iron loss Pi r and the copper loss Pcop do not change compared with the case of the MOSFET. On the other hand, the switching loss Psw and the conduction loss Pcon change. The semiconductor switching element Q1* is always off and does not perform switching, so the conduction loss and turn-off loss related to the semiconductor switching element Q1* are not generated. Therefore, the conduction loss Eon_Q1 and the turn-off loss Eoff_Q1 in Equation (6) are zero. Regarding the conduction loss and turn-off loss related to the semiconductor switching element Q2* and the recovery loss of the freewheeling diode D1, Equations (11) to (16) are used for calculation. In addition, in the case of an IGBT, the collector is equivalent to the drain of the MOSFET, and the emitter is equivalent to the source of the MOSFET.

[0180] Regarding the conduction loss Pcon, since the semiconductor switching element Q1* is not conducting, it is generated in the freewheeling diode D1 and the semiconductor switching element Q2*. The conduction loss Pcon in this case is calculated using the following Equation (29).

[0181] [Equation 29]

[0182] P con =(V on_CE +V fwd_D1 )I L_det

[0183] …(29)

[0184] In Equation (29), Von_CE is the conduction voltage between the collector and the emitter, which is a value determined by the current flowing through the semiconductor switching element Q2*.

[0185] As described above, in Embodiment 5, a part of each loss constituting the loss calculation value Ploss is different from that in Embodiment 1. On the other hand, the variation of the inductor current IL caused by the influence of the switch is the same as that in Embodiment 1. Therefore, similar to Embodiment 1, the input voltage, output voltage, and inductor current are detected and the frequency command value is changed synchronously with the carrier, so that the same effect as in Embodiment 1 can also be obtained in Embodiment 5. In addition, the case of the boost chopper is described in Embodiment 5, but the buck chopper is the same.

[0186] Embodiment 6

[0187] Next, based on Figure 21A and Figure 21B Embodiment 6 will be described. In Embodiment 6, in the buck-boost chopper described in Embodiment 4, an IGBT is used as the semiconductor switching element instead of the MOSFET. Figure 21A is a circuit diagram of the power conversion device in Embodiment 6, which is a circuit diagram when an IGBT is used as the switching element of the buck-boost chopper. In addition, Figure 21A the description of the control unit is omitted. The power conversion unit 602 is connected to the DC voltage source 101 and is driven by the control unit 103 (not shown). The power conversion unit 602 has a buck-boost chopper circuit, and this buck-boost chopper circuit includes semiconductor switching elements Q1* and Q2* that are each constituted by an IGBT and are connected in series with each other. Comparing Figure 21A with Figure 15A it can be seen that in Embodiment 6, only the semiconductor switching elements Q1 and Q2 in Embodiment 4 are replaced with semiconductor switching elements Q1* and Q2*, respectively, so the description of other structures is omitted.

[0188]

Figure 20

[0189] As Figure 21B shown, the turn-off loss (Eoff_Q1) of the semiconductor switching element Q1* is generated at the moment t31, and the turn-on loss (Eon_Q1) of the semiconductor switching element Q1* and the recovery loss (Erec) of the freewheeling diode D2 are generated at the moment t32. In addition, the inductor current IL is equal to the inductor current detection value IL_det in the initial state, but then varies with a period T. The variation amplitude during the vibration period is ΔIL.

[0190] In the case of using an IGBT as a semiconductor switching element of a buck-boost chopper, the iron loss Pir and copper loss Pcop are unchanged from the case of a MOSFET. On the other hand, the switching loss Psw and conduction loss Pcon change. The semiconductor switching element Q2* is always turned off without switching, so no conduction loss and turn-off loss associated with the semiconductor switching element Q2* are generated. Therefore, the conduction loss Eon_Q2 and turn-off loss Eoff_Q2 in Equation (6) are zero. Regarding the conduction loss and turn-off loss associated with the semiconductor switching element Q1*, and the recovery loss of the freewheeling diode D2, Equations (19), (20), and (25) to (28) are used for calculation.

[0191] Regarding the conduction loss Pcon, since the semiconductor switching element Q2* is not conducting, it is generated in the freewheeling diode D2 and the semiconductor switching element Q1*. The conduction loss Pcon in this case is calculated using the following Equation (30).

[0192] [Equation 30]

[0193] P con =(V on _ CE +V fwd_D2 )I L_det …(30)

[0194] Von_CE in Equation (30) is the conduction voltage between the collector and emitter, and is a value determined by the current flowing through the semiconductor switching element Q1*.

[0195] As described above, in Embodiment 6, a part of each loss constituting the loss calculation value Ploss is different from that in Embodiment 4. On the other hand, the variation of the inductor current IL caused by the influence of the switch is the same as that in Embodiment 4. Therefore, similarly to Embodiment 4, the input voltage, output voltage, and inductor current are detected synchronously with the carrier, and the frequency command value is changed, so that the same effect as in Embodiment 4 can also be obtained in Embodiment 6.

[0196] Although this application describes various exemplary embodiments and examples, the various features, methods, and functions described in one or more embodiments are not limited to the application of a specific embodiment, and can be applied alone or in various combinations to the embodiments.

[0197] Therefore, countless unillustrated variations can be envisioned within the scope of the technology disclosed in this application.

[0198] Reference Numeral Explanation

[0199] 100 Power conversion device

[0200] 101 DC voltage source

[0201] 102, 302, 402, 502, 602 Power conversion units

[0202] 103 Control unit

[0203] 105 Reactor

[0204] 109, 111 Voltage detectors

[0205] 110 Current detector

[0206] 112 Voltage control unit

[0207] 113, 213 Frequency control units

[0208] 113a Loss calculation unit

[0209] 113a3 Iron loss calculation unit

[0210] 113a4 Copper loss calculation unit

[0211] 113a5 Switching loss calculation unit

[0212] 113a6 Conduction loss calculation unit

[0213] 113b, 213b Frequency determination units

[0214] 114 Carrier generation unit

[0215] D1, D2 Flyback diodes

[0216] Q1, Q1*, Q2, Q2* Semiconductor switching elements

[0217] IL Inductor current

[0218] IL_det Inductor current detection value

[0219] Vin Input voltage

[0220] Vin_det Input voltage detection value

[0221] Vout Output voltage

[0222] Vout_det Output voltage detection value

[0223] Dref Duty ratio command value

[0224] fsw Switching frequency

[0225] fsw_ref Frequency command value

[0226] car Carrier

[0227] Ploss Loss calculation value

[0228] Pir iron loss

[0229] Pcop copper loss

[0230] Psw switching loss

[0231] Pcon conduction loss.

Claims

1. A power conversion device that converts an input voltage from a DC voltage source and outputs it as an output voltage, wherein the power conversion device is characterized by comprising: A plurality of semiconductor switching elements connected in series between the DC voltage source and the output side; An inductor connected between the DC voltage source and the output side; A control unit that controls the switching frequency of the plurality of semiconductor switching elements; And A detection unit that detects the voltage value of the input voltage, the voltage value of the output voltage, and the current value of the inductor current flowing through the inductor, The control unit includes: A voltage control unit that calculates an output voltage command value based on the voltage value of the input voltage and the voltage value of the output voltage obtained through the detection; A frequency control unit that calculates a frequency command value based on the voltage value of the input voltage, the voltage value of the output voltage, the current value of the inductor current, and the output voltage command value; and A carrier generation unit that generates a carrier based on the frequency command value, At a timing synchronized with the carrier, the detection based on the detection unit and the change of the switching frequency based on the control unit are performed.

2. The power conversion device according to claim 1, wherein The frequency control unit includes: A loss calculation unit that calculates the loss generated during the operation of the power conversion device as a loss calculation value; and A frequency determination unit that obtains the switching frequency at which the loss is minimized based on the loss calculation value and outputs it as the frequency command value.

3. The power conversion device according to claim 2, wherein When the carrier is a triangular wave carrier, the detection and the change of the switching frequency are performed at a timing when the phase of the triangular wave carrier is 0 degrees or 180 degrees.

4. The power conversion device according to claim 2, wherein When the carrier is a sawtooth wave carrier, the timing is shifted from the peak and trough of the sawtooth wave carrier to perform the detection and the change of the switching frequency.

5. The power conversion device according to claim 2, wherein The loss calculation value includes at least one of the iron loss of the inductor, the copper loss of the inductor, the switching loss of the plurality of semiconductor switching elements, and the conduction loss of the plurality of freewheeling diodes connected in anti-parallel with the plurality of semiconductor switching elements.

6. The power conversion device according to claim 2, wherein The frequency determination unit obtains the switching frequency at which the loss calculation value is minimized by the hill climbing method.

7. The power conversion device according to claim 6, wherein The loss calculation value includes at least one of the iron loss of the inductor, the copper loss of the inductor, the switching loss of the plurality of semiconductor switching elements, and the conduction loss of the plurality of freewheeling diodes connected in anti-parallel with the plurality of semiconductor switching elements.

8. The power conversion device according to claim 7, wherein: the loss calculation unit calculates the switching loss by using a data table indicating the relationship between the current flowing through the plurality of semiconductor switching elements, the voltage between each terminal, and the switching loss.

9. The power conversion device according to claim 7, wherein: the loss calculation unit calculates the switching loss by using an approximate formula composed of the current flowing through the plurality of semiconductor switching elements and the voltage between each terminal.

10. The power conversion device according to claim 1, wherein: when the carrier wave is a triangular wave carrier, the detection and the change of the switching frequency are performed at a timing when the phase of the triangular wave carrier is 0 degrees or 180 degrees.

11. The power conversion device according to claim 1, wherein: when the carrier wave is a sawtooth wave carrier, the timing is shifted from the peak and trough of the sawtooth wave carrier to perform the detection and the change of the switching frequency.

12. The power conversion device according to any one of claims 1 to 11, wherein: the voltage control unit changes a control parameter according to the frequency command value.

13. The power conversion device according to any one of claims 1 to 11, wherein: it includes a boost chopper, a buck chopper, or a buck-boost chopper.

Images