Synchronous control method for a two-way wireless charging system

Abstract

The present invention discloses a synchronous control method for a bidirectional wireless charging system. The synchronous control method includes the following steps: After power-on, calculate the mutual inductance of the bidirectional wireless charging system, and calculate the reference bypass duty ratio of the secondary rectifier during synchronization according to the reference value of the output voltage or current; When the output voltage or current rises to the reference value, turn on the power loop and the synchronization loop, and assign an initial value to the output bypass duty ratio of the power loop; Among them, the power loop performs closed-loop control of the output bypass duty ratio; The synchronization loop performs closed-loop phase-shift control of the synchronization signal, thereby controlling the phase synchronization between the synchronization signal and the zero-crossing point of the input current of the secondary rectifier. The present invention can achieve zero-voltage turn-on of the Mos in the secondary rectifier, does not require an AC sensor, reduces the system cost and improves the reliability, and solves the phase perturbation problem caused by the crystal oscillator frequency deviation between the primary and secondary controllers in the bidirectional wireless charging system.

Description

Technical Field

[0001] The present invention relates to the technical field of wireless charging, and more particularly to a synchronous control method for a bidirectional wireless charging system. Background Art

[0002] Wireless charging technology can transfer electrical energy from a power source to a load in a non-contact manner, with the advantages of convenience, speed, safety and reliability. In the field of electric vehicle charging, bidirectional wireless charging technology can achieve two-way energy allocation between the vehicle side and the power grid, which can improve the energy utilization efficiency and the power supply flexibility of the power grid, and has broad application prospects.

[0003] In a bidirectional wireless charging system, the primary side and the secondary side each adopt an independent controller to output control signals with the same frequency. However, due to the slight deviation of the crystal oscillator frequencies of different controllers, the phase difference between the primary and secondary control signals will change periodically during the control process, resulting in output power oscillation. Even if the primary and secondary controllers perform wireless communication, existing wireless communication means such as Bluetooth and Wi-Fi have a communication delay of the order of milliseconds, while the control signal period of the wireless charging system is generally of the order of microseconds, which is difficult to meet the real-time synchronization requirements of the control signals. Moreover, wireless communication signals are easily affected by electromagnetic interference, which is not conducive to the safety and stability of the system.

[0004] Therefore, in order to ensure the stability of power transmission in a bidirectional wireless charging system, a suitable phase synchronization method is needed to control the phase difference between the primary and secondary control signals. Currently, the existing technologies can be divided into two categories. The first category indirectly obtains the phase information of the primary side by adding additional hardware devices on the secondary side for synchronization, such as adding an auxiliary coil or an AC current sensor for high-frequency current sampling, and then controlling the phase difference between the primary and secondary sides. This method requires high hardware accuracy for the sampling part, and when the current harmonic content is high, the sampled phase information will shift and compensation is required, increasing the cost and design complexity of the system. The second category does not require additional hardware circuits and only requires the DC information of the load. Phase synchronization is achieved by the perturbation method to track the extreme value of the output current or the extreme value of the phase angle of the active rectifier bridge closed-loop bypass. This method is not affected by system parameter drift and detuning, but the system performance is affected by the perturbation period and the perturbation amount. When the frequency difference between the primary and secondary controllers is large, there is a disadvantage of large output ripple, and the perturbation method is only applicable to the symmetric bypass control of the active rectifier bridge (the rectifier circuit current is in phase with the voltage fundamental wave). For the soft-switching bypass control that can achieve the soft-switching of the rectifier bridge Mos (the rectifier circuit current leads the voltage fundamental wave), the extreme value of the output current or the phase angle of the rectifier circuit bypass does not correspond to the synchronization point, and the perturbation method cannot achieve phase synchronization at this time. Summary of the Invention

[0005] The object of the present invention is to provide a synchronous control method for a bidirectional wireless charging system. The present invention can achieve zero-voltage turn-on and synchronous control of the secondary rectifier, reduce the system cost and improve the reliability, and solve the phase perturbation problem caused by the crystal oscillator frequency deviation between the primary and secondary controllers in the bidirectional wireless charging system.

[0006] The technical solution of the present invention is as follows: A synchronous control method for a bidirectional wireless charging system, the bidirectional wireless charging system includes a primary circuit and a secondary circuit; the primary circuit includes an inverter circuit connected to a DC voltage source, and the output end of the inverter circuit is connected with a primary series compensation capacitor and a transmitting coil; the secondary circuit includes a secondary rectifier connected to a load, and the input end of the secondary rectifier is connected with a secondary series compensation capacitor and a receiving coil; the bidirectional wireless charging system realizes the control of the output voltage or current and the output power direction of the bidirectional wireless charging system by controlling the output bypass duty ratio and the phase duty ratio of the synchronous signal of the secondary rectifier, and the synchronous control method includes the following steps:

[0007] Step 1: After power-on, the primary circuit inputs a small power, the secondary rectifier enters an uncontrolled rectification mode, calculates the mutual inductance of the bidirectional wireless charging system, and calculates the reference bypass duty ratio of the secondary rectifier at synchronization according to the reference value of the output voltage or current.

[0008] Step 2: When the output voltage or current rises to the reference value, turn on the power loop and the synchronization loop, and assign an initial value to the output bypass duty ratio; wherein, the power loop performs closed-loop control of the output bypass duty ratio to reduce the error between the output voltage or current and the reference value; the synchronization loop performs closed-loop phase-shift control of the synchronous signal to reduce the error between the output bypass duty ratio and the reference bypass duty ratio, and further controls the phase synchronization between the synchronous signal and the zero-crossing point of the input current of the secondary rectifier.

[0009] In the above synchronous control method for a bidirectional wireless charging system, in step 1, the mutual inductance calculation is that the primary circuit transmits the primary DC bus voltage U of the small power input at this time to the secondary circuit through low-speed non-real-time wireless communication. in1 The secondary circuit calculates the mutual inductance M according to the following formula:

[0010]

[0011] In the formula: I O is the DC bus current in the secondary circuit; ω is the angular frequency of the bidirectional wireless charging system, ω = 2πf, and f is the operating frequency of the bidirectional wireless charging system.

[0012] In the foregoing synchronous control method for a bidirectional wireless charging system, the calculation of the reference bypass duty ratio of the secondary rectifier includes that when the secondary rectifier performs constant current control, the reference bypass duty ratio dref The calculation formula is as follows:

[0013]

[0014] Wherein, I O_ref is the reference value of the DC bus current of the secondary circuit, U in_rated is the DC bus voltage of the primary circuit under the rated power, M is the mutual inductance, ω is the angular frequency of the bidirectional wireless charging system, and d a_ref is the reference value of the duty cycle of the synchronization signal phase of the secondary circuit.

[0015] For the synchronization control method of the above-mentioned bidirectional wireless charging system, the calculation of the reference bypass duty cycle of the secondary rectifier includes that when the secondary circuit performs constant voltage control, the reference bypass duty cycle d ref The calculation formula is as follows:

[0016]

[0017] Wherein, U o_ref is the reference value of the DC bus voltage of the secondary circuit, U in_rated is the DC bus voltage of the primary circuit under the rated power, M is the mutual inductance, ω is the angular frequency of the bidirectional wireless charging system, and d a_ref is the reference value of the duty cycle of the synchronization signal phase of the secondary circuit; R L is the equivalent resistance of the load, U o and I o are the sampled values of the output DC voltage and current respectively.

[0018] For the synchronization control method of the above-mentioned bidirectional wireless charging system, the method of assigning the initial value of the output bypass duty cycle is as follows:

[0019] Set the reference current threshold I o_ref1 , when the reference current I o_ref is less than the reference current threshold I o_ref1 , the initial value d initial is the reference bypass duty cycle d ref ;

[0020] When the reference current I o_ref is greater than the reference current threshold I o_ref1 , the initial value d initial is the bypass duty cycle threshold d1, and the calculation of the initial phase duty cycle is as follows:

[0021]

[0022] For the synchronization control method of the above-mentioned bidirectional wireless charging system, the reference current threshold I o_ref1 is solved by the following equation:

[0023]

[0024] The synchronization control method of the aforementioned bidirectional wireless charging system, the synchronization control method further includes the reverse flow of the output power. At this time, the output bypass duty ratio of the power loop remains unchanged, and the synchronization loop controls the synchronization signal to gradually shift the phase by 180°. After that, the power loop is restored to closed-loop control to achieve the reverse flow of power.

[0025] Compared with the prior art, the present invention has the following beneficial effects:

[0026] 1. The present invention only needs to collect the output information of the secondary side DC side of the bidirectional wireless charging system, does not require an AC current sensor to collect the high-frequency phase information of the current, and thus does not require a complex high-frequency filtering circuit, reducing the design complexity of the system and lowering the system cost.

[0027] 2. The present invention does not require real-time and fast wireless communication between the primary and secondary sides, and only requires low-speed and non-real-time wireless communication during the startup phase to achieve phase synchronization between the primary and secondary sides, solving the phase perturbation problem caused by the crystal oscillator frequency deviation of the primary and secondary side controllers in the bidirectional wireless charging system and improving the system reliability.

[0028] 3. Compared with the synchronization control method of the WPT system that collects DC information by the perturbation method, the synchronization method of the present invention adopts closed-loop feedback control, has a faster response speed and good dynamic performance, does not have the problem that the system performance of the perturbation method is affected by the perturbation period and the perturbation amount, and is designed for the soft-switching bypass control (the current of the rectifier circuit leads the voltage fundamental wave) that can realize the soft switching of the rectifier bridge Mos, reducing the switching loss of the rectifier bridge and being beneficial to improving the system efficiency. Description of the Drawings

[0029] Figure 1 is the basic circuit topology diagram of the bidirectional wireless charging system of the present invention;

[0030] Figure 2 is the drive control waveform of the 4 Mos transistors on the secondary side and the Sync waveform of the synchronization signal;

[0031] Figure 3 is the schematic diagram of the current i e flow path at stage a;

[0032] Figure 4 is the schematic diagram of the current i e flow path at stage b;

[0033] Figure 5 is the schematic diagram of the current i e flow path at stage c;

[0034] Figure 6It is the current i at stage d e Schematic diagram of the current flow path;

[0035] Figure 7 It is the current i at stage e e Schematic diagram of the current flow path;

[0036] Figure 8 It is the current i at stage f e Schematic diagram of the current flow path;

[0037] Figure 9 It is the schematic diagram of the principle of the control method of the present invention;

[0038] Figure 10 It is the output DC current I O and the relationship curve of the duty cycle d of the synchronization signal Sync phase a ;

[0039] Figure 11 It is the reference bypass duty cycle d ref and the relationship curve of the duty cycle d of the synchronization signal Sync phase a ;

[0040] Figure 12 It is the output DC current I O , the relationship diagram of the output bypass duty cycle d and the duty cycle d of the synchronization signal Sync phase a ;

[0041] Figure 13 It is the duty cycle d of the synchronization signal Sync phase a When it is less than 0, the output power loop DC current I O Variation relationship curve with the output bypass duty cycle d;

[0042] Figure 14 It is the duty cycle d of the synchronization signal Sync phase a When it is greater than 0, the output power loop DC current I O Variation relationship curve with the output bypass duty cycle d;

[0043] Figure 15 It is the reference bypass duty cycle d ref and the relationship curve of the duty cycle d of the synchronization signal Sync phase a ;

[0044] Figure 16 It is the initial phase duty cycle |d a_initial |, the reference bypass duty cycle d ref and the relationship curve with the reference current I O_ref ;

[0045] Figure 17 It is the 3A constant current control simulation waveform of the resistive load;

[0046] Figure 18 is the simulation waveform of constant voltage 60V control for resistive load;

[0047] Figure 19 is the simulation waveform of 3A constant current control for battery load.

[0048] Figure 20 is the experimental waveform of constant current 3.5A to constant voltage 52.9V for battery load;

[0049] Figure 21 is the experimental waveform of constant current 3.5A and 2.5A for battery load. Detailed implementation manners

[0050] The present invention will be further described below in conjunction with the accompanying drawings and embodiments, but it shall not be used as a basis for limiting the present invention. The protection scope of the present invention shall include all the contents of the claims, and through the following embodiments, those of ordinary skill in the art can more comprehensively understand the present invention.

[0051] Embodiment: A synchronous control method for a bidirectional wireless charging system, the basic circuit topology of the bidirectional wireless charging system is as Figure 1 shown, including a primary circuit and a secondary circuit; the primary circuit includes an inverter circuit connected to a DC voltage source, and the output end of the inverter circuit is connected with a primary series compensation capacitor and a transmitting coil; the secondary circuit includes a secondary rectifier connected to a load, and the input end of the secondary rectifier is connected with a secondary series compensation capacitor and a receiving coil;

[0052] In this embodiment, the inverter circuit is composed of an H-bridge formed by four Mos transistors S1, S2, S3, and S4, and is used to convert direct current into alternating current. The transmitting coil is used to transmit the electric energy transmitted by the inverter circuit through a magnetic field, and the receiving coil is used to receive the energy transmitted by the transmitting coil.

[0053] The primary series compensation capacitor C p is configured according to the formula:

[0054]

[0055] wherein, L p is the self-inductance of the transmitting coil, ω is the angular frequency of the bidirectional wireless charging system, ω = 2πf, and f is the operating frequency of the bidirectional wireless charging system.

[0056] The parameters of the secondary series compensation capacitor C S are configured according to the formula:

[0057]

[0058] wherein, Ls is the self - inductance of the receiving coil.

[0059] The bidirectional wireless charging system configured according to the above formula has a constant - current source output characteristic, and the input current I of the rectifier e is not affected by the load change, and the formula is as follows.

[0060]

[0061] Among them, I e is the amplitude of the input current of the secondary rectifier, and U in is the input voltage of the primary DC bus (i.e., the voltage of the DC voltage source).

[0062] In this embodiment, the input voltage of the primary DC bus, the output current and voltage of the secondary DC bus are collected by a DC current - voltage sensor.

[0063] The rectifying circuit is composed of 4 Mos transistors S5, S6, S7, and S8, and is used to convert alternating current into direct current. The drive - control waveforms of the 4 Mos transistors of the rectifying circuit and the Sync waveform of the synchronization signal are as Figure 2 shown, where β is the phase angle of the 0 - level of the secondary rectified voltage, is the phase difference between the secondary synchronization signal Sync and the zero - crossing point of the rectifier input current. When the synchronization signal Sync lags behind the zero - crossing point of the secondary rectifier input current For the convenience of subsequent description, define d as the bypass duty ratio of the secondary rectifier, Define d a as the phase - duty ratio of the secondary synchronization signal Sync and the zero - crossing point of the rectifier input current,

[0064] The Mos transistors S5, S6, S7, S8 and the synchronization signal Sync are all digital square waves with a 0.5 duty ratio inside the secondary controller, and the frequency is preset to the system operating frequency. The control logic is: the rising edge of the drive - control waveform of S5 is triggered and synchronized by the rising edge of the synchronization signal Sync. The pulse widths of S5 and S7 are (π / 2 + β) / (2πf), the phase angle by which S7 lags behind S5 is fixed at π / 2, the drive - control waveform of S6 is complementary to that of S5, and the drive - control waveform of S7 is complementary to that of S8.

[0065] According to Figure 2 , the secondary control waveform can be specifically divided into 6 stages:

[0066] Stage a: During the positive - half cycle of the rectifier input current i e , due to the lag of the synchronization signal Sync, when the current i e passes through the zero - crossing point, S6 and S7 are turned on, and the current - flowing path of the current i e is asFigure 3 As shown, the instantaneous power of the system flows reversely from the DC side to the AC side.

[0067] Phase b: S6 is turned off and S5 is turned on. At this time, S5 and S7 are conducting, and the secondary side is in a bypass state. The bypass duty ratio d of the secondary rectifier is determined by the secondary controller, and the current i e flows as shown in Figure 4 As shown, the instantaneous power of the system is not transferred to the DC side.

[0068] Phase c: After the bypass ends, S7 is turned off and S8 is turned on. At this time, S5 and S8 are conducting, and the current i e flows as shown in Figure 5 As shown, the instantaneous power of the system flows forward from the AC side to the DC side.

[0069] Phase d: During the negative half-cycle of the rectifier input current i e , due to the lag of the synchronization signal Sync, the current i e passes through the zero-crossing point. At this time, S5 and S8 are still conducting, and the current i e flows as shown in Figure 6 As shown, the instantaneous power of the system flows reversely from the DC side to the AC side.

[0070] Phase e: S5 is turned off and S6 is turned on. At this time, S6 and S8 are conducting, and the secondary side is in a bypass state. The bypass duty ratio d of the secondary rectifier is determined by the secondary controller, and the current i e flows as shown in Figure 7 As shown, the instantaneous power of the system is not transferred to the DC side.

[0071] Phase f: After the bypass ends, S8 is turned off and S7 is turned on. At this time, S6 and S7 are conducting, and the current i e flows as shown in Figure 8 As shown, the instantaneous power of the system flows forward from the AC side to the DC side.

[0072] From Figure 2 the relationship between the output DC current I O of the secondary side and the output bypass duty ratio d of the secondary rectifier and the phase duty ratio d a of the synchronization signal Sync can be derived as follows:

[0073]

[0074] It can be seen that the bidirectional wireless charging system of the present invention realizes the control of the output voltage or current and the output power direction of the bidirectional wireless charging system by controlling the output bypass duty ratio d of the secondary rectifier and the phase duty ratio d a of the synchronization signal, and due to the current i eThe zero-crossing leading synchronization signal Sync can achieve zero-voltage turn-on of the four Mosfets on the secondary side, reducing switching losses.

[0075] Based on the above bidirectional wireless charging system, as Figure 9 shown, the synchronization control method in this embodiment includes the following steps:

[0076] Step 1: After power-on, the primary circuit inputs a small power. The secondary rectifier enters the uncontrolled rectification mode, calculates the mutual inductance of the bidirectional wireless charging system, and calculates the reference bypass duty ratio of the secondary rectifier at synchronization according to the reference value of the output voltage or current.

[0077] The mutual inductance calculation is that the primary circuit transmits the primary DC bus voltage U in1 at this time of small power input and the primary DC bus voltage U in_rated at rated power input to the secondary circuit through low-speed non-real-time wireless communication. The secondary circuit calculates the mutual inductance M according to the following formula:

[0078]

[0079] In the formula: I o is the DC bus current in the secondary circuit; ω is the angular frequency of the bidirectional wireless charging system, ω = 2πf, and f is the operating frequency of the bidirectional wireless charging system. In other embodiments, the present invention can also adopt other common mutual inductance identification methods.

[0080] After calculating the mutual inductance, the calculation of the reference bypass duty ratio of the secondary rectifier includes that when the secondary rectifier performs constant current control, the reference bypass duty ratio d ref is calculated as follows:

[0081]

[0082] Among them, I o_ref is the reference value of the DC bus current in the secondary circuit, U in_rated is the primary circuit DC bus voltage under rated power, M is the mutual inductance, ω is the angular frequency of the bidirectional wireless charging system, and d a_ref is the reference value of the duty ratio of the synchronization signal phase in the secondary circuit.

[0083] When the secondary performs constant voltage control, the reference bypass duty ratio d ref is calculated as follows:

[0084]

[0085] Among them, U o_ref is the reference value of the DC bus voltage in the secondary circuit, U in_ratedis the DC bus voltage of the primary circuit under the rated power, M is the mutual inductance, ω is the angular frequency of the bidirectional wireless charging system, d a_ref is the reference value of the duty cycle of the synchronous signal phase in the secondary circuit; R L is the equivalent resistance of the load, U o and I o are the sampled values of the output DC voltage and current respectively.

[0086] Step 2: Input the rated DC bus voltage on the primary side. When the output voltage or current rises to the reference value, turn on the power loop and the synchronization loop, and assign an initial value d to the output bypass duty cycle d initial ; Among them, the power loop performs closed-loop control of the output bypass duty cycle to reduce the error between the output voltage or current and the reference value; the synchronization loop performs closed-loop phase-shifting control of the synchronization signal Sync to reduce the error between the output bypass duty cycle d and the reference bypass duty cycle d ref , and then control the phase synchronization of the synchronization signal and the zero-crossing point of the input current of the secondary rectifier, that is, control d a to be a suitable constant value d a_ref .

[0087] When power reverse flow is required, at this time, the output bypass duty cycle of the power loop remains unchanged, the synchronization loop controls the synchronization signal to gradually shift phase by 180°, and then makes the power loop resume closed-loop control to achieve power reverse flow.

[0088] In this step, the method of assigning the initial value of the output bypass duty cycle is as follows:

[0089] Set the reference current threshold I o_ref1 , and the reference current threshold I o_ref1 is solved by the following equation:

[0090]

[0091] For the sake of simple calculation, in this embodiment, an approximate numerical solution formula is used as the simplification of the above formula, and this approximate numerical solution formula is as follows:

[0092] In this formula, d a_ref = 0.1;

[0093] When the reference current I o_ref is less than the reference current threshold I o_ref1 , the initial value d initial is the reference bypass duty cycle d ref ;

[0094] When the reference current I o_ref is greater than the reference current threshold I o_ref1 , the initial value d initialFor the bypass duty cycle threshold d1, the calculation of d1 is as follows:

[0095]

[0096] It can be seen that through the closed-loop control of the power loop and the synchronization loop, the present invention realizes the control of the output voltage / current, the output power direction and the synchronization state of the bidirectional wireless charging system, and is applicable to both resistive and battery loads.

[0097] To further illustrate the beneficial effects of the present invention, this embodiment explains the power disturbance caused by the frequency difference between the primary and secondary side controllers of the bidirectional wireless charging system and the inapplicability of the extreme value perturbation method to the soft-switching bypass synchronization control in the background art.

[0098] First, for the scheme in which the primary side and the secondary side of the bidirectional wireless charging system each adopt an independent controller. Among them, the control signal period of the primary side controller determines the period of the input current i e of the secondary side rectifier, and the control signal period of the secondary side controller determines the period of the secondary side synchronization signal Sync. Due to the slight deviation of the crystal oscillator frequencies of different controllers, there is also a slight period difference ΔT in the control signal periods of the primary and secondary side controllers. The period difference ΔT will cause the phase duty cycle d e of i a and the synchronization signal Sync to change periodically. As can be seen from the formula, the periodic change of d a will cause the periodic change of the output DC current I O , as shown in Figure 10 .

[0099] Taking the crystal oscillator accuracy of the primary and secondary side controllers as 50 ppm as an example (1 ppm = 10 -6 ), when the system operating frequency is set to 85 kHz and the main control frequencies of the primary and secondary side single-chip microcomputers are selected as 72 MHz, the control signal period difference ΔT between the primary and secondary sides is 0.59 ns. According to the formula:

[0100]

[0101] it can be calculated that the fluctuation period T power of the system output current at this time is 0.24 s. Therefore, a suitable synchronization control method is required to keep d a constant.

[0102] Regarding the inapplicability of the extreme value perturbation method to the soft-switching bypass synchronization control in the background art, it is explained. As can be seen from Figure 2 , the secondary side control method of the present invention is soft-switching bypass control, and the synchronization signal Sync needs to lag behind the current i e to realize the ZVS (zero voltage turn-on) of the four Mos on the secondary side, that is, d aIt needs to be greater than or equal to 0. If the current extreme value perturbation method is used for synchronization, from Figure 10 it can be seen that when the power is transmitted in the forward direction and the output bypass duty cycle d is greater than 0, the output current I o corresponding to the maximum value of d a is less than 0. At this time, ZVS of the 4 Mos on the secondary side cannot be achieved, that is, the synchronization method of the traditional current extreme value perturbation method is not applicable to the soft-switching bypass control.

[0103] When the bypass duty cycle extreme value perturbation method is used for synchronization, taking the reference current I O_ref as 3A as an example, the reference bypass duty cycle d ref can be obtained from the formula, and the relationship with the duty cycle d a of the synchronization signal Sync phase is as Figure 11 shown.

[0104] From Figure 11 it can be seen that the extreme point of the reference bypass duty cycle d ref also corresponds to the interval where d a is less than 0, and there is also a situation where the same reference current I a corresponds to two bypass duty cycles d o_ref in the interval where d ref is less than 0. It can be seen that the bypass duty cycle extreme value perturbation method is also not applicable to the synchronization of the soft-switching bypass control in the present invention.

[0105] Therefore, for the soft-switching bypass control, the present invention proposes a synchronization control method based on closed-loop feedback control according to the reference value of the secondary rectifier bypass duty cycle during synchronization. The description of the power loop and the synchronization loop in the solution of the present invention is as follows:

[0106] According to the formula in the previous text, the output DC-side current I o of the secondary side under steady state, the output bypass duty cycle d of the secondary rectifier, and the duty cycle d a of the synchronization signal Sync phase are obtained, and the relationship is as Figure 12 shown.

[0107] Considering forward charging and keeping the duty cycle d a of the synchronization signal Sync phase unchanged, the curve of the output power loop DC current I o changing with the bypass duty cycle d can be obtained, as Figure 13 and Figure 14 shown.

[0108] From Figure 13 it can be seen that when d a is less than 0, that is, the synchronization signal Sync leads the rectifier circuit i e , I oThe curve showing the variation relationship with the bypass duty ratio d is non - monotonic. Among them, is determined by the following formula:

[0109]

[0110] Let We can obtain the d corresponding to the maximum output current, and the calculation formula is as follows:

[0111] d1 = |d a |;

[0112] Assume that when d is equal to 0, I o is equal to I o_ref Then d1 can be further expressed as:

[0113]

[0114] If 0 < d < d1, then the output DC current I o increases with the increase of the output bypass duty ratio d, showing monotonicity. At this time, the control monotonicity is If d1 < d < 1, then the output DC current I O decreases with the increase of the output bypass duty ratio d, showing monotonicity. At this time, the control monotonicity is It can be seen that the overall control logic of the DC current I o and the output bypass duty ratio d is non - monotonic.

[0115] From Figure 14 it can be seen that when d a is greater than 0, that is, when the synchronization signal Sync lags behind the rectifier circuit i e , the output DC current I o decreases with the increase of the output bypass duty ratio d, showing monotonicity. At this time, the control monotonicity is The overall control logic is monotonic.

[0116] Since all the rectifier Mos need to achieve soft - switching when the steady - state d a is greater than or equal to 0, so the control monotonicity of the monotonic power loop is selected as That is, the output DC current I o decreases with the increase of the output bypass duty ratio d as the control logic of the power loop. Considering that when d a is too large, the reactive component of the rectifier impedance will increase, resulting in a decrease in the AC - AC efficiency. Therefore, the phase - reference duty ratio d a_ref of the synchronization signal Sync is set to 0.1. The steady - state operating point of the power loop is point C as shown in Figure 14 , Figure 13 and Figure 14 the output DC current I ois 3A, and the output DC current I corresponding to point D o is d a is the maximum current when

[0117] For the synchronization loop, taking the reference current I o_ref as 3A as an example, d a_initial is defined as keeping the output bypass duty cycle d at 0 when starting up. When the output DC current I o rises to the reference current I o_ref , the reference bypass duty cycle d ref and the phase duty cycle d of the synchronization signal Sync a are related as Figure 15 shown, and the monotonicity is determined by the following formula

[0118]

[0119] Let be equal to 0, and the phase duty cycle d corresponding to the maximum value of the reference bypass duty cycle d ref can be obtained as a :

[0120]

[0121] It can be found from the above formula that the absolute value of d a1 is the same as that of d1.

[0122] From Figure 15 it can be seen that there are also two monotonicities between the reference bypass duty cycle d ref and the phase duty cycle d a , and when d a < d a1 , d ref has two solutions. Since the phase reference duty cycle d of the synchronization signal Sync a_ref is 0.1, the control monotonicity is selected, that is, the reference bypass duty cycle d ref decreases as the phase duty cycle d a increases as the control logic of the synchronization loop. Since the synchronization loop performs closed-loop phase-shifting control according to the output bypass duty cycle d a of the power loop, the speed of the synchronization loop needs to be slower than that of the power loop. It is set that the synchronization loop performs phase-shifting control every N power loop control periods, and the control period of the power loop is the same as the system resonance period. Figure 13 、 Figure 14 and Figure 15 meet the same conditions as points A, B, and C in

[0123] Furthermore, in the solution of the present invention, the initial value d of the output bypass duty cycle of the power loop initialThe description is as follows:

[0124] From Figure 13 , Figure 14 and Figure 15 it can be seen that when d a < 0, the control monotonicity of the synchronous loop and the power loop may be opposite to the design, which may cause the output of the controller not to converge. Therefore, it is necessary to design appropriate initial values of the control variables and controller parameters to ensure that the control monotonicity of the power loop and the synchronous loop always conforms to the design.

[0125] From Figure 10 it can be seen that when the output DC current I o rises to the reference current I o_ref when d a may be greater than 0 or less than 0. When the primary control period is greater than the secondary control period, the secondary synchronous signal Sync continuously leads the rectifier circuit i e , d a will continuously decrease; when the output DC current I o rises to the reference current I o_ref when d a is greater than 0, the control monotonicity of the power loop and the synchronous loop are respectively and in line with the design. When the primary control period is less than the secondary control period, the secondary synchronous signal Sync continuously lags behind the rectifier circuit i e , d a will continuously increase. When the output DC current I o rises to the reference current I o_ref when d a is less than 0. At this time, in the synchronous loop, the operating point moves from point A to the right, and the control monotonicity is in line with the design. However, the control monotonicity of the power loop is Therefore, it is necessary to design an appropriate initial value for the output bypass duty ratio d to ensure that its control monotonicity is From Figure 13 it can be seen that when the output DC current I o rises to the reference current I o_ref after that, when the initial value of the output bypass duty ratio d satisfies d initial ≥ d1, the operating point of the power loop will jump from point A to the right of point D. At this time, the control monotonicity is

[0126] The relationship between the bypass duty ratio threshold d1, the reference bypass duty ratio d ref and the reference current I o_ref is as shown in Figure 16 . In order to ensure that the output of the system quickly follows the reference value, the initial value d initial needs to be as close as possible to the reference bypass duty ratio dref Therefore, the principle for selecting the initial value can be described as:

[0127]

[0128] In this embodiment, the reference current threshold I of the system o_ref1 has a calculated value of 2.72 A.

[0129] It should be noted that when the reference current I o_ref is greater than the reference current threshold I o_ref1 and the initial value d initial is selected as the reference bypass duty cycle d ref , the system can also remain stable, but the overshoot of the startup current will be relatively large.

[0130] To verify the effectiveness of the present invention, a simulation platform is built in Plecs using a bidirectional wireless power transfer system as an example, and the system parameters are shown in Table 1.

[0131] Parameter Name Parameter Value Parameter Name Parameter Value M 72.17 μH Cp 6.83 nF Lp 514.9 μH Cs 42.92 nF Ls 81.72 μH Co 95 nF <![CDATA[R Lp > 0.98 Ω <![CDATA[R Ls > 0.11 Ω

[0132] Table 1

[0133] Set the input voltage U of the primary DC bus in to 380 V (by keeping S3 of the primary inverter off and S4 on, making it operate in a half-bridge mode, then the actual equivalent input DC voltage U in is 190 V). When the load is a resistor, set the load resistance to 17.8 Ω. When the load is a battery, the battery is equivalent to a constant voltage source in series with a resistor. The constant voltage source is set to 50.84 V and the resistor is set to 0.1 Ω. Among them, the reference current is set to I o_ref of 3 A. Case 1 is to select the initial value d initial as the reference bypass duty cycle d ref , and Case 2 is to select the initial value d initial as the bypass duty cycle threshold d1. The simulation waveforms of the 3 A constant current control for the resistive load are shown in Figure 17, the simulation waveforms of the constant voltage 60 V control are shown in Figure 18, and the simulation waveforms of the 3 A constant current control for the battery load are shown in Figure 19. Figure 20 And Figure 21 is the experimental waveform diagram for the battery load. In Figure 21 , after the system starts, the output is first controlled to a constant current output of 3.5 A. When the battery voltage rises to 52.9 V, the system switches to a constant voltage output of 52.9 V. In Figure 21 , the reference value of the output current of the system switches from 3.5 A to 2.5 A, and the output remains stable.

[0134] As can be seen from the above waveform diagram, through the closed-loop control of the power loop and the synchronization loop, the present invention realizes the control of the output voltage / current, the output power direction and the synchronization state of the bidirectional wireless charging system, and is applicable to both resistive and battery loads.

[0135] In summary, the present invention only needs to collect the output information of the secondary DC side of the bidirectional wireless charging system, without the need for an AC current sensor to collect the high-frequency phase information of the current and the real-time wireless communication between the primary and secondary sides. It can realize the zero-voltage turn-on and synchronization control of the Mos components of the secondary rectifier, reduce the system cost and improve the reliability, and solve the phase perturbation problem caused by the crystal oscillator frequency deviation of the primary and secondary controllers in the bidirectional wireless charging system.

Claims

1. A synchronous control method for a bidirectional wireless charging system, the bidirectional wireless charging system comprising a primary circuit and a secondary circuit; the primary circuit includes an inverter circuit connected to a DC voltage source, and the output end of the inverter circuit is connected with a primary series compensation capacitor and a transmitting coil; the secondary circuit includes a secondary rectifier connected to a load, and the input end of the secondary rectifier is connected with a secondary series compensation capacitor and a receiving coil; the bidirectional wireless charging system realizes the control of the output voltage or current and the output power direction of the bidirectional wireless charging system by controlling the output bypass duty ratio of the secondary rectifier and the phase duty ratio of the synchronous signal, and is characterized in that: The synchronous control method includes the following steps: Step 1: After power-on, the primary circuit inputs a small power, the secondary rectifier enters the uncontrolled rectification mode, calculates the mutual inductance of the bidirectional wireless charging system, and calculates the reference bypass duty ratio of the secondary rectifier at synchronization according to the reference value of the output voltage or current; Step 2: When the output voltage or current rises to the reference value, turn on the power loop and the synchronization loop, and assign an initial value to the output bypass duty ratio; among them, the power loop performs closed-loop control of the output bypass duty ratio to reduce the error between the output voltage or current and the reference value; the synchronization loop performs closed-loop phase-shift control of the synchronization signal to reduce the error between the output bypass duty ratio and the reference bypass duty ratio, and further controls the phase synchronization between the synchronization signal and the zero-crossing point of the input current of the secondary rectifier; In Step 1, the mutual inductance calculation is that the primary circuit transmits the primary DC bus voltage U with low-power input at this time to the secondary circuit through low-speed non-real-time wireless communication. in1 The secondary circuit calculates the mutual inductance M according to the following formula: Where: I o is the DC bus current in the secondary circuit; ω is the angular frequency of the bidirectional wireless charging system, ω = 2πf, where f is the operating frequency of the bidirectional wireless charging system; The calculation of the reference bypass duty ratio of the secondary rectifier includes that when the secondary rectifier performs constant current control, the reference bypass duty ratio d ref The calculation formula is as follows: Among them, I o_ref is the reference value of the DC bus current of the secondary circuit, U in_rated is the DC bus voltage of the primary circuit under the rated power, M is the mutual inductance, ω is the angular frequency of the bidirectional wireless charging system, d a_ref is the reference value of the duty cycle of the phase of the synchronous signal of the secondary circuit; The calculation of the reference bypass duty ratio of the secondary rectifier includes that when the secondary side performs constant voltage control, the reference bypass duty ratio d ref The calculation formula is as follows: Among them, U o_ref is the reference value of the DC bus voltage of the secondary circuit, U in_rated is the DC bus voltage of the primary circuit under the rated power, M is the mutual inductance, ω is the angular frequency of the bidirectional wireless charging system, d a_ref is the reference value of the duty cycle of the phase of the synchronous signal of the secondary circuit; R L is the equivalent resistance of the load, U o and I o are the sampled values of the output DC voltage and current respectively; The way to assign the initial value of the output bypass duty ratio is as follows: Set the reference current threshold I o_ref1 , when the reference current I o_ref is less than the reference current threshold I o_ref1 , the initial value d initial is the reference bypass duty cycle d ref ; When the reference current I o_ref is greater than the reference current threshold I o_ref1 , the initial value d initial is the bypass duty cycle threshold d1, and the calculation of the initial phase duty cycle is as follows:

2. The synchronous control method of the bidirectional wireless charging system according to claim 1, characterized in that: The reference current threshold I o_ref1 is solved by the following equation:

3. The synchronous control method of the bidirectional wireless charging system according to claim 1, wherein: The synchronous control method further includes the reverse flow of the output power. At this time, the output bypass duty ratio of the power loop remains unchanged, the synchronization loop controls the synchronization signal to gradually shift the phase by 180°, and then makes the power loop resume closed-loop control to achieve the reverse flow of power.

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