Joint dynamic ranging method based on energy centroid and full-phase FFT

Abstract

This invention discloses a joint dynamic ranging method based on energy centroid and full-phase FFT, mainly addressing the problem that existing methods cannot achieve high-precision dynamic ranging. It includes: 1) acquiring an interference signal containing distance information based on a double-sideband frequency scanning interferometric ranging system; 2) using energy centroid FFT and full-phase FFT to coarsely and finely measure the phase difference between the upper and lower sidebands respectively, resolving phase ambiguity issues, and jointly obtaining a high-precision phase difference calculation result; 3) constructing a dynamic ranging formula based on the double-sideband frequency scanning interferometric ranging system, and substituting the result into the formula to calculate the absolute distance; 4) repeatedly measuring distance using a sliding window and averaging the results to reduce random ranging errors; 5) obtaining the average value within each frequency sweep cycle to achieve dynamic tracking measurement of the target. The absolute ranging accuracy of this invention reaches the sub-micron level, realizing high-precision dynamic absolute distance measurement over long distances.

Description

Technical Field

[0001] This invention belongs to the field of laser measurement technology, and further relates to dynamic ranging methods, specifically a joint dynamic ranging method based on energy centroid and full-phase fast Fourier transform (FFT), which can be used for long-distance, high-precision dynamic absolute distance measurement. Background Technology

[0002] High-precision absolute distance measurement technology is crucial for both basic scientific research and advanced manufacturing. Particularly in the field of space science, spacecraft formation flight technology, exemplified by the European Darwin Project, the Gravitational Wave Project, and the Sound Search Project, places extremely high demands on the accuracy of on-orbit measurements. Frequency-scanning interferometric absolute ranging (FSI) technology can achieve static absolute distance measurement, possessing advantages such as simple hardware system structure, no need for guide rails, high precision, and no non-ambiguity distance limitations, making it one of the most promising and valuable absolute ranging technologies currently available. However, absolutely stationary targets do not exist in real measurement scenarios. Factors such as the mechanical vibration and noise of the target, and temperature fluctuations in the measurement environment, can cause dynamic changes in the measured optical path difference of the FSI absolute ranging system during the optical frequency scanning process. Simultaneously, FSI is extremely sensitive to minute movements of the target; amplification of motion errors by thousands of times can obliterate the ranging accuracy in an absolutely static state, resulting in significant ranging errors. Therefore, improving the inherent defect of FSI systems being extremely sensitive to changes in optical path difference and achieving high-precision dynamic absolute distance measurement is the most challenging problem in FSI absolute ranging technology.

[0003] In 2017, Di Mo published a paper titled "A Double-Side Band Frequency Scanning Interferometry Method for Long-Distance Dynamic Absolute Measurement" in the journal *Applied Physics Series B: Lasers & Optics*. This paper proposed a double-side band frequency scanning interferometry system that uses a fixed-frequency laser and a Mach-Zehnder modulator to generate two scanning signals with opposite frequencies. IQ demodulation is then used to distinguish the frequencies of the two signals, thus compensating for ranging errors caused by target motion. However, this ranging method directly uses the main spectral line in the FFT spectrum to calculate the phase difference between the upper and lower sidebands, resulting in significant spectral errors due to spectral leakage and the picket fence effect, making it impossible to obtain high-precision absolute distance measurement results.

[0004] In 2018, Keshu Zhang published a paper titled "A Double-Side Frequency Scanning Interferometry Method for Distance Measurement in Outdoor Environments" in the journal *Optical Communications*. This paper utilizes the same double-side frequency scanning interferometry system and proposes an All-phase FFT (apFFT) phase measurement method. This method effectively suppresses phase estimation errors caused by spectral leakage, achieving a standard deviation of 16.59 μm for kilometer-distance measurements under stationary conditions, thus realizing high-precision long-distance outdoor measurements. However, this method is only applicable when the target is stationary or moving at a constant speed, where the interference signal frequency remains almost constant. For more common real-world conditions such as uniformly accelerated motion and harmonic motion, the interference signal frequency changes, resulting in significant spectral broadening and rendering the ranging algorithm ineffective. Summary of the Invention

[0005] The purpose of this invention is to address the shortcomings of existing technologies by providing an absolute distance measurement formula for targets under arbitrary motion patterns, based on a double-sideband frequency scanning interferometric measurement system. Furthermore, it proposes a joint dynamic ranging method based on energy centroid FFT and full-phase FFT. When the target is in dynamic motion, the interference signal is obtained through the double-sideband frequency scanning interferometric ranging system. The phase difference between the upper and lower sidebands is coarsely and finely measured using energy centroid FFT and full-phase FFT, respectively, to construct a dynamic ranging formula. The absolute distance is calculated using the phase difference, and a sliding window is used to further improve the ranging accuracy, resulting in high-precision absolute distance measurements of the target under various motion states. This invention, by being immune to dynamic ranging errors, ultimately achieves dynamic tracking and measurement of distant moving targets.

[0006] To achieve the above objectives, the technical solution of the present invention includes the following:

[0007] (1) Using the double-sideband frequency scanning interferometric ranging system, two orthogonal electrical signals are output as the real and imaginary parts, respectively, to construct the complex form of the interferometric signal; and an interferometric signal data segment with a data length of M within one frequency sweep cycle is extracted. Let the number of spectral lines of the full-phase FFT spectrum be N. The Nth point and the M-2Nth point of the interferometric signal data segment are recorded as the starting point and ending point of the dynamic ranging. The extracted interferometric signal data segment is divided into the first data segment [N~M-2N], the second data segment [1~2N-1], and the third data segment [M-2N+2~M].

[0008] (2) Using the energy centroid FFT algorithm, perform Fast Fourier Transform (FFT) on the first data segment [N~M-2N] to obtain a coarse measurement of the phase difference between the upper and lower sidebands. and

[0009] (3) The full-phase FFT algorithm is used to obtain the precise value of the phase difference between the upper and lower sidebands. and

[0010] (4) Take and The integer part, taking and The decimal parts are added together to obtain a high-precision result. and Measured values, of which and These represent the phase difference between the upper and lower sidebands, respectively.

[0011] (5) Construct a dynamic ranging formula based on a double-sideband frequency scanning interferometric ranging system, substitute the phase difference calculation result into the dynamic ranging formula, and solve for the absolute distance S; the implementation steps are as follows:

[0012] (5.1) Construct the phase difference of the upper sideband within the measurement time T based on the expression of the interference signal. Phase difference with the lower sideband The expression is as follows:

[0013]

[0014] (5.2) The dynamic ranging formula is derived, namely the expressions for the initial optical path L and the change in optical path ΔL:

[0015]

[0016] (5.3) The high precision obtained by combining and Substituting this into the dynamic ranging formula in step (5.2), the initial optical path L is calculated, and the absolute distance S is further calculated using the following formula:

[0017]

[0018] Where n air It is the refractive index of air.

[0019] (6) Based on a sliding window, perform multiple repeated distance measurements and take the average value as the estimated absolute distance. The steps are as follows:

[0020] (6.1) Within the same scanning period, the starting point and ending point of the interference signal interception are both moved one position backward to construct a sliding window as a new interference signal interception segment. The ranging algorithm is then executed to calculate the absolute distance S.

[0021] (6.2) The sliding window is executed repeatedly to perform repeated distance measurements. The average of the obtained absolute distances S is calculated, and the average absolute distance is used as the final absolute distance measurement value.

[0022] (7) Obtain the final absolute distance measurement value within each frequency sweep cycle to realize dynamic tracking measurement of the target under test.

[0023] Compared with the prior art, the present invention has the following advantages:

[0024] First, this invention employs a double-sideband frequency scanning interferometric dynamic ranging system and proposes a joint phase difference algorithm based on energy centroid FFT and full-phase FFT, which combines the advantages of low computational load and high accuracy, achieving absolute distance measurement accuracy at the micrometer level. The further improved algorithm based on the sliding window reduces the ranging error to the sub-micrometer level, thereby improving the accuracy of absolute distance measurement.

[0025] Second, this invention derives the absolute distance measurement formula for the target under arbitrary motion laws, applies the improved energy centroid spectrum analysis to solve the phase difference of the frequency conversion signal, solves the problem of spectrum broadening caused by complex motion, and reveals the mechanism of its role in offsetting double-sideband ranging errors. This makes the proposed joint phase difference ranging algorithm applicable to dynamic ranging scenarios such as uniform acceleration motion and simple harmonic motion that are more common in real-world environments, thereby realizing the dynamic tracking and measurement of long-distance moving targets. Attached Figure Description

[0026] Figure 1 This is a schematic diagram of the overall structure of an existing double-sideband frequency scanning interferometric ranging system;

[0027] Figure 2 This is a flowchart illustrating the implementation of the method of the present invention;

[0028] Figure 3 This is a schematic diagram of an existing double-sideband ranging algorithm;

[0029] Figure 4 This is a double-sideband amplitude spectrum of simple harmonic motion with an amplitude of 100µm and a frequency of 100Hz in an embodiment of the present invention;

[0030] Figure 5 This is a simulation result of a target undergoing simple harmonic motion at a distance of 100m in an embodiment of the present invention.

[0031] Figure 6 This is a simulation result of a 100m distance measurement of a target undergoing uniform acceleration in an embodiment of the present invention;

[0032] Figure 7 This is a simulation result diagram of the dynamic tracking and measurement of simple harmonic motion in an embodiment of the present invention;

[0033] Figure 8This is a simulation result diagram of the target simple harmonic motion dynamic tracking measurement error based on the sliding window in this invention. Detailed Implementation

[0034] The embodiments and effects of the present invention will be further described in detail below with reference to the accompanying drawings:

[0035] Example 1: Refer to Appendix Figure 2 The present invention proposes a joint dynamic ranging method based on energy centroid and full-phase FFT, comprising the following steps:

[0036] Step 1. Using a double-sideband frequency scanning interferometric ranging system, output two orthogonal electrical signals, respectively as the real and imaginary parts, to construct the complex form of the interference signal; and extract an interference signal data segment of length M within one sweep cycle. Since there is a time delay between the measurement optical signal and the reference optical signal, there are discontinuities between adjacent sweep cycles. These discontinuities are detected, hence the extraction of an interference signal data segment of length M within one sweep cycle for subsequent distance measurement calculation; let N be the number of spectral lines in the full-phase FFT spectrum, and then... Points N and M-2N of the signal data segment are denoted as the starting and ending points of dynamic ranging. The intercepted interferometric signal data segment is divided into the first data segment [N~M-2N], the second data segment [1~2N-1], and the third data segment [M-2N+2~M]. The [N~M-2N] data segment is used for coarse phase difference measurement using the energy centroid FFT algorithm, the [1~2N-1] data segment is used for fine phase measurement of the starting point using the full-phase FFT algorithm, and the [M-2N+2~M] data segment is used for fine phase measurement of the ending point using the full-phase FFT algorithm.

[0037] Step 2. Using the energy centroid FFT algorithm, perform a Fast Fourier Transform (FFT) on the first data segment [N~M-2N] to obtain a coarse measurement of the phase difference between the upper and lower sidebands. and The specific steps are as follows:

[0038] An FFT is performed on the truncated [N~M-2N] interference signal data segment to obtain the double-sideband amplitude spectrum. Since the main spectral line is unknown and the double-side discrete spectrum exhibits some spectral leakage, and to reduce the computational burden of spectrum estimation, the sum of squares of the amplitude spectrum is approximated by the summation of spectral lines over a certain bandwidth. Based on this, a threshold coefficient ε is designed and defined as follows:

[0039] ε=|X| max / α

[0040] Where, |X| max It represents the maximum value of the amplitude spectrum of each sideband, where α is the bandwidth coefficient, which determines the value used to calculate the energy centroid.

[0041] The number of spectral lines is set to an empirical value of 20 in this embodiment;

[0042] (2.1) Define the threshold coefficients of the upper and lower sidebands as ε upper and ε lower :

[0043] ε upper =|X upper | max / α,

[0044] ε lower =|X lower | max / α,

[0045] Among them, |X upper | max It is the maximum value of the sideband on the amplitude spectrum, |X lower | max It is the maximum value of the lower sideband of the amplitude spectrum, and α is the bandwidth coefficient;

[0046] (2.2) Calculate the energy centroid frequencies f of the upper and lower sidebands according to the following formula. eg,upper and f eg,lower :

[0047]

[0048]

[0049] Among them, f s Let n be the sampling frequency, n be the FFT length, and |X k | represents the amplitude of the k-th spectral line;

[0050] (2.3) Obtain the phase difference between the upper and lower sidebands based on the energy centroid frequency. and

[0051]

[0052]

[0053] Step 3. Use the full-phase FFT algorithm to obtain the precise measurement value of the phase difference between the upper and lower sidebands. and The implementation steps are as follows:

[0054] (3.1) Convolve two identical N-point Hamming windows to obtain the convolution Hamming window function, and multiply it with the second data segment [1~2N-1] to achieve weighted processing of the interference signal;

[0055] (3.2) Take the Nth point of the weighted interference signal as the starting point of the preprocessing data, and shift and add the weighted interference discrete signal in steps of N data points to obtain the subsequent preprocessing data. Finally, a total of N discrete preprocessed interference signal data points are obtained.

[0056] (3.3) Perform FFT on N discrete preprocessed interference signal data points to obtain the double-sideband amplitude spectrum of the full-phase FFT;

[0057] (3.4) Take the imaginary and real parts of the amplitude of the main frequency line of the upper and lower sidebands respectively, and solve the arctangent to obtain the phase of the ranging starting point of the upper and lower sidebands. The range of values ​​is:

[0058] (3.5) Replace the second data segment [1~2N-1] in step (3.1) with the third data segment [M-2N+2~M], and use the full-phase FFT algorithm to obtain the ranging termination point phase of the upper and lower sidebands;

[0059] (3.6) Subtract the phase of the starting point of the ranging from the phase of the ranging termination point of the upper and lower sidebands to obtain the precise phase difference values ​​of the upper and lower sidebands, respectively. and

[0060] Step 4. Take and The integer part, taking and The decimal parts are added together to obtain a high-precision result. and Measured values, of which and These represent the phase difference between the upper and lower sidebands, respectively; specifically as follows:

[0061] (4.1) Coarsely obtained using the energy centroid FFT algorithm and calculate and Phase difference precisely measured using the full-phase FFT algorithm and calculate and

[0062] (4.2) Due to detailed measurement and The value range is [-2π, 2π], and there are two detailed measurements that are complementary with respect to 2π. Taking the coarse measurement... and For the fractional part, the fine measurement result with the smallest error is taken as the true fine measurement result, thereby solving the "phase ambiguity" problem between the phase difference calculated by full-phase FFT and the true phase difference;

[0063] (4.3) Take the rough measurement results and The integer part will be obtained through detailed measurement. and As the fractional part, combined, we obtain high precision. and Measured value.

[0064] Step 5. Construct a dynamic ranging formula based on a double-sideband frequency scanning interferometric ranging system, substitute the phase difference calculation result into the dynamic ranging formula, and solve for the absolute distance S; the implementation steps are as follows:

[0065] (5.1) Construct the phase difference of the upper sideband within the measurement time T based on the expression of the interference signal. Phase difference with the lower sideband The expression is as follows:

[0066]

[0067] (5.2) The dynamic ranging formula is derived, namely the expressions for the initial optical path L and the change in optical path ΔL:

[0068]

[0069] (5.3) The high precision obtained by combining and Substituting this into the dynamic ranging formula in step (5.2), the initial optical path L is calculated, and the absolute distance S is further calculated using the following formula:

[0070]

[0071] Where n air It is the refractive index of air.

[0072] Step 6. Based on a sliding window, perform multiple repeated distance measurements and take the average value as the estimate of the absolute distance. The steps are as follows:

[0073] (6.1) Within the same scanning period, the starting point and ending point of the interference signal interception are both moved one position backward to construct a sliding window as a new interference signal interception segment. The ranging algorithm is then executed to calculate the absolute distance S.

[0074] (6.2) The sliding window is executed repeatedly to perform repeated distance measurements. The average of the obtained absolute distance S is calculated and the average absolute distance is used as the final absolute distance measurement value to further reduce the impact of random errors in dynamic distance measurement.

[0075] Step 7. Obtain the final absolute distance measurement value within each frequency sweep cycle to achieve dynamic tracking measurement of the target under test, as detailed below:

[0076] (7.1) After obtaining the absolute distance measurement value of a scanning cycle segment, extract the interference signal data segment of the next scanning cycle based on the discontinuity between two adjacent scanning cycles obtained by detection;

[0077] (7.2) By repeating steps (1)-(6), the absolute distance measurement value of the interference signal data segment of the next scanning cycle is obtained;

[0078] (7.3) Set the data update time interval to one scan time, and repeat steps (7.1)-(7.2) to obtain the absolute distance measurement values ​​of all frequency sweep period interference signal data segments, so as to realize the dynamic tracking measurement of the target under test.

[0079] Example 2: Refer to Appendix Figure 1 The overall implementation steps of the ranging method proposed in this embodiment are the same as those in Embodiment 1. The specific implementation is based on a double-sideband frequency scanning interferometric ranging system for high-precision absolute distance measurement and tracking of dynamic targets. A specific example is given below for further detailed description of the double-sideband frequency scanning interferometric ranging system involved in this invention:

[0080] The double-sideband frequency scanning interferometric ranging system used in this example includes a 1550nm narrow-linewidth laser (LASER), a swept-frequency signal source, a driver, a Mach-Zehnder intensity modulator (MZM), an optical fiber beam splitter, an optical power amplifier (EDFA), a circulator, a transceiver optical antenna, a corner cube prism, a 90° optical mixer, a balanced detector (BPD), a data acquisition card, and a programmable gate array (FPGA).

[0081] A 1550nm narrow linewidth laser (LASER) is connected to a Mach-Zehnder intensity modulator (MZM) via optical fiber, and the output optical carrier is used for sideband modulation.

[0082] A frequency sweep signal source, connected to a driver, outputs a linear frequency sweep electrical signal, which is used for double-sideband modulation to generate a modulated optical signal with linear frequency variation;

[0083] The driver output is connected to a Mach-Zehnder intensity modulator (MZM) to change the amplitude of the sweep frequency signal, thereby changing the double-sideband modulation depth.

[0084] The Mach-Zehnder intensity modulator (MZM) outputs a fiber optic beam splitter connected via optical fiber to generate up-and-down sweeping positive and negative first-order double-sideband modulated signals.

[0085] The fiber optic beam splitter has two outputs for laser beam splitting. One optical output is connected to the optical power amplifier (EDFA) as the measurement optical path, and the other optical output is directly input to the 90° optical mixer as the reference optical path.

[0086] The optical power amplifier EDFA outputs through an optical fiber connected to circulator interface 1 for measuring the optical power amplification of the optical path.

[0087] The circulator has an interface 2 that connects to an optical antenna and an output interface 3 that connects to a 90° optical mixer via optical fiber. The unique structure of the interface is used for the integrated transceiver optical path design.

[0088] The transceiver optical antenna has one end connected to a circulator via an optical fiber, and the other end is in free space, which is used to achieve coupling between the optical fiber and free space.

[0089] A corner prism, rigidly mounted on the target under test, moves with the target under test and is used to reflect the beam emitted from the optical antenna back to the optical antenna in parallel.

[0090] The 90° optical mixer outputs four optical interference signals, each with a phase difference of 90°. Among them, two optical interference signals with a phase difference of 180° are connected to the balanced detector BPD through polarization-maintaining fiber, which are used to measure the coherence between the light and the reference light and to form orthogonal interference signals.

[0091] The balanced detector (BPD) outputs an electrical signal that is connected to the data acquisition card. It is used to convert optical signals into electrical signals and eliminate the DC component in the interference signal.

[0092] The data acquisition card outputs digital signals to the programmable gate array (FPGA) for high-speed acquisition of analog signals, facilitating subsequent signal processing.

[0093] The programmable gate array (FPGA) is connected to the output signal of the data acquisition card to perform data processing calculations and determine the absolute distance to the target.

[0094] Example 3: Refer to Figure 1 The overall implementation steps of the ranging method proposed in this embodiment are the same as those in Embodiment 1. The specific implementation steps of the two-sided frequency scanning interferometric ranging system that outputs two orthogonal electrical signals are described in detail below:

[0095] (1a) The fixed-frequency laser emitted by the laser is subjected to double-sideband carrier suppression modulation by a Mach-Zehnder intensity modulator (MZM). In the nth scanning cycle, the optical modulation signal emitted by the double-sideband frequency scanning interferometric ranging system is used as the reference optical signal and is represented as follows:

[0096] S t (t)=S t,upper (t)+S t,lower(t)

[0097] =exp[j2π(f c +f0)t+jπKt 2 ]+exp[j2π(f c -f0)t-jπiKt 2 ]

[0098] Among them, S t,upper (t) represents the upper sideband of the reference optical signal, S t,lower (t) represents the lower sideband of the reference optical signal, t∈[0,T], where T is the time to complete one frequency scan, f c f0 is the frequency of the narrow linewidth laser, f0 is the initial frequency of the radio frequency signal source, and K is the frequency scanning rate.

[0099] (1b) Allow the target to move arbitrarily, and at any time transmit the measurement light signal S. r (t) represents the reference optical signal S. t (t) Time delay τ based on the relative position of the measured target:

[0100]

[0101] Where c is the speed of light, L is the initial optical path length of the target at the interception time, and ΔL(t) is the relative optical path length of the target at time t. The optical path propagates bidirectionally, transmitting the measured optical signal S. r (t) represents the following:

[0102]

[0103] Among them, S r,upper (t) represents the upper sideband of the measured optical signal, S r,lower (t) represents the lower sideband of the measured optical signal;

[0104] (1c) Let the reference light and the measurement light interfere in a 90° optical mixer, and realize the conversion of optical signal to electrical signal through a balanced detector BPD. The obtained interference signal I(t) is:

[0105]

[0106] (1d) The interference signal I(t) obtained by the balanced detector BPD is sampled at high speed through the data acquisition card to obtain two orthogonal electrical signals, which are used as the real part and the imaginary part respectively. The complex form of the interference signal is constructed to facilitate the acquisition of the Fourier transform double-sideband spectrum.

[0107] The effects of this invention can be further illustrated by the following simulation results.

[0108] I. Simulation Conditions

[0109] Using MATLAB simulation software, the speed of light is set to 3 × 10⁻⁶. 8 m / s, laser wavelength of 1550nm, initial frequency of sweep signal source of 5GHz, sweep period of 1ms, sweep bandwidth of 5GHz, full phase FFT calculation points of 256, initial distance of 100m, sampling frequency of 100MHz, and sampling resolution of 10bit.

[0110] II. Simulation Content and Results

[0111] Simulation 1: Under the above simulation conditions, with the target undergoing simple harmonic motion at an amplitude of 100µm and a frequency of 100Hz, the FFT double-sideband amplitude spectrum of the interference signal is simulated, as follows: Figure 4 As shown, because the instantaneous frequency of the interference signal is constantly changing, there is a large spectral broadening. According to... Figure 3 Existing double-sideband ranging algorithms use the extreme value spectral line as the main spectral line, resulting in a large error compared to the true ranging spectral line. Therefore, existing ranging algorithms cannot perform dynamic ranging.

[0112] The energy centroid method was used to obtain more accurate frequency positions, and its application to frequency conversion signal spectrum analysis was extended, although... Figure 4 Errors still exist between the mid-energy barycentric spectral line and the true spectral line of the ranging measurement, but the phase errors of the frequency variation amplification terms in the upper and lower sidebands are basically the same, and since the initial optical path L is mainly calculated by... Since the phase measurement error is canceled out during the distance calculation process, the distance measurement method of this invention can theoretically realize dynamic distance measurement.

[0113] Simulation 2: The combined ranging algorithm of this invention is used to simulate the simple harmonic motion of a target at a distance of 100m. The ranging accuracy at a distance of 100m is as follows: Figure 5 As shown, for simple harmonic motion with a frequency of 200Hz and an amplitude of less than 200µm, the ranging accuracy can reach the micrometer level.

[0114] Simulation 3: The joint ranging algorithm of this invention is used to simulate the uniformly accelerated motion of a target at a distance of 100m. The ranging accuracy at a distance of 100m is as follows: Figure 6 As shown, the horizontal axis represents the set acceleration of the target being measured, and the vertical axis represents the ranging error. For a speed of 10 m / s², the vertical axis represents the ranging error. 2 For the following uniformly accelerated motion, the ranging accuracy can also reach the micrometer level.

[0115] Figure 5 and Figure 6 Dynamic simulation examples show that the method of the present invention can truly realize dynamic absolute distance measurement of the target, with ranging accuracy reaching the micrometer level.

[0116] Simulation 4: Simulation of tracking and measuring the simple harmonic motion of a 100m target using the joint ranging algorithm of this invention, such as... Figure 7 As shown, the data update time is 1ms, which can reproduce the motion trajectory of the target being tested.

[0117] Simulation 5: Using the improved ranging algorithm based on the sliding window of this invention, the simple harmonic motion of the target at 100m was simulated. Compared with the single ranging error results, the sliding window further reduced the dynamic tracking measurement error based on the joint phase difference algorithm, and the ranging accuracy reached the sub-micron level.

[0118] The simulation results above show that the method of the present invention can improve the ranging accuracy of traditional methods to the sub-micron level, and truly realize long-distance high-precision dynamic absolute distance measurement.

[0119] The above simulation analysis proves the correctness and effectiveness of the method proposed in this invention.

[0120] The parts of this invention not described in detail are common knowledge to those skilled in the art.

[0121] The above description is merely a preferred embodiment of the present invention and is not intended to limit the present invention. Obviously, those skilled in the art, after understanding the content and principle of the present invention, may make various modifications and changes in form and detail without departing from the principle and structure of the present invention. However, these modifications and changes based on the concept of the present invention are still within the scope of protection of the claims of the present invention.

Claims

1. A joint dynamic ranging method based on energy centroid and full-phase FFT, characterized in that, When the target under test is in dynamic motion, the interference signal is obtained through a double-sideband frequency scanning interferometric ranging system. The phase difference between the upper and lower sidebands is coarsely measured and finely measured using energy centroid fast Fourier transform (FFT) and full-phase fast Fourier transform (FFT), respectively. A dynamic ranging formula is constructed, and the absolute distance is calculated using the phase difference. A sliding window is used to further improve the ranging accuracy, ultimately achieving dynamic tracking and measurement of the target. The implementation steps include the following: (1) Using the double-sideband frequency scanning interferometric ranging system, two orthogonal electrical signals are output as the real and imaginary parts, respectively, to construct the complex form of the interferometric signal; and an interferometric signal data segment with a data length of M within one frequency sweep cycle is extracted. Let the number of spectral lines of the full-phase FFT spectrum be N. The Nth point and the M-2Nth point of the interferometric signal data segment are recorded as the starting point and ending point of the dynamic ranging. The extracted interferometric signal data segment is divided into the first data segment [N~M-2N], the second data segment [1~2N-1], and the third data segment [M-2N+2~M]. (2) Using the energy centroid FFT algorithm, perform Fast Fourier Transform (FFT) on the first data segment [N~M-2N] to obtain a coarse measurement of the phase difference between the upper and lower sidebands. and (3) The full-phase FFT algorithm is used to obtain the precise value of the phase difference between the upper and lower sidebands. and (4) Take and The integer part, taking and The decimal parts are added together to obtain a high-precision result. and Measured values, of which and These represent the phase difference between the upper and lower sidebands, respectively. (5) Construct a dynamic ranging formula based on a double-sideband frequency scanning interferometric ranging system, substitute the phase difference calculation results into the dynamic ranging formula, and solve for the absolute distance S; The implementation steps are as follows: (5.1) Construct the phase difference of the upper sideband within the measurement time T based on the expression of the interference signal. Phase difference with the lower sideband The expression is as follows: Where c is the speed of light, f c f0 is the frequency of the narrow linewidth laser, f0 is the initial frequency of the radio frequency signal source, and K is the frequency scanning rate. (5.2) The dynamic ranging formula is derived, namely the expressions for the initial optical path L and the change in optical path ΔL: (5.3) The high precision obtained by combining and Substituting this into the dynamic ranging formula in step (5.2), the initial optical path L is calculated, and the absolute distance S is further calculated using the following formula: Where n air It is the refractive index of air. (6) Based on a sliding window, perform multiple repeated distance measurements and take the average value as the estimated absolute distance. The steps are as follows: (6.1) Within the same scanning period, the starting point and ending point of the interference signal interception are both moved one position backward to construct a sliding window as a new interference signal interception segment. The ranging algorithm is then executed to calculate the absolute distance S. (6.2) The sliding window is executed repeatedly to perform repeated distance measurements. The average of the obtained absolute distances S is calculated, and the average absolute distance is used as the final absolute distance measurement value. (7) Obtain the final absolute distance measurement value within each frequency sweep cycle to realize dynamic tracking measurement of the target under test.

2. The method according to claim 1, characterized in that: In step (1), two orthogonal electrical signals are output using a double-sideband frequency scanning interferometric ranging system. The specific implementation steps are as follows: (1a) The fixed-frequency laser emitted by the laser is subjected to double-sideband carrier suppression modulation by a Mach-Zehnder intensity modulator (MZM). In the nth scanning cycle, the optical modulation signal emitted by the double-sideband frequency scanning interferometric ranging system is used as the reference optical signal and is represented as follows: S t (t)=S t,upper (t)+S t,lower (t) =exp[j2π(f c +f0)t+jπKt 2 ]+exp[j2π(f c -f0)t-jπKt 2 ] Among them, S t,upper (t) represents the upper sideband of the reference optical signal, S t,lower (t) represents the lower sideband of the reference optical signal, t∈[0,T], where T is the time to complete one frequency scan; (1b) Allow the target to move arbitrarily, and at any time transmit the measurement light signal S. r (t) represents the reference optical signal S. t (t) Time delay τ based on the relative position of the measured target: Where L is the initial optical path of the target at the interception time, ΔL(t) is the relative optical path of the target at time t, and the optical path propagates bidirectionally, transmitting the measurement optical signal S r (t) represents the following: Among them, S r,upper (t) represents the upper sideband of the measured optical signal, S r,lower (t) represents the lower sideband of the measured optical signal; (1c) Let the reference light and the measurement light interfere in a 90° optical mixer, and convert the optical signal into an electrical signal through a balanced detector BPD to obtain the interference signal I(t): (1d) The interference signal I(t) obtained by the balanced detector BPD is sampled at high speed through the data acquisition card to obtain two orthogonal electrical signals.

3. The method according to claim 1, characterized in that: The coarse measurement of the upper sideband phase difference in step (2) Coarse measurement of phase difference with lower sideband Specifically, it is obtained through the following steps: (2.1) Define the threshold coefficients of the upper and lower sidebands as ε upper and ε lower : e upper =|X upper | max / a, e lower =|X lower | max / a, Among them, |X upper | max It is the maximum value of the sideband on the amplitude spectrum, |X lower | max It is the maximum value of the lower sideband of the amplitude spectrum, and α is the bandwidth coefficient; (2.2) Calculate the energy centroid frequencies f of the upper and lower sidebands according to the following formula. eg,upper and f eg,lower : Among them, f s Let n be the sampling frequency, n be the FFT length, and |X k | represents the amplitude of the k-th spectral line; (2.3) Obtain the phase difference between the upper and lower sidebands based on the energy centroid frequency. and 4. The method according to claim 1, characterized in that, Step (3) Use the full-phase FFT algorithm to obtain the precise measurement value of the upper sideband phase difference. Precise measurement of phase difference between the lower sideband and the lower sideband The implementation is as follows: (3.1) Convolve two identical N-point Hamming windows to obtain the convolution Hamming window function, and multiply it with the second data segment [1~2N-1] to achieve weighted processing of the interference signal; (3.2) Take the Nth point of the weighted interference signal as the starting point of the preprocessing data, and shift and add the weighted interference discrete signal in steps of N data points to obtain the subsequent preprocessing data. Finally, a total of N discrete preprocessed interference signal data points are obtained. (3.3) Perform FFT on N discrete preprocessed interference signal data points to obtain the double-sideband amplitude spectrum of the full-phase FFT; (3.4) Take the imaginary and real parts of the amplitude of the main frequency line of the upper and lower sidebands respectively, and solve the arctangent to obtain the phase of the ranging starting point of the upper and lower sidebands. The range of values ​​is: (3.5) Replace the second data segment [1~2N-1] in step (3.1) with the third data segment [M-2N+2~M], and use the full-phase FFT algorithm to obtain the ranging termination point phase of the upper and lower sidebands; (3.6) Subtract the phase of the starting point of the ranging from the phase of the ranging termination point of the upper and lower sidebands to obtain the precise phase difference values ​​of the upper and lower sidebands, respectively. and 5. The method according to claim 1, characterized in that: The dynamic tracking and measurement of the target object in step (7) is achieved as follows: (7.1) After obtaining the absolute distance measurement value of a scanning period segment, extract the interference signal data segment of the next scanning period based on the discontinuity between two adjacent scanning periods obtained by detection; (7.2) By repeating steps (1)-(6), the absolute distance measurement value of the interference signal data segment of the next scanning cycle is obtained; (7.3) Set the data update time interval to one scan time, and repeat steps (7.1)-(7.2) to obtain the absolute distance measurement values ​​of all frequency sweep period interference signal data segments, so as to realize the dynamic tracking measurement of the target under test.

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