A discrete vector proportional resonance control method

Abstract

The present invention discloses a discrete vector proportional resonant control method, which belongs to the field of power electronic control technology. It comprises a negative frequency pole resonance link, a positive frequency pole resonance link, and a zero point link. Aiming at the challenges of low switching frequency and control frequency, high regulation delay, limited stability and dynamic performance of high-power converters, the present invention proposes a discrete vector proportional resonant control method, which has high accuracy of discrete scheme and good consistency with the original continuous domain controller. And this discrete scheme is simpler to calculate than the discrete scheme of traditional bilinear transformation. Therefore, through the discrete vector proportional resonant control of the present invention, the vector proportional resonant controller of large-capacity converter under low carrier ratio conditions can be realized with higher accuracy and lower calculation amount when discretized.

Description

Technical Field

[0001] The invention belongs to the technical field of power electronic control, and in particular relates to a vector proportional resonance control method. Background Art

[0002] As an energy conversion device, three-phase large-capacity converters are increasingly widely used in industrial fields such as electrified transportation and ship power systems. Such converters usually operate under low carrier ratio conditions, with significant control and modulation delays and insufficient control loop stability margin, which affects their dynamic performance.

[0003] The three-phase converter can use a vector proportional resonant controller to control the three-phase current in a stationary coordinate system. However, due to the switching frequency of high-power devices, which is usually less than one kilohertz, the system control delay can reach milliseconds, and its phase margin and corresponding dynamic performance are seriously insufficient.

[0004] Therefore, it is necessary to design a new solution to improve the system stability and dynamic performance under low carrier ratio conditions where large-capacity converters are located. Summary of the invention

[0005] In order to improve the dynamic performance of large-capacity converters, the present invention proposes a discrete vector proportional resonant control method, which avoids the problem of algebraic loops or calculation complexity due to increased order compared to traditional discrete vector proportional resonant controllers.

[0006] To achieve the above purpose, the technical solution adopted by the present invention is as follows:

[0007] A discrete vector proportional resonance control method comprises the following steps:

[0008] 1) Sample the current of each phase in the controlled converter, and obtain the current i in the stationary coordinate system through abc / αβ coordinate transformation α and i β , and define the current sampling value i αβ =i α +ji β , where i α with i β are the current values ​​of the α-axis and β-axis in the stationary coordinate system, i αβ is a complex vector, j is an imaginary unit;

[0009] 2) Set the current reference value i αβ_R Subtract the current sampling value i αβ , and get the current error i αβ_E ;

[0010] 3) Current error i αβ_E After the negative frequency pole resonance link and the positive frequency pole resonance link, we can get mαβ_n and m αβ_p ;m αβ_n and m αβ_p Add together to get the total output m of the two resonant links αβ_pn :

[0011] 4) The total output of the two resonant links is m αβ_pn The total output m of the control loop is obtained through the zero point link αβ :

[0012] 5) Total output of control loop m αβ After αβ / abc coordinate transformation, the three-phase modulation wave m is obtained. a 、m b 、m c , and compared with the carrier in the modulation and drive module to generate a drive signal to drive the controlled converter to achieve power conversion.

[0013] Furthermore, the current error i αβ_E The output m of the negative frequency pole resonance link is obtained through the formula I corresponding to the negative frequency pole resonance link. αβ_n :

[0014] m αβ_n =i αβ_E ·T s ·e -jθ ' / [1-z -1 e^(-jω0T s )] Formula I

[0015] Among them, T s is the control period, θ' is the improved phase correction angle, z is the operator of discrete transformation, e is the base of the natural logarithm function, and ω0 is the fundamental angular frequency;

[0016] Furthermore, the current error i αβ_E The output m of the positive frequency pole resonance link is obtained through equation II corresponding to the positive frequency pole resonance link. αβ_p :

[0017] m αβ_p =i αβ_E ·T s ·e jθ ' / [1-z -1 e^(jω0T s Formula II

[0018] Furthermore, the total output m of the two resonant links is αβ_pn The total output m of the control loop is obtained by equation III corresponding to the zero point link: αβ :

[0019] m αβ =mαβ_pn ·K·R / L·[1-z -1 e^(-T s ·R / L)] / [1-e^(-T s ·R / L)] Formula III

[0020] Wherein, K is the proportional coefficient, L and R are the inductance value and equivalent series resistance value of the grid-connected filter inductor of the converter respectively;

[0021] Furthermore, the improved phase correction angle θ' is the original continuous domain phase correction angle θ plus the phase compensation angle θ c , corresponding to formula IV:

[0022] θ'=θ+θ c Formula IV

[0023] Where, the phase compensation angle θ c Usually 0.5ω0T s .

[0024] The present invention has the following beneficial effects:

[0025] Under the low carrier ratio working condition corresponding to the large-capacity converter, for the control of the three-phase converter in the αβ stationary coordinate system, for the traditional discrete vector proportional resonant controller based on the bilinear transformation method, the equation V corresponding to the positive frequency pole resonance link is taken as an example:

[0026] m αβ_p =i αβ_E ·e jθ / [k w (1-z -1 )-jω0(1+z -1 Formula V

[0027] The coefficient of dispersion k w =ω0 / tan(ω0T s / 2). In order to avoid algebraic loops during calculation, this scheme usually requires multiplying the numerator and denominator by k. w (1-z -1 )+jω0(1+z -1 ), let the denominator be real, that is, the positive frequency pole resonance link after eliminating the algebraic loop corresponds to formula VI:

[0028] m αβ_p =i αβ_E ·e jθ (k w (1-z -1 )+jω0(1+z -1 )) / [k w 2 (1-z -1 )2 +ω0 2 (1+z -1 ) 2 Formula VI

[0029] It faces new problems of higher order and more complicated calculation. However, the present invention improves the discrete scheme, has high discrete accuracy, avoids the problem of algebraic loop or the problem of complicated calculation due to increased order, and achieves beneficial technical effects. BRIEF DESCRIPTION OF THE DRAWINGS

[0030] Figure 1 is a schematic diagram of a converter;

[0031] Figure 2 It is the overall control block diagram;

[0032] Figure 3 This is a block diagram of the discrete vector proportional resonant controller proposed by the present invention;

[0033] Figure 4 It is the block diagram of the vector proportional resonant controller under the traditional bilinear transformation discrete;

[0034] Figure 5 This is the block diagram of the traditional continuous domain vector proportional resonant controller;

[0035] Figure 6 The Bode diagram comparison between the discrete domain controller proposed in the present invention and the traditional continuous domain controller. DETAILED DESCRIPTION

[0036] The following takes the current loop control of a converter with a three-phase bridge inverter topology as an example, and describes in detail the purpose, scheme and advantages of the present invention in combination with the accompanying drawings and embodiments.

[0037] Figure 1 The schematic diagram of the converter with three-phase bridge inverter topology is shown in Figure 1. Figure 2 The overall control block diagram is shown in Figure 1, where the three-phase current sampling obtains the AC side current i a 、i b 、i c , and after abc / αβ coordinate transformation, the current i in the stationary coordinate system is obtained α 、i β , as the input of the control loop. Next, the control loop outputs the modulation wave m in the stationary coordinate system α 、m β , after αβ / abc coordinate transformation, we get the three-phase modulation wave m a 、m b 、m c , and compared with the carrier in the modulation and drive module to generate a drive signal to drive the converter to achieve power conversion.

[0038] Figure 3This is a specific implementation block diagram of the control loop, that is, the discrete vector proportional resonance control method proposed in the present invention, including a negative frequency pole resonance link, a positive frequency pole resonance link, and a zero point link. Here, the expression of complex vectors and complex transfer functions is adopted, and the current sampling value i in the stationary coordinate system is taken as αβ For example, the complex vector represents i α +ji β , where j is the imaginary unit, i α with i β represent the current values ​​of the α-axis and β-axis respectively. The definitions of other complex vectors with subscripts αβ are the same.

[0039] The control loop obtains the corresponding i from the sample of the controlled object αβ , and outputs the modulation wave m αβ To control the controlled object. The steps of the vector proportional resonance control method corresponding to the control loop are as follows:

[0040] 1) In the stationary coordinate system, the current reference value i αβ_R Subtract the current sampling value i αβ , and get the current error i αβ_E .

[0041] 2) Current error i αβ_E After the negative frequency pole resonance link and the positive frequency pole resonance link, we can get m αβ_n and m αβ_p ;m αβ_n and m αβ_p Add together to get the total output m of the two resonant links αβ_pn :

[0042] 3) The total output of the two resonant links is m αβ_pn The total output m of the control loop is obtained through the zero point link αβ :

[0043] 4) Total output of control loop m αβ After αβ / abc coordinate transformation, the three-phase modulation wave m is obtained. a 、m b 、m c , and compared with the carrier in the modulation and drive module to generate a drive signal to drive the controlled converter to achieve power conversion.

[0044] In a specific implementation of the present invention, see Figure 3 , the current error i αβ_E The output m of the negative frequency pole resonance link is obtained through the formula I corresponding to the negative frequency pole resonance link. αβ_n :

[0045] m αβ_n =i αβ_E ·T s·e -jθ ' / [1-z -1 e^(-jω0T s )] Formula I

[0046] Among them, T s is the control period, θ' is the improved phase correction angle, z is the operator of discrete transformation, e is the base of the natural logarithm function, and ω0 is the fundamental angular frequency.

[0047] The current error i αβ_E The output m of the positive frequency pole resonance link is obtained through equation II corresponding to the positive frequency pole resonance link. αβ_p :

[0048] m αβ_p =i αβ_E ·T s ·e jθ ' / [1-z -1 e^(jω0T s Formula II

[0049] The total output m of the two resonant links is αβ_pn The total output m of the control loop is obtained by equation III corresponding to the zero point link: αβ :

[0050] m αβ =m αβ_pn ·K·R / L·[1-z -1 e^(-T s ·R / L)] / [1-e^(-T s ·R / L)] Formula III

[0051] Wherein, K is the proportional coefficient, L and R are the inductance value and equivalent series resistance value of the grid-connected filter inductor of the converter, respectively.

[0052] In this embodiment, the improved phase correction angle θ' of the proposed discrete solution is the original continuous domain phase correction angle θ plus the phase compensation angle θ c , corresponding to formula IV:

[0053] θ'=θ+θ c Formula IV

[0054] Where, the phase compensation angle θ c Usually 0.5ω0T s .

[0055] Figure 4 It is the block diagram of the vector proportional resonant controller under the traditional bilinear transformation discrete; Figure 5 This is the block diagram of the traditional continuous domain vector proportional resonant controller; Figure 6The Bode plots of the proposed discrete domain controller and the traditional continuous domain controller are compared. These three plots are used for analysis in application examples.

[0056] An application example of the present invention is given below.

[0057] for Figure 1 The three-phase power conversion circuit shown in the figure has a common control scheme: three-phase current sampling to obtain the AC side current i a 、i b 、i c , and after abc / αβ coordinate transformation, the current i in the stationary coordinate system is obtained α 、i β , as the input of the control loop. Here, the specific implementation process of the control loop is the same as the above description. Compared with the discrete vector proportional resonant controller proposed in the present invention, the traditional discrete vector proportional resonant controller based on the bilinear transformation method is Figure 4 As shown, take the formula V corresponding to the positive frequency pole resonance link as an example:

[0058] m αβ_p =i αβ_E ·e jθ / [k w (1-z -1 )-jω0(1+z -1 Formula V

[0059] Among them, the dispersion coefficient k w =ω0 / tan(ω0T s / 2). In this scheme, in order to avoid algebraic loops during calculation, it is usually necessary to multiply the numerator and denominator by k. w (1-z -1 )+jω0(1+z -1 ), let the denominator be real, that is, the positive frequency pole resonance link after eliminating the algebraic loop corresponds to formula VI:

[0060] m αβ_p =i αβ_E ·e jθ (k w (1-z -1 )+jω0(1+z -1 )) / [k w 2 (1-z -1 ) 2 +ω0 2 (1+z -1 ) 2 Formula VI

[0061] As can be seen from the above formula, it faces new problems of higher order and more complicated calculation. However, the present invention improves the discrete solution, has high discrete accuracy, avoids the problem of algebraic loop or the problem of complicated calculation due to increased order, and achieves beneficial technical effects.

[0062] Next, a frequency domain comparison analysis is performed between the conventional solution and the solution of the present invention. The parameters are set as follows: the frequency-carrier ratio is 9, the fundamental frequency ω0 is 314 rad / s, and the corresponding control period T s is 1 / 900s. The proportional coefficient K is 0.5ω0L, and the phase correction angle θ is 30°. The phase compensation angle θ of the proposed discrete scheme is c 0.5ω0T s That is, 10°, the improved phase correction angle θ' is θ+θ c That is 40°.

[0063] like Figure 6 As shown, Figure 3 The Bode diagram of the proposed discrete domain controller system is shown in Figure 5 The Bode diagrams of the system under the original traditional continuous domain controller shown in the figure basically overlap. Within the crossover frequency, the system amplitude-frequency characteristic error is less than 0.1dB, and the phase characteristic error is less than 1°, which shows that the discrete scheme has high accuracy and good consistency with the original continuous domain controller. In addition, as analyzed above, the discrete scheme is simpler to calculate than the discrete scheme of the traditional bilinear transformation. Therefore, through the discrete vector proportional resonant control of the present invention, the vector proportional resonant controller of the large-capacity converter under the low carrier ratio condition can achieve higher accuracy and lower calculation amount when discretely implemented, and achieves beneficial technical effects.

[0064] The present invention is not limited to the above specific embodiments. Those skilled in the art can adopt various other embodiments based on the contents disclosed in the present invention, such as Figure 1 The two-level converter topology in the embodiment is replaced with a three-level topology, etc. Therefore, the claims are intended to cover all variations that are within the true concept and scope of the invention.

Claims

1. A discrete vector proportional resonance control method, characterized in that: The following steps are involved: 1) Sample the current of each phase in the controlled converter, and obtain the current i in the stationary coordinate system through abc / αβ coordinate transformation α and i β , and define the current sampling value i αβ =i α +ji β , where i α with i β are the current values ​​of the α-axis and β-axis in the stationary coordinate system, i αβ is a complex vector, j is an imaginary unit; 2) Set the current reference value i αβ_R Subtract the current sampling value i αβ , and get the current error i αβ_E ; 3) Current error i αβ_E After the negative frequency pole resonance link and the positive frequency pole resonance link, we can get m αβ_n and m αβ_p ;m αβ_n and m αβ_p Add together to get the total output m of the two resonant links αβ_pn : The current error i αβ_E The output m of the negative frequency pole resonance link is obtained through the formula I corresponding to the negative frequency pole resonance link. αβ_n : m αβ_n = i αβ_E ·T s ·e -jθ ' / [1 - z -1 e^(-jω0T s )] Equation I Among them, T s is the control period, θ' is the improved phase correction angle, z is the operator of discrete transformation, e is the base of the natural logarithm function, and ω0 is the fundamental angular frequency; The current error i αβ_E The output m of the positive frequency pole resonance link is obtained through equation II corresponding to the positive frequency pole resonance link. αβ_p : m αβ_p = i αβ_E ·T s ·e jθ ' / [1 - z -1 e^(jω0T s )] Equation II Among them, T s is the control period, θ' is the improved phase correction angle, z is the operator of discrete transformation, e is the base of the natural logarithm function, and ω0 is the fundamental angular frequency; The improved phase correction angle θ' is the sum of the phase correction angle θ and the phase compensation angle θ of the original continuous domain. c sum; 4) The total output of the two resonant links is m αβ_pn The total output m of the control loop is obtained through the zero point link αβ : 5) Total output of control loop m αβ After αβ / abc coordinate transformation, the three-phase modulation wave m is obtained. a 、m b 、m c , and compared with the carrier in the modulation and drive module to generate a drive signal to drive the controlled converter to achieve power conversion.

2. A discrete vector proportional resonance control method according to claim 1, characterized in that: The total output m of the two resonant links is αβ_pn The total output m of the control loop is obtained by equation III corresponding to the zero point link: αβ : m αβ = m αβ_pn ·K·R / L·[1 - z -1 e^(-T s ·R / L)] / [1 - e^(-T s ·R / L)] Equation III Among them, K is the proportional coefficient, L and R are the inductance value and equivalent series resistance value of the converter filter inductor respectively, T s is the control period, z is the operator of discrete transformation, and e is the base of the natural logarithm function.

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