Reliable communication method for weak signals of Ku transponders of broadcast satellites based on the transform domain

Abstract

The present invention discloses a reliable communication method for weak signals of a Ku transponder of a broadcast satellite based on the transform domain, belonging to the field of satellite communication. Using transform domain communication as the weak signal communication technology system, research is carried out on the weak signal communication technology that realizes weak signals with a power spectral density lower than the noise by 15 dB or more under a low code rate by using a transparent transponder in the Ku band of a broadcast satellite, and methods for spectrum sensing and rapid capture and tracking of weak signals are designed; research is carried out on the synchronization and demodulation technology of multi-carrier aggregation deep spread spectrum signals with an extremely low spectral density to achieve optimal bit error performance.

Description

Technical Field

[0001] The present invention belongs to the field of satellite communication, and particularly relates to a reliable communication method for weak signals of a Ku transponder of a broadcast satellite based on the transform domain. Background Art

[0002] The satellite transmission technology of radio and television mainly consists of three major parts: an uplink transmitting station, an on-board transponder, and a ground receiver. It contains devices for sending and receiving signals like general transmission technologies, and on this basis, a satellite as a relay system is added.

[0003] Limited by orbital resources, the total number of broadcast communication satellites that can be accommodated in the geostationary orbit is limited, making it difficult to meet the growing demand for radio and television. Usually, a single broadcast satellite has only dozens of transponders, and a certain amount of protection bandwidth needs to be reserved between transponders and between satellite signals of each satellite. Therefore, there are quite a lot of unused spectral gaps on the transponder, which undoubtedly causes a waste of precious satellite resources.

[0004] The present invention makes use of the redundancy and power margin existing in the available channels of the Ku-band transparent transponder of the broadcast satellite, places weak signals with a power spectral density lower than the noise by 15 dB or more in the spectral gaps of the transponder, and realizes the connection of weak signals without changing the original use effect of the broadcast signals of the transponder, thereby improving the utilization rate of satellite communication resources. Summary of the Invention

[0005] In order to solve the technical problems existing in the background art, the present invention provides a reliable communication method for weak signals of a Ku transponder of a broadcast satellite based on the transform domain. Using transform domain communication as the weak signal communication technology system, it conducts research on the communication technology of weak signals with a power spectral density lower than the noise by 15 dB or more using the Ku-band transparent transponder of the broadcast satellite at a low code rate, and designs a spectrum sensing and weak signal fast acquisition and tracking method; conducts research on the synchronization and demodulation technology of multi-carrier aggregation deep spread spectrum signals with an extremely low spectral density to achieve the optimal bit error performance.

[0006] In order to solve the technical problems, the technical solution of the present invention is as follows:

[0007] A reliable communication method for weak signals of a Ku transponder of a broadcast satellite based on the transform domain, the method comprising:

[0008] Step S1: During the data frame gap, the transmitting end performs spectrum detection on the Ku transponder of the broadcast satellite in the downlink to provide real-time spectrum sensing results for the generation of the baseband waveform; (To ensure transceiver matching, the receiving end also performs the same spectrum sensing operation);

[0009] Step S2: The transmitter selects available frequency points according to the spectrum sensing result, generates a baseband signal through transform domain communication technology, up-converts the baseband signal to the corresponding Ku-band uplink radio frequency, and then performs filtering processing to obtain a filtered signal;

[0010] Step S3: The transmitter suppresses the peak-to-average ratio of the filtered signal, and after transmitting through the transmitting antenna, it is forwarded by the Ku repeater;

[0011] Step S4: The receiver part performs band-pass filtering on the received signal, down-converts it to the intermediate frequency, performs low-pass filtering, and obtains an intermediate frequency output signal after processing by the noise amplifier;

[0012] Step S5: Through the direct digital control module DDC in the FPGA of the receiver part, the intermediate frequency output signal is down-converted to the baseband orthogonally, and then the high-frequency components are filtered out by the FIR filter to obtain two I and Q baseband signals; perform sparse FFT on the I and Q signals, conjugate multiply with the sparse FFT signals of the local baseband waveform, and then obtain the correlation peak of the signal after inverse sparse FFT, and obtain the demodulated data after reverse mapping.

[0013] Further, in the step S1, the spectrum environment is sampled, and using the flat amplitude shaping method, the spectrum estimation result PSD(k) is compared with the preset threshold Th and quantized into a spectrum amplitude matrix A(k);

[0014]

[0015] If the k-th frequency point in the spectrum is interfered (exceeds the threshold), the corresponding frequency point in A(k) is set to 0, indicating that this frequency point is unavailable; otherwise, it is set to 1, indicating that this frequency point is available.

[0016] Further, in the step S2, the pseudo-random phase θ(k) is generated by M-sequence mapping, and the signal power is adjusted by adding a scaling factor where N and N1 are the total number of frequency points and the number of non-interfered frequency points respectively, and the frequency domain form of the basis function is obtained as:

[0017] S(k) = CA(k)e jθ(k)

[0018] Perform inverse fast Fourier transform IFFT on the frequency domain form of the basis function to obtain its corresponding time-domain basis function s(n):

[0019]

[0020] Further, the step S3 specifically includes:

[0021] Construct a complementary spectrum amplitude matrix corresponding to the spectrum amplitude matrix A(k) It is the complement of A(k) and the all-one matrix I(k), and the relationship is as follows:

[0022]

[0023] The data signal, namely the aforementioned time domain basis function s(n), is composed of the spectrum amplitude matrix A(k) and the pseudo-random phase e jθ(k) The complementary peak suppression signal CPRSc(n) is generated by the complement matrix of the spectrum amplitude matrix A(k) and the same pseudo-random phase generation; Unlike traditional TDCS, data modulation no longer uses a time domain basis function, but a composite signal composed of the data signal and its corresponding CPRS. The composite signal s after peak-to-average ratio suppression CPRS (n) can be expressed as

[0024]

[0025] The modulation mode of the system is cyclic shift keying, and the modulation is a composite signal s CPRS (n) is the prototype for the cyclic shift operation; assuming that the modulation order of the cyclic shift keying modulation is M CCSK , the i-th modulation phase is The corresponding CCSK modulation symbol is expressed as:

[0026]

[0027] The modulation symbols to be transmitted are mapped into corresponding symbol waveforms through Gray code and sent out through the transmitter.

[0028] Furthermore, the step S4 specifically includes:

[0029] After passing through the channel, the received signal y(t) includes the transmitted signal x(t), channel noise w(t), and interference j(t). The receiving end needs to perform synchronous capture and uses the direct time correlation method for signal capture. By inserting a synchronization header into the transmitted signal, the received signal is correlated with the local synchronization header. When the correlation value is greater than the preset threshold, the capture is successful. Otherwise, the local synchronization header is moved point by point and correlation is continued until the capture is successful. Alternatively, synchronization can be performed using the German synchronization method. The difference is that before correlating with the local synchronization header, the signal-to-noise ratio is improved by integrating the odd and even points of the received signal in the frequency domain. Assuming that the system is accurately synchronized, the i-th received signal after sampling can be expressed as:

[0030] y i (n) = x i (n)+w(n)+j(n)

[0031] The received signal is correlated with the locally generated time-domain basis function, and the resulting correlation vector is R i (τ):

[0032]

[0033] where X i (k), Y i (k) are the frequency-domain forms of the i-th transmitted signal and the received signal respectively, τ i is the position of the correlation peak corresponding to the modulated CCSK symbol in the i-th transmitted signal, and conj(·) represents the conjugate form of a complex signal; since when the transmitted signal matches the received signal, the correlation vector R i (τ) should be a real number, so peak decision and data demodulation are achieved by detecting the real part of the correlation vector;

[0034] data i = argmax[real(R i (τ))], τ ∈ [0, 1, …, N - 1]

[0035] where real(·) represents the real part of a complex signal;

[0036] Through the above operations, an intermediate-frequency output signal is obtained after processing.

[0037] Furthermore, the specific steps of step S5 include:

[0038] (1): Window the received in-phase and quadrature spread-spectrum signals, and then preprocess the time domain of the spread-spectrum signals; Windowing means multiplying the spread-spectrum signals by a window function to obtain a finite-length sequence with a length of N; N is a natural number; The obtained finite-length sequence is equally spaced into N / P rows and P columns, and the data in each row is superimposed to obtain an N / P-point sequence;

[0039] (2): Perform FFT operation on the N / P-point sequence obtained in step (1) to obtain a sequence;

[0040] (3): Perform envelope detection and notch filtering on the sequence obtained in step (2); Envelope detection means comparing the sequence obtained in step (2) with a set threshold to obtain the frequency corresponding to when the amplitude crosses the threshold;

[0041] (4): Preprocess the local PN code sequence; The preprocessing means equally spacing the PN code sequence into N / P rows and P columns, and superimposing the data in each row to obtain an N / P-point sequence, and performing FFT operation on the N / P-point sequence, and storing the obtained FFT operation result in the satellite payload;

[0042] (5): Multiply the sequence after notch processing in step (3) with the FFT operation result obtained in step (4) through conjugate multiplication operation;

[0043] (6): Perform N / P-point IFFT operation on the conjugate multiplication result obtained in step (5), thus completing the parallel code phase correlation operation;

[0044] (7): Perform capture decision and verification on the parallel code phase correlation operation obtained in step (6) to obtain the correlation peak of the signal, and obtain the demodulated data after inverse mapping.

[0045] Compared with the prior art, the advantages of the present invention are as follows:

[0046] 1. The present invention extracts the unoccupied part within the bandwidth in the form of available frequency points, constructs waveforms based on this, and simultaneously designs peak suppression signals using the orthogonal characteristics of available frequency points and interfered frequency points for peak-to-average ratio suppression, solving the peak-to-average ratio suppression problem of the TDCS system without the need for sideband information, providing an important technological breakthrough for the specific implementation of TDCS.

[0047] 2. The present invention uses a deep spread spectrum capture and interference suppression method based on sparse Fourier transform, and reduces the number of FFT points through sparse FFT technology, ensuring the real-time processing of extremely high spread spectrum gain for weak signal communication on a spaceborne processor. BRIEF DESCRIPTION OF THE DRAWINGS

[0048] Figure 1 、Structure diagram of weak signal communication system;

[0049] Figure 2 、Structure diagram of transform domain communication;

[0050] Figure 3 、Overall algorithm block diagram of weak signal receiver;

[0051] Figure 4 、Schematic diagram of the principle of complementary peak suppression signal method (CPRS);

[0052] Figure 5 、Flow chart of TDCS based on complementary peak suppression signal method;

[0053] Figure 6 、Schematic diagram of sparse FFT principle; [[ID=4y]]

[0054] Figure 7 、Schematic diagram of the principle of pseudocode parallel capture based on sparse FFT;

[0055] Figure 8 、Schematic diagram of the transmitted signal spectrum generated by constructing spectral notch (the notch is the position of the broadcast television satellite signal). DETAILED DESCRIPTION OF THE INVENTION

[0056] The specific embodiments of the present invention will be described below in conjunction with the embodiments:

[0057] It should be noted that the structures, ratios, sizes, etc. shown in this specification are only used to cooperate with the content disclosed in the specification for those familiar with this technology to understand and read, and are not used to limit the limiting conditions under which the present invention can be implemented. Any modification of the structure, change of the proportional relationship, or adjustment of the size, without affecting the effects that the present invention can produce and the purposes that can be achieved, should still fall within the scope covered by the technical content disclosed in the present invention.

[0058] At the same time, the terms such as "upper", "lower", "left", "right", "middle", and "one" cited in this specification are only for the convenience of clear narration, and are not used to limit the scope under which the present invention can be implemented. The change or adjustment of their relative relationships, without substantial change in the technical content, should also be regarded as the scope within which the present invention can be implemented.

[0059] Embodiment 1:

[0060] As Figure 1 shown in the structural diagram of the weak signal communication system, the entire system transmitter mainly includes two major parts: the generation and processing of baseband signals. First, the baseband signal is generated through transform domain communication technology, and the signal avoids the original signals of radio and television satellites. The emission spectrum is as Figure 8 shown, and through the up-conversion module and the temperature-compensated crystal oscillator, the frequency band is adjusted to the radio frequency (such as 14.0 - 14.5 GHz of Zhongxing 9 satellite). Then, filtering is performed through a band-pass filter to remove the image. Next, amplification is performed through a high-frequency power amplifier. The digital control attenuation module is used to control the high-frequency power amplifier to achieve adjustable output power. Finally, the signal is sent out through the transmitting antenna.

[0061] In order to enable the transmitting end to obtain spectrum sensing ability through a single antenna, the transmitting end performs spectrum detection during the data frame gap through a transceiver switch to provide real-time spectrum sensing results for baseband waveform generation.

[0062] The receiver part of the system performs band-pass filtering on the received signal. In order to ensure the stability of the intermediate frequency output power, the receiver uses automatic gain control for adjustment to output an intermediate frequency signal with stable power.

[0063] To ensure the reliability of the down-conversion module, the receiver is down-converted to the intermediate frequency, low-pass filtered, and processed through a noise amplifier to obtain the intermediate frequency output signal.

[0064] Finally, through the direct digital control (DDC) module in the FPGA, the intermediate frequency signal is down-converted to the baseband orthogonally, and the high-frequency components are filtered out through the FIR filter to obtain the I and Q baseband signals.

[0065] 1) Transformed domain weak signal waveform design. As Figure 2 shown, the generation and modulation of the baseband signal are mainly realized through the transformed domain communication technology. First, the spectrum environment is sampled, and using the flat amplitude shaping method, the spectrum estimation result PSD(k) is compared with the preset threshold Th and quantized into the spectrum amplitude matrix A(k). If the k-th frequency point in the spectrum is interfered (exceeds the threshold), the corresponding frequency point in A(k) is set to 0, indicating that this frequency point is unavailable; otherwise, it is set to 1, indicating that this frequency point is available.

[0066]

[0067] The system ensures the multi-access ability, the noise-like characteristics of the time-domain signal, and the good autocorrelation characteristics through the pseudo-random phase θ(k). Commonly used pseudo-random sequences include M-sequence, Gold sequence, Kasami sequence, chaotic sequence, etc. In this solution, the pseudo-random phase is generated by M-sequence mapping.

[0068] To ensure that the basis function has the same power under different spectrum amplitude matrices, the signal power must be adjusted. Assuming the signal power is 1, by adding the scaling ratio where N and N1 are the total number of frequency points and the number of non-interfered frequency points respectively, the frequency-domain form of the basis function is obtained as:

[0069] S(k) = CA(k)e jθ(k)

[0070] Perform the inverse fast Fourier transform (IFFT) on the frequency-domain form of the basis function to obtain its corresponding time-domain basis function s(n).

[0071]

[0072] 2) Peak-to-average ratio suppression.

[0073] The modulation method of the system is cyclic shift keying, and the modulation performs cyclic shift operations with the time-domain basis function as the prototype. This method utilizes the pseudo-random characteristics of the time-domain basis function, that is, the basis function has good autocorrelation, while the cross-correlation with the cyclic shift waveform is poor. Without peak-to-average ratio suppression, assuming the modulation order of the cyclic shift keying modulation is M CCSK , the i-th modulation phase is Then the corresponding CCSK modulation symbol can be expressed as:

[0074]

[0075] An important feature that distinguishes the transform domain communication system from other communication systems (such as OFDM) is that the peak-to-average power ratio (PAPR) of the transmitted signal varies with the system's pseudo-random phase and the spectrum sensing matrix; moreover, since the system generates modulation symbols through cyclic shift, the PAPR of each modulation symbol is the same, that is, the system PAPR does not change with the variation of the modulation data.

[0076] This scheme proposes to design the peak reduction signal (PRS) by utilizing the orthogonality between the available frequency points and the interfered frequency points. Different from the subcarrier reservation methods that use reserved subcarriers and the extended constellation methods that use all subcarriers for PAPR suppression, this scheme constructs the PRS using all the interfered frequency points for PAPR suppression. The interfered frequency points and the available frequency points are orthogonal to each other, and the generated PRS will not affect the data signal part.

[0077] Figure 4 For the schematic diagram of the principle of the proposed method, if the system spectrum amplitude matrix is A(k), then its complement with the all-ones matrix I(k) is defined as As shown in the figure, A = [1 1 1 1 0 0 1 1 1 0], I = [1 1 1 1 1 1 1 1 1 1]. It can be seen that regardless of how the spectrum situation changes, once the spectrum amplitude matrix is determined, its corresponding complementary spectrum amplitude matrix always maintains the orthogonal and complementary characteristics in the frequency domain, and their relationship is as follows:

[0078]

[0079] The data signal s(n) is generated by the spectrum amplitude matrix A(k) and the pseudo-random phase e jθ(k) while the complementary peak reducing signal (CPRS) c(n) is generated by the complement matrix of the spectrum amplitude matrix A(k) and the same pseudo-random phase. Therefore, it inherits the orthogonal and complementary characteristics of the two amplitude matrices. Different from the traditional TDCS, the data modulation no longer uses the time-domain basis function, but a composite signal composed of the data signal and its corresponding CPRS. After PAPR suppression, the composite signal s CPRS (n) can be expressed as

[0080]

[0081] Taking the composite signal s CPRS (n) as the prototype for cyclic shift operation, the corresponding CCSK modulation symbol can be expressed as:

[0082]

[0083] The modulated symbols to be transmitted are mapped to corresponding symbol waveforms through Gray coding and sent out by the transmitter. If the composite signal wants to obtain a lower peak-to-average power ratio, the pseudo-random phase e jθ(k) must match the all-ones matrix I(k). Therefore, the problem of peak-to-average power ratio suppression for the dynamic spectrum amplitude matrix of TDCS is transformed into a problem of peak-to-average power ratio suppression for a fixed matrix.

[0084] As Figure 5 shown, the entire system process is mainly divided into two parts. The preprocessing stage is used to generate the pseudo-random sequence with low PAPR required by the system during the system initialization stage. The specific implementation steps are similar to the SLM method. First, generate N scan groups of different pseudo-random sequences with length N. After multiplying them with the all-ones matrix I(k) with the same length N and then performing IFFT, N scan groups of time-domain alternative waveforms are obtained. All the alternative waveforms pass through a PAPR comparator to select the waveform with the minimum PAPR and its corresponding pseudo-random sequence. This selected pseudo-random sequence is the optimal pseudo-random sequence, which will be used as the known information for the transmitter and receiver to generate TDCS symbols. Compared with the traditional TDCS, the TDCS transmitter based on CPRS is somewhat simplified, and the minimum peak-to-average power ratio signal generated in the preprocessing stage can be directly used in the generation of modulated symbols. And at the receiving end, only the pseudo-random sequence needs to be replaced with the known optimal pseudo-random sequence. Other parts are the same as those of the traditional TDCS transmitter and receiver.

[0085] It should be noted that the preprocessing stage only needs to run once, and the generated optimal pseudo-random phase and minimum PAPR waveform will not change with the change of the spectrum environment (spectrum amplitude matrix). That is to say, the system can achieve effective peak-to-average power ratio suppression without sideband information. According to the definition of PAPR, the suppressed peak-to-average power ratio can be expressed as

[0086]

[0087] 3) Design of weak signal receiver. After passing through the channel, the received signal y(t) includes the transmitted signal x(t), channel noise w(t), and interference j(t). The receiving end needs to perform synchronous acquisition. Generally, the direct time correlation method is used for signal acquisition. By inserting a synchronization header in the transmitted signal and correlating the received signal with the local synchronization header, when the correlation value is greater than the preset threshold, the acquisition is successful; otherwise, the local synchronization header is moved point by point and correlated again until the acquisition is successful. It is also possible to perform synchronization through the German synchronization method. The difference is that before correlating with the local synchronization header, the signal-to-noise ratio is improved by integrating the even and odd points in the frequency domain of the received signal. Assuming that the system is accurately synchronized, the i-th received signal after sampling can be expressed as:

[0088] yi r(n) = x i (n) + w(n) + j(n)

[0089] The received signal is correlated with the locally generated time-domain basis function, and the resulting correlation vector is R i (τ):

[0090]

[0091] where X i (k), Y i (k) are the frequency-domain forms of the i-th transmitted signal and the received signal respectively, τ i is the position of the correlation peak corresponding to the modulated CCSK symbol in the i-th transmitted signal, and conj(·) represents the conjugate form of a complex signal. Since when the transmitted signal and the received signal match, the correlation vector R i (τ) should be a real number, so peak decision and data demodulation can be achieved by detecting the real part of the correlation vector.

[0092] data i = arg max[real(R i (τ))], τ ∈ [0, 1, …, N - 1]

[0093] where real(·) represents the real part of a complex signal.

[0094] The signal constructed by transform-domain communication technology has a Gaussian-like distribution in the time domain due to the introduction of pseudo-random phases and spreading at available frequency points within the bandwidth, and is orthogonal to environmental interference and has a suppressed power spectral density in the frequency domain, which is suitable for the design of weak signal waveforms

[0095] In order not to affect the power of the transponder and the original signals on it and improve the low-detection characteristics of the signal, a modulation scheme of non-integer period long code spreading is proposed. The receiver algorithm structure is as Figure 3 shown, which is divided into four parts: acquisition, bit synchronization, pseudo-code tracking, and carrier tracking. Since transform-domain communication technology is a frame-burst communication of a multi-carrier system, when using transform-domain communication technology as the main scheme, only the acquisition and bit synchronization parts are needed. In order to take into account direct sequence spreading, the scheme is designed according to the most complex acquisition and tracking.

[0096] In order to reduce the number of FFT points in the synchronization process, the sparse FFT technology is utilized, and multi-frequency channel SFFT-ISFFT fast correlation is adopted. At the same time, the code Doppler compensation technology is used to compensate for the code Doppler shift accumulated during the Tong detection stay, and finally, the Tong detection judgment is completed, ending the fast acquisition process of the pseudo-code at extremely low signal-to-noise ratio. In the tracking part, the code phase tracking is completed by pseudo-code tracking, then symbol synchronization is carried out, and the carrier tracking stage is entered. In the tracking stage, the carrier tracking loop assists the pseudo-code tracking loop to improve the dynamic adaptability.

[0097] 4) Sparse FFT technology. The principle of sparse FFT is as Figure 6 shown. The method for deep spread-spectrum acquisition and interference suppression based on sparse Fourier transform has the steps as Figure 7 shown:

[0098] (1) Perform windowing processing on the received spread-spectrum signal, and then preprocess the time domain of the spread-spectrum signal. Windowing processing means multiplying the spread-spectrum signal by a window function to obtain a finite-length sequence with a length of N. N is a natural number. The obtained finite-length sequence is equally spaced into N / P rows and P columns, and the data in each row is superimposed to obtain an N / P-point sequence.

[0099] (2) Perform FFT operation on the N / P-point sequence obtained in step (1) to obtain a sequence.

[0100] (3) Perform envelope detection and notch filtering on the sequence obtained in step (2). Envelope detection means comparing the sequence obtained in step (2) with a set threshold to obtain the frequency corresponding to when the amplitude crosses the threshold.

[0101] (4) Preprocess the local PN code sequence. The preprocessing mentioned refers to equally spacing the PN code sequence into N / P rows and P columns, superimposing the data in each row to obtain an N / P-point sequence, performing FFT operation on the N / P-point sequence, and storing the obtained FFT operation result in the satellite payload.

[0102] (5) Perform conjugate multiplication operation on the sequence after notch filtering in step (3) and the FFT operation result obtained in step (4).

[0103] (6) Perform N / P-point IFFT operation on the conjugate multiplication result obtained in step (5), and thus complete the parallel code phase correlation operation.

[0104] (7) Perform acquisition decision and verification on the parallel code phase correlation operation obtained in step (6) to obtain the correlation peak of the signal, and obtain the demodulated data after inverse mapping.

[0105] The following makes a detailed description of the present invention in combination with specific signal examples:

[0106] In this simulation experiment, in this example, the carrier frequency f0 of the transmitted signal is 14.2 GHz, the communication modulation method is 32CCSK, the symbol rate of the TDCS system is 20 sps, the information rate is 100 bps, the occupied bandwidth is 3.5 MHz, the spreading gain is 49.5 dB, and the baseband signal spectrum is as Figure 8 shown. After the signal is received by a parabolic antenna with a diameter of 2.4 m, it enters a Ku-band downconverter. The downconverter has a fixed gain of 100 dB and outputs an intermediate frequency of 140 MHz. The 140 MHz intermediate frequency signal is low-pass sampled at 560 MSPS and then undergoes subsequent demodulation processing. Test results: When the transmit power is -17 dBm, the receive power is -137 dBm, and the receiver Eb / No is 10.44 dB. During the experiment, the ratio of the power spectral density of the broadcast television user signal (single-channel code rate of 2.4 kbps) to the power spectral density of the weak transmitted signal is approximately 31 dB, which is much greater than the demodulation signal-to-noise ratio of the broadcast television user signal (about 10 dB), so it will not affect the communication of broadcast television users.

[0107] Example 2:

[0108] The present invention will be described in detail below in conjunction with specific signal examples:

[0109] In this simulation experiment, in this example, the carrier frequency f0 of the transmitted signal is 14.2 GHz, the communication modulation method is 32CCSK, the symbol rate of the TDCS system is 20 sps, the information rate is 100 bps, the occupied bandwidth is 3.5 MHz, the spreading gain is 49.5 dB, and the baseband signal spectrum is as Figure 8 shown. After the signal is received by a parabolic antenna with a diameter of 2.4 m, it enters a Ku-band downconverter. The downconverter has a fixed gain of 100 dB and outputs an intermediate frequency of 140 MHz. The 140 MHz intermediate frequency signal is low-pass sampled at 560 MSPS and then undergoes subsequent demodulation processing. Test results: When the transmit power is -17 dBm, the receive power is -137 dBm, and the receiver Eb / No is 10.44 dB. During the experiment, the ratio of the power spectral density of the broadcast television user signal (single-channel code rate of 2.4 kbps) to the power spectral density of the weak transmitted signal is approximately 31 dB, which is much greater than the demodulation signal-to-noise ratio of the broadcast television user signal (about 10 dB), so it will not affect the communication of broadcast television users.

[0110] The preferred embodiments of the present invention have been described in detail above. However, the present invention is not limited to the above embodiments. Within the knowledge scope of those of ordinary skill in the art, various changes can be made without departing from the gist of the present invention.

[0111] Many other changes and modifications can be made without departing from the concept and scope of the present invention. It should be understood that the present invention is not limited to specific embodiments, and the scope of the present invention is defined by the appended claims.

Claims

1. A reliable communication method for weak signals of a broadcast satellite Ku transponder based on the transform domain, characterized in that, The method includes: Step S1: During the data frame gap, the transmitting end performs spectrum detection on the broadcast satellite downlink Ku repeater to provide real-time spectrum sensing results for baseband waveform generation; in the said Step S1, the spectrum environment is sampled, and using the flat amplitude shaping method, the spectrum estimation result is compared with a preset threshold to be quantized into a spectrum amplitude matrix ; If the th frequency point in the spectrum is interfered, the corresponding frequency point in is set to 0, indicating that this frequency point is unavailable; otherwise it is set to 1, indicating that this frequency point is available; Step S2: The transmitting end selects available frequency points according to the spectrum sensing result, generates a baseband signal through transform domain communication technology, up-converts the baseband signal to the corresponding Ku-band uplink radio frequency, and then performs filtering processing to obtain a filtered signal; Step S3: The transmitting end suppresses the peak-to-average ratio of the filtered signal, emits it through the transmitting antenna, and then forwards it through the Ku transponder; Structure and spectral amplitude matrix Corresponding complementary spectral amplitude matrix , is the complement of the all-ones matrix and their relationship is as follows: Data signal, i.e., time-domain basis function Generated by the spectral amplitude matrix And the pseudo-random phase While the complementary peak suppression signal Is generated by the spectral amplitude matrix Of the complementary set matrix And the same pseudo-random phase; Different from the traditional TDCS, the data modulation no longer uses the time-domain basis function, but a composite signal composed of the data signal and its corresponding CPRS; The composite signal after peak-to-average ratio suppression Can be expressed as The modulation method of the system is cyclic shift keying, and the modulation performs cyclic shift operations with a composite signal as the prototype; assuming that the modulation order of cyclic shift keying modulation is , the th modulation phase is , then the corresponding CCSK modulation symbol is expressed as: The modulated symbols to be transmitted are mapped to corresponding symbol waveforms through Gray coding and sent out by the transmitter; Step S4: The receiver part performs band-pass filtering on the received signal, down-converts it to the intermediate frequency, performs low-pass filtering, and obtains an intermediate frequency output signal after processing through a noise amplifier; Step S5: Through the direct digital control module DDC in the FPGA of the receiver part, the intermediate frequency output signal is quadrature down-converted to the baseband, and then the high-frequency components are filtered out through the FIR filter to obtain I and Q baseband signals; perform sparse FFT on the I and Q signals, conjugate multiply them with the sparse FFT signals of the local baseband waveform, and then obtain the correlation peak of the signal after inverse sparse FFT, and obtain the demodulation data after inverse mapping.

2. The reliable communication method for weak signals of a broadcast satellite Ku transponder based on the transform domain according to claim 1, wherein The step S2 generates a pseudo-random phase through M-sequence mapping , and adjusts the power of the signal by adding a scaling factor , where and are the total number of frequency points and the number of non-interfered frequency points respectively, and the frequency-domain form of the basis function is obtained as follows: Perform an inverse fast Fourier transform (IFFT) on the frequency-domain form of the basis function to obtain its corresponding time-domain basis function : 。 3. A reliable communication method for weak signals of a broadcast satellite Ku transponder based on the transform domain according to claim 1, characterized in that The specific content of step S4 includes: After passing through the channel, the received signal includes the transmitted signal , channel noise and interference ; The receiving end needs to perform synchronization capture. The direct time correlation method is used for signal capture. By inserting a synchronization header in the transmitted signal, the received signal is correlated with the local synchronization header. When the correlation value is greater than the preset threshold, the capture is successful; otherwise, the local synchronization header is moved point by point and the correlation continues until the capture is successful; or synchronization is performed by the German synchronization method. The difference is that before correlating with the local synchronization header, the signal-to-noise ratio is improved by integrating the odd and even points in the frequency domain of the received signal; Assuming accurate system synchronization, the th received signal after sampling is expressed as: The received signal is correlated with the locally generated time-domain basis function, and the resulting correlation vector is :[[]] Among them , are the frequency-domain forms of the th transmitted signal and received signal respectively, is the position of the correlation peak corresponding to the modulated CCSK symbol in the th transmitted signal, represents the conjugate form of a complex signal; since when the transmitted signal matches the received signal, the correlation vector should be a real number, peak decision and data demodulation are realized by detecting the real part of the correlation vector; wherein represents the real part of a complex signal; Through the above operations, that is, the intermediate frequency output signal is obtained after processing.

4. A reliable communication method for weak signals of a broadcast satellite Ku transponder based on the transform domain according to claim 1, characterized in that, The specific content of step S5 includes: (1): Perform windowing processing on the received I and Q spread spectrum signals, and then perform preprocessing on the time domain of the spread spectrum signals; windowing processing means multiplying the spread spectrum signal by a window function to obtain a finite-length sequence, the length of which is N; N is a natural number; the obtained finite-length sequence is equally spaced into N / P rows and P columns, and the data in each row is superimposed to obtain an N / P-point sequence; (2): Perform FFT operation on the N / P-point sequence obtained in step (1) to obtain a sequence; (3): Perform envelope detection and notch processing on the sequence obtained in step (2); envelope detection means comparing the sequence obtained in step (2) with a set threshold to obtain the frequency corresponding to the amplitude passing through the threshold; (4): Preprocess the local PN code sequence; the preprocessing means equally spacing the PN code sequence into N / P rows and P columns, and superimposing the data in each row to obtain an N / P-point sequence, and performing FFT operation on the N / P-point sequence, and storing the obtained FFT operation result in the satellite payload; (5): Perform conjugate multiplication operation on the sequence after notch processing in step (3) and the FFT operation result obtained in step (4); (6): Perform N / P-point IFFT operation on the conjugate multiplication result obtained in step (5), and thus complete the parallel code phase correlation operation; (7): Perform capture decision and verification on the parallel code phase correlation operation obtained in step (6) to obtain the correlation peak of the signal, and obtain the demodulation data after inverse mapping.

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