A Control Method for a Flying Capacitor Seven-Level Soft-Switching Power Amplifier

Abstract

A control method for a flying capacitor seven-level soft-switching power amplifier, which relates to the field of ultra-precision motor drive control. The method is as follows: Determine the given current i set , and require that i set be equal to the average current i ave of a single cycle of the trapezoidal filtered current, and build the shape of the trapezoidal filtered current based on this; by analyzing the curve of the current changing with time, extract the slope of the current change. According to the maximum level V dc , the minimum level -V dc and the output voltage V out , determine the slopes α and β, select the voltage level closest to the output voltage V out as the intermediate level of the trapezoidal filtered current, and obtain the slope γ; based on the principle that the charging charge Q c and the discharging charge Q f of the flying capacitor are equal under a single switching cycle, combined with the obtained slopes and the average current i ave equivalent principle, build a mathematical expression of the filtered current, calculate the time t [k] at each moment and the corresponding current i Lf[k] , generate a PWM control signal with periodic variation through the calculated time t [k] ; establish the upper and lower boundaries of the flying capacitor voltage, sample the flying capacitor voltage V cqx in real time and compensate it into the trapezoidal filtered current calculation model; control the switching tubes of the flying capacitor seven-level soft-switching power amplifier according to the PWM signal to obtain the output voltage V out and the output current i out , and build a current control closed-loop system. It solves the disadvantages that the switching loss of the traditional multi-level power amplifier increases sharply with the increase of the switching frequency and the electromagnetic interference is serious under the hard-switching condition.

Description

Technical Field

[0001] The present invention relates to the field of ultra-precision motor drive control, and particularly to a control method for a flying capacitor seven-level soft-switching power amplifier. Background Art

[0002] The flying capacitor seven-level power amplifier can generate seven possible voltage levels. The most commonly used method for the traditional flying capacitor seven-level power amplifier is the carrier phase-shifted PWM method. However, at a low modulation index, the harmonic performance of the output voltage of this method is poor and it is not suitable for the semiconductor field. The position error of the moving workbench in a lithography machine is determined by the non-linear error of the power amplifier, and it is required that the current distortion is less than -100 dB, which poses strict requirements on the output current accuracy and bandwidth of the flying capacitor power amplifier involved in motor drive. Therefore, researching the control method of multi-level power amplifiers with wide bandwidth, high voltage and high power is of great significance to the performance of ultra-precision motor systems.

[0003] Figure 2 For the traditional flying capacitor seven-level power amplifier, seven-level output is achieved on the load side. The multi-level synthesis method is adopted to reduce the distortion of the load-side voltage (current), and at the same time, the voltage stress of the device is also reduced. However, since the device operates under hard-switching conditions, the system efficiency is limited. In occasions where extremely high positioning accuracy is required, a high-dynamic and high-precision response current needs to be provided. The output current ripple and loss of the traditional carrier phase-shifted modulation strategy are still large and cannot meet the application requirements. Summary of the Invention

[0004] The present invention is to solve the problem that in occasions where extremely high positioning accuracy is required for the traditional flying capacitor seven-level power amplifier, the output current ripple and loss of the traditional carrier phase-shifted modulation strategy are large and cannot meet the application requirements, and provides a control method for a flying capacitor seven-level soft-switching power amplifier.

[0005] The technical solution adopted by the present invention is as follows:

[0006] A control method for a flying capacitor seven-level soft-switching power amplifier, which is also applicable to a topology-reconfigurable modular multi-level power amplifier, and only needs to adjust the control method according to the number of levels. The present invention describes this method with a flying capacitor seven-level soft-switching power amplifier ( Figure 2 ) as the topology structure, and this method includes the following steps:

[0007] Step 1: Apply a DC voltage source V dc , to obtain a trapezoidal filtered current shape, and the stage gradients include α, γ, and β;

[0008] Step 2: The average current i in a single cycle setCompare with the output current \(i\) amplified by \(k\) times out and then amplify by \(C\) e times to obtain the average current set value \(i\) of a single cycle set * Send the average current set value \(i\) of a single cycle set * to the filtered current generation calculation model;

[0009] Step 3: After calculation by the filtered current generation calculation model, obtain the calculated time \(t\) [k] and the filtered current \(i\) Lf[k] Generate a PWM control signal with periodic variation through the calculated time \(t\) [k] and the filtered current \(i\) Lf[k] ;

[0010] Step 4: Input the PWM control signal into the flying capacitor seven-level soft-switching power amplifier to obtain the output voltage \(V\) out and the output current \(i\) out ;

[0011] Step 5: Amplify the output current \(i\) out by \(k\) times to obtain the output current set value \(i\) out * and return to Step 2.

[0012] Preferably, the specific obtaining method of the filtered current generation calculation model in Step 2 is as follows:

[0013] Step 2-1: Select the optimal intermediate level \(V\) among 2, 1, 0, -1, -2 according to the DC voltage source \(V\) dc and the output voltage \(V\) obtained in Step 4 out , and at the same time obtain the DC voltage source \(m\) sn ; [k] ;

[0014] Step 2-2: Calculate the trapezoidal filtered current slope according to the optimal intermediate level \(V\) sn , the DC voltage source \(V\) dc and the output voltage \(V\) out to obtain the stage gradients \(\alpha\) [k] , \(\gamma\) [k] , \(\beta\) [k] ; 1 ≤ \(k\) ≤ \(n\), \(n\) is an integer;

[0015] Step 2-3: Select the calculation method for generating a trapezoidal wave according to the DC voltage source \(m\) [k] recorded in Step 2-1 and the stage gradients \(\alpha\) [k] , \(\gamma\) [k] , \(\beta\) [k] recorded in Step 2-2;

[0016] Step 24: Under the joint action of establishing the upper and lower boundary constraint conditions of the trapezoidal filter current and the capacitor voltage circulating current equalization method, obtain the filter current generation calculation model according to the calculation method for generating a trapezoidal wave selected in Step 23.

[0017] Preferably, the equation for calculating the slope of the trapezoidal filter current in Step 22 is as follows:

[0018]

[0019] L f represents the filter inductor.

[0020] Preferably, the filter current generation calculation model includes 10 switching modes, specifically as follows:

[0021] Switching mode 1: In the time period from t0 to t1, switches S1…S6 are turned on, and S7…S 12 are turned off. The upper DC power supply V dc provides energy to the load, and the current flowing through the inductor L f increases linearly with a slope of α. At this time, V sn =V dc ;

[0022] Switching mode 2: In the time period from t1 to t2, switches S1…S5 are turned on, and S6…S 12 are turned off. The upper DC power supply V dc charges the parallel capacitor of S7 reversely through the flying capacitor C q1 until the capacitor voltage of the parallel capacitor becomes zero;

[0023] Switching mode 3: In the time period from t2 to t3, switches S1…S5 and S7 are turned on, and S6, S8…S 12 are turned off. At this time, since the voltage of the parallel capacitor of S7 becomes zero, the current flows through the switching diode to the inductor L f with a slope of γ. At this time, V sn =2 / 3V dc ;

[0024] Switching mode 4: In the time period from t3 to t4, switch S7 is turned on, and switches S1…S6, S8…S 12 are turned off. The inductor current i Lf charges the parallel capacitors of switches S8, S9, S 10 , S 11 , S 12 reversely until the capacitor voltage of the parallel capacitor becomes zero;

[0025] Switching mode 5: In the time period from t4 to t5, switches S7…S 12Conduct, switch tubes S1…S6 are turned off, inductor current i Lf Passes through switches S7…S 12 The reverse diode conducts, the inductor current continuously decreases with a slope of β, and finally the current becomes zero. At this time, V sn =-V dc ;

[0026] Switch mode 6: In the time period from t5 to t6, the states of the switch tubes are the same as those in mode 5, but at this time the inductor current is reversed, -V dc Is applied to inductor L f , and the inductor current increases linearly in the reverse direction with a slope of β. At this time, V sn =-V dc ;

[0027] Switch mode 7: In the time period from t6 to t7, switch tube S7 conducts, switch tubes S1…S6, S8…S 12 Are turned off, V dc Charges the parallel capacitors of S1…S5 in the reverse direction through the flying capacitor C q1 Until the capacitance voltage of the parallel capacitor becomes zero;

[0028] Switch mode 8: In the time period from t7 to t8, switch tubes S1…S5, S7 conduct, switch tubes S6, S8…S 12 Are turned off. At this time, since the voltage of the parallel capacitor of S1…S5 becomes zero, the current flows back to the power supply through the switch diode with a slope of γ. At this time, V sn =2 / 3V dc ;

[0029] Switch mode 9: In the time period from t8 to t9, switch tubes S1…S5 conduct, switch tubes S6…S 12 Are turned off, and the current charges the parallel capacitor of S6 in the reverse direction until the capacitance voltage of the parallel capacitor becomes zero;

[0030] Switch mode 10: In the time period from t9 to t 10 , switch tubes S1…S6 conduct, switch tubes S7…S 12 Are turned off, inductor current i Lf Conducts through the reverse diodes of switch tubes S1…S6, the inductor current continuously decreases with a slope of α, and finally the current becomes zero, and finally returns to switch mode 1. At this time, V sn =V dc .

[0031] Preferably, the specific selection method for the calculation method of generating a trapezoidal wave in the second and third steps is as follows:

[0032] Step 231: By giving the charge and discharge times T1, T2 of the flying capacitor and the average current i of a single cycleset Calculate the trapezoidal current generation calculation model within a single period;

[0033] Step 232: Since the dead time is much smaller than the time of each switching mode, the dead time is first ignored in the calculation process;

[0034] Step 233: Although the charge amount of the flying capacitor charging and discharging within each period is calculated, it is difficult to keep it equal within a sine period.

[0035] Therefore, in order to maintain the voltage balance of the capacitors C q1 ~C q5 in the flying capacitor type power amplifier and make the charging and discharging charges of the flying capacitor equal within a single period, as shown in Equation (2):

[0036]

[0037] t1 represents the time duration when the inductor current i Lf starts from zero, the switching tubes S1…S6 are turned on, and S7…S 12 is in the off mode;

[0038] t2 represents the time duration when the switching tubes S7…S 12 are turned on, S1…S6 are turned off, until the inductor current i Lf becomes zero;

[0039] t3 represents the time duration when the inductor current i Lf starts from zero, the switching tubes S7…S 12 are turned on, and S1…S6 are in the off mode;

[0040] t4 represents the time duration when the switching tubes S1…S6 are turned on, S7…S 12 are turned off, until the inductor current i Lf becomes zero;

[0041] Step 234: Equation (2) effectively ensures that the flying capacitor remains at a stable value within each period. However, since i1, i2, i3, and i4 are unknown at this time, additional conditions are required;

[0042] Step 235: Assume that the seven-level soft-switching power amplifier outputs sinusoidal current and voltage, and the trapezoidal current is equal to a constant value after filtering, that is, the average current i set required for a single period. As shown in Equation (3), the product of the filtered average value of the trapezoidal current and the period is equal to the change in the charge amount;

[0043]

[0044] i1 represents the inductor current iLf Starting from zero, switching transistors S1…S6 are turned on, and S7…S 12 The current magnitude at the end of the turn-off mode;

[0045] i2 represents the inductor current i Lf When it is greater than zero, switching transistors S1…S5 and S7 are turned on, and S6, S8…S 12 The current magnitude at the end of the turn-off mode;

[0046] i3 represents the current magnitude of switching transistors S7…S 12 When turned on, the current magnitude at the end of the turn-off mode of S1…S6;

[0047] i4 represents the inductor current i Lf When it is less than zero, switching transistors S1…S5 and S7 are turned on, and S6, S8…S 12 The current magnitude at the end of the turn-off mode;

[0048] Step 236: Generate α, β, and γ for each time period in the calculation model according to the filtered current. Therefore, the relationships between i1, i2, i3, i4 and each time period are shown in Equations (4) and (5) as follows:

[0049]

[0050] i2 = i1 + γT, i3 = i4 - γT2 (5)

[0051] Preferably, the specific selection method for the calculation method of generating a trapezoidal wave includes four types:

[0052] The first type: the average current i of a single cycle set <0, V sn -V out >0;

[0053] The second type: the average current i of a single cycle set >0, V sn -V out >0;

[0054] The third type: the average current i of a single cycle set <0, V sn -V out >0;

[0055] The fourth type: the average current i of a single cycle set >0, V sn -V out <0;

[0056] According to the actual situation, select the above four types as the basis for subsequent simulation.

[0057] Preferably, i set has both positive and negative values. Depending on the actual situation, the magnitudes of T1 and T2 are different; when i set > 0, T1 < T2; when i set < 0, T1 > T2.

[0058] Preferably, for the stage gradients α and β in the filtered current generation calculation model, V dc and -V dc are selected as the required distribution levels. For the levels V sn corresponding to the intermediate stage part of the trapezoidal filtered current, the selected levels depend on the magnitude of the output voltage V out .

[0059] Beneficial effects: A control method for a flying capacitor seven-level soft-switching power amplifier according to the present invention will be described by taking the control method of the flying capacitor seven-level soft-switching power amplifier as an example. The topological structure of the flying capacitor seven-level soft-switching power amplifier adds resonant capacitors at both ends of the MOS transistors in the traditional form. The present invention proposes a control method for a flying capacitor seven-level soft-switching power amplifier, which is a high-precision control method for a flying capacitor seven-level soft-switching power amplifier based on trapezoidal filtered current. The proposed control method based on trapezoidal filtered current can make the power devices operate under soft-switching conditions, reducing switching losses and electromagnetic interference. In addition, the three-level modulation strategy based on trapezoidal filtered current proposed by the present invention further reduces the system losses and current distortion on the output side compared with the traditional two-level modulation, and can meet the requirements in high-power or ultra-precise positioning applications.

[0060] The present invention solves the disadvantages of traditional multi-level power amplifiers operating under hard-switching conditions, where the switching losses increase sharply with the increase of the switching frequency and the electromagnetic interference is serious (EMI). At the same time, it reduces the peak value and root mean square of the filtered current, resulting in lower switching losses, lower conduction losses, improving the power density of the system, and the harmonic content of the trapezoidal filtered current control method is relatively low. BRIEF DESCRIPTION OF THE DRAWINGS

[0061] Figure 1 is a schematic diagram of the control flow of a control method for a flying capacitor seven-level soft-switching power amplifier;

[0062] Figure 2 is a diagram of a traditional flying capacitor seven-level power amplifier;

[0063] Figure 3 is Figure 1 a schematic diagram of Mode 1 of

[0064] Figure 4 is Figure 1 a schematic diagram of Mode 2 of

[0065] Figure 5 is Figure 1 a schematic diagram of Mode 3;

[0066] Figure 6 is Figure 1 a schematic diagram of Mode 4;

[0067] Figure 7 is Figure 1 a schematic diagram of Mode 5;

[0068] Figure 8 is Figure 1 a schematic diagram of Mode 6;

[0069] Figure 9 is Figure 1 a schematic diagram of Mode 7;

[0070] Figure 10 is Figure 1 a schematic diagram of Mode 8;

[0071] Figure 11 is Figure 1 a schematic diagram of Mode 9;

[0072] Figure 12 is Figure 1 a schematic diagram of Mode 10;

[0073] Figure 13 is a schematic diagram of the level correspondence relationship between the trapezoidal filter current and the distribution;

[0074] Figure 14 is a schematic diagram of the schematic description of the overall controller of the trapezoidal filter current;

[0075] Figure 15 is a schematic diagram of the structure of the topology-reconfigurable modular multilevel power amplifier;

[0076] Figure 16 is a schematic diagram of two-stage modulation. Detailed implementation manners

[0077] Detailed implementation manner 1. Refer to Figures 1 to 15 to illustrate this implementation manner. A control method for a flying-capacitor seven-level soft-switching power amplifier described in this implementation manner is applicable to a topology-reconfigurable modular multilevel power amplifier. Taking a flying-capacitor seven-level soft-switching power amplifier as the topology structure to describe this method, this method includes the following steps:

[0078] Step 1: Apply a DC voltage source V dc , and obtain the trapezoidal filter current shape, and the stage gradients include α, γ, and β;

[0079] Step 2: The average current i in a single cycleset Compare with the output current \(i\) amplified by \(k\) times out After comparison, amplify it by \(C\) times e times to obtain the average current set value \(i\) of a single cycle set * Send the average current set value \(i\) of a single cycle set * into the filtered current generation calculation model;

[0080] Step 3: After calculation by the filtered current generation calculation model, obtain the calculated time \(t\) [k] and the filtered current \(i\) Lf[k] Generate a PWM control signal with periodic variation through the calculated time \(t\) [k] and the filtered current \(i\) Lf[k] Generate a PWM control signal with periodic variation;

[0081] Step 4: Input the PWM control signal into the flying capacitor seven-level soft-switching power amplifier to obtain the output voltage \(V\) out and the output current \(i\) out ;

[0082] Step 5: Amplify the output current \(i\) by \(k\) times to obtain the output current set value \(i\) out and return to Step 2. out *

[0083] Specific Embodiment 2. This embodiment is a further description of the control method of a flying capacitor seven-level soft-switching power amplifier described in Embodiment 1

[0084] The specific obtaining method of the filtered current generation calculation model in Step 2 is as follows:

[0085] Step 2-1: According to the DC voltage source \(V\) dc and the output voltage \(V\) obtained in Step 4 out Select the optimal intermediate level \(V\) among 2, 1, 0, -1, -2 sn and obtain the calculation method selection basis \(m\) at the same time [k] ;

[0086] Step 2-2: According to the optimal intermediate level \(V\) sn , the DC voltage source \(V\) dc and the output voltage \(V\) out Calculate the trapezoidal filtered current slope to obtain the stage gradients \(\alpha\) [k] , \(\gamma\) [k] , \(\beta\) [k] ; 1 ≤ k ≤ n, n is an integer;

[0087] Step 2-3: According to the calculation method selection basis \(m[k]\) recorded in Step 2-1 and the stage gradients \(\alpha\) recorded in Step 2-2​[k] , γ [k] , β [k] Select the calculation method for generating a trapezoidal wave;

[0088] Step 24: Under the joint action of establishing the upper and lower boundary constraint conditions of the trapezoidal filter current and the capacitor voltage circulating current equalization method, obtain the filter current generation calculation model according to the calculation method for generating a trapezoidal wave selected in Step 23.

[0089] Specific Embodiment 3: This embodiment further describes a control method for a flying capacitor seven-level soft-switching power amplifier described in Embodiment 2.

[0090] The equation for calculating the slope of the trapezoidal filter current in Step 22 is as follows:

[0091]

[0092] Specific Embodiment 4: This embodiment further describes a control method for a flying capacitor seven-level soft-switching power amplifier described in Embodiment 2. The filter current generation calculation model includes 10 switching modes, specifically as follows:

[0093] Switching Mode 1: In the time period from t0 to t1, the switching transistors S1...S6 are turned on, and S7...S 12 are turned off. The upper DC power supply V dc provides energy to the load, and the current flowing through the inductor L f increases linearly, with a slope of α. At this time, V sn = V dc ;

[0094] Switching Mode 2: In the time period from t1 to t2, the switching transistors S1...S5 are turned on, and S6...S 12 are turned off. The upper DC power supply V dc charges the parallel capacitor of S7 reversely through the flying capacitor C q1 until the capacitor voltage of the parallel capacitor becomes zero;

[0095] Switching Mode 3: In the time period from t2 to t3, the switching transistors S1...S5, S7 are turned on, and S6, S8...S 12 are turned off. At this time, since the voltage of the parallel capacitor of S7 becomes zero, the current flows through the switching diode to the inductor L f above, with a slope of γ. At this time, V sn = 2 / 3V dc ;

[0096] Switching Mode 4: In the time period from t3 to t4, the switching transistor S7 is turned on, and the switching transistors S1...S6, S8...S 12 are turned off. The inductor current i LfReverse charge the parallel capacitors of switches S8, S9, S 10 , S 11 , S 12 until the capacitance voltage of the parallel capacitor becomes zero;

[0097] Switch mode 5: In the time period from t4 to t5, switch tubes S7…S 12 conduct, switch tubes S1…S6 turn off, and the inductor current i Lf flows through the reverse diodes of switches S7…S 12 and decreases continuously with a slope of β. Eventually, the current becomes zero. At this time, V sn =-V dc ;

[0098] Switch mode 6: In the time period from t5 to t6, the states of the switch tubes are the same as those in mode 5, but at this time the inductor current is reversed, and -V dc is applied to the inductor L f . The inductor current increases linearly in the reverse direction with a slope of β. At this time, V sn =-V dc ;

[0099] Switch mode 7: In the time period from t6 to t7, switch tube S7 conducts, switch tubes S1…S6, S8…S 12 turn off, and V dc charges the parallel capacitors of S1…S5 reversely through the flying capacitor C q1 until the capacitance voltage of the parallel capacitor becomes zero;

[0100] Switch mode 8: In the time period from t7 to t8, switch tubes S1…S5, S7 conduct, switch tubes S6, S8…S 12 turn off. At this time, since the voltage of the parallel capacitor of S1…S5 becomes zero, the current flows back to the power supply through the switch diode with a slope of γ. At this time, V sn =2 / 3V dc ;

[0101] Switch mode 9: In the time period from t8 to t9, switch tubes S1…S5 conduct, switch tubes S6…S 12 turn off, and the current charges the parallel capacitor of S6 reversely until the capacitance voltage of the parallel capacitor becomes zero;

[0102] Switch mode 10: In the time period from t9 to t 10 , switch tubes S1…S6 conduct, switch tubes S7…S 12 turn off, and the inductor current i Lf flows through the reverse diodes of switches S1…S6 and decreases continuously with a slope of α. Eventually, the current becomes zero and finally returns to switch mode 1. At this time, V sn= V dc 。

[0103] Specific Embodiment 5. This embodiment further elaborates on the control method of a flying capacitor seven-level soft-switching power amplifier described in Embodiment 2. The specific selection method for choosing the calculation method for generating a trapezoidal wave in Steps 23 is as follows:

[0104] Step 231: Calculate the trapezoidal current generation calculation model within a single period by given charge and discharge times T1, T2 of the flying capacitor and the average current i in a single period set ;

[0105] Step 232: Since the dead time is much smaller than each switching mode time, the dead time is first ignored in the calculation process;

[0106] Step 233: Although the charge amount of the flying capacitor during charge and discharge in each period is calculated, it is difficult to keep it equal within a sine period.

[0107] Therefore, in order to maintain the voltage balance of capacitors C q1 ~C q5 in the flying capacitor type power amplifier and make the charge of the flying capacitor during charge and discharge equal within a single period, as shown in Equation (2):

[0108]

[0109] Step 234: Equation (2) effectively ensures that the flying capacitor is maintained at a stable value in each period. However, since i1, i2, i3, and i4 are unknown at this time, additional conditions are required;

[0110] Step 235: Assume that the seven-level soft-switching power amplifier outputs sinusoidal current and voltage, and the trapezoidal current is equal to a fixed value after filtering, that is, the average current i set required in a single period, as shown in Equation (3). The product of the filtered average value of the trapezoidal current and the period is equal to the change amount of the charge amount;

[0111]

[0112] Step 236: According to α, β, and γ in each time period in the filtered current generation calculation model, the relationships between i1, i2, i3, i4 and each time period are shown in Equations (4) and (5):

[0113]

[0114] i2 = i1 + γT, i3 = i4 - γT2 (5)

[0115] Specific Embodiment Six. This embodiment further elaborates on the control method of a flying-capacitor seven-level soft-switching power amplifier described in Embodiment Five. The specific selection method for choosing the calculation method of generating a trapezoidal wave includes four types:

[0116] The first type: The average current i in a single cycle set <0, V sn -V out >0;

[0117] The second type: The average current i in a single cycle set >0, V sn -V out >0;

[0118] The third type: The average current i in a single cycle set <0, V sn -V out >0;

[0119] The fourth type: The average current i in a single cycle set >0, V sn -V out <0;

[0120] According to the actual situation, select the above four types as the basis for subsequent simulation.

[0121] Specific Embodiment Seven. This embodiment further elaborates on the control method of a flying-capacitor seven-level soft-switching power amplifier described in Embodiment Six. i set has both positive and negative values. According to the actual situation, the sizes of T1 and T2 are different; when i set >0, T1 < T2; when i set <0, T1 > T2.

[0122] Specific Embodiment Eight. This embodiment further elaborates on the control method of a flying-capacitor seven-level soft-switching power amplifier described in Embodiment One.

[0123] For the stage gradients α and β in the filtered current generation calculation model, select V dc and -V dc as the required allocated levels. For the levels V sn corresponding to the middle stage part of the trapezoidal filtered current, the selected levels depend on the magnitude of the output voltage V out .

[0124] Working principle: The non-linear output capacitance (i.e., the parallel resonant capacitance) of the switching device MOSFET can form a resonant circuit with the filter inductor. Due to the relatively large filter inductor, the filter current i Lf[k]The non - linear output capacitance can be quickly discharged, enabling rapid conversion of the drain - source voltage of the switching device, reducing the non - linear voltage error introduced during the resonance process, and suppressing non - linear current distortion.

[0125] By optimizing the intermediate level and switching method, the active balance of the flying - capacitor voltage can be achieved. Based on this, a resonant commutation hybrid mode of the power amplifier is established. By controlling the operating state and time of the resonant circuit of the non - linear output capacitance of the switching device, zero - voltage turn - on / off (ZVS) of the switching device can be realized. Meanwhile, during the zero - voltage switching process, the charging and discharging conditions of the resonant capacitor can be regulated to minimize the dead - time while ensuring the conversion of the resonant commutation hybrid mode, thereby reducing the distortion of the output current. The working modes of the proposed topology are as Figures 3 to 12 shown, namely, the 10 switching modes of the filter - current generation calculation model.

[0126] The present invention solves the disadvantages of traditional multilevel power amplifiers that the switching loss increases sharply with the increase of the switching frequency and the serious electromagnetic interference (EMI) under hard - switching conditions. At the same time, it reduces the peak value and root - mean - square value of the filter current, resulting in lower switching losses, lower conduction losses, and improved power density of the system. Meanwhile, the harmonic content of the trapezoidal filter - current control method is relatively low. Based on the condition of the trapezoidal filter current, to achieve the active balance of the flying - capacitor voltage, the present invention provides a complete control scheme. By giving the charging and discharging times T1, T2 of the flying - capacitor and the average current i set in a single period to calculate the filter - current generation calculation model.

[0127] Figure 1 The sampling circuit, MCU control circuit, and driving circuit in [description] are the hardware parts of this control method when making a physical object, which will not be described here as they are common knowledge in the art.

[0128] The present invention proposes a control method applicable to flying - capacitor multilevel power amplifiers. When an additional voltage level is actively applied, a trapezoidal filter - current shape is obtained, as Figure 13 shown. For the stage gradients α and β, V dc and - V dc are selected as the required distribution levels. For the level V sn corresponding to the middle - stage part of the trapezoidal filter current, the selected level depends on the output voltage V out . According to the level closest to V out , combined with the resonant mode and commutation characteristics of soft - switching, the optimal distribution is carried out. The slope equation of the T - type current is:

[0129]

[0130] This will result in a reduction in the RMS current and peak current through the filter inductor, and the time and current value of the ladder filter current can be fully described in the controller, by controlling the operating state and time of the device's resonant capacitor resonant circuit, thereby achieving zero voltage on / off of the switching device.

[0131] 2. Ladder filter current

[0132] Ladder filter current Figure 14 As shown, by giving the flying capacitor charging and discharging time T1, T2 and the average current i of a single cycle set To calculate a complete description of the time and current changes of each operating mode of the ladder filter current within a single cycle.

[0133] Since the dead time is much smaller than the switching mode time, the dead time can be ignored in the calculation process. Although the charge amount of the flying capacitor in each cycle can be calculated, it is difficult to keep it equal in a sinusoidal cycle. Therefore, in order to maintain the capacitor C q1 ~C q5 The voltage balance of this scheme keeps the charge and discharge charge of the flying capacitor equal in a single cycle, as shown in formula (2):

[0134]

[0135] Formula 2 can effectively ensure that the flying capacitor is kept at a stable value in each cycle, but since i1, i2, i3, and i4 are unknown at this time, additional conditions are required. Assuming that the seven-level soft-switching power amplifier outputs sinusoidal current and voltage, the trapezoidal current is equal to a constant value after filtering, that is, the required i set The value is, as shown in equation (3), the product of the filtered average value of the trapezoidal current and the period and the change in the charge amount.

[0136]

[0137] In addition, we have given α, β and γ of each time period of the trapezoidal current, so the relationship between i1, i2, i3, i4 and each time period can be expressed, as shown in equations (4) and (5).

[0138]

[0139] i2=i1+γT i3=i4-γT2 (5)

[0140] By combining equations (1)-(5), the values of each time period of the T-type filter current and the magnitude of the current in a single cycle can be obtained, and a complete description of the current in a single cycle can be obtained.

[0141] Since i set has both positive and negative values, the magnitudes of T1 and T2 will be different according to the shown situation. When i set > 0, T1 < T2. When i set < 0, T1 > T2.

[0142] Figure 15 is the structural schematic diagram of the topology-reconfigurable modular multilevel power amplifier;

[0143] The multilevel circuit includes power switches S1, S2, S3, …, S 2n-3 , S 2n-2 , resonant capacitors C r1 , C r2 , C r3 , …, C r(2n-3) , C r(2n-2) and fast capacitors C q1 , C q2 , C q3 , …, C q(n-3) , C q(n-2) ; n is a positive integer greater than or equal to 3;

[0144] The source of power switch S1 is connected to the drain of power switch S2, the source of power switch S2 is connected to the drain of power switch S3, the source of power switch S3 is connected to the drain of power switch S4, …, the source of power switch S 2n-3 is connected to the drain of power switch S 2n-2 ;

[0145] The resonant capacitors C r1 , C r2 , C r3 , …, C r(2n-3) , C r(2n-2) are respectively connected in parallel with power switches S1, S2, S3, …, S 2n-3 , S 2n-2 ;

[0146] The fast capacitor C q1 is connected between the source of power switch S n-2 and the drain of power switch S n+1 . The fast capacitor C q2 is connected between the source of power switch S n-3 and the drain of power switch S n+2 . The fast capacitor C q3 is connected between the source of power switch S n-4 and the drain of power switch S n+3 . …, The fast capacitor C q(n-3) is connected between the source of power switch S2 and the drain of power switch S 2n-3 . The fast capacitor Cq(n-2) is connected between the source of the power switch S1 and the drain of the power switch S 2n-2 ;

[0147] The drain of the power switch S1 is connected to the positive terminal of the power supply, and the source of the power switch S 2n-2 is connected to the negative terminal of the power supply;

[0148] The source of the power switch S n-1 is connected to the input end of the filter circuit, and the output end of the filter circuit is connected to the motor coil winding load.

[0149] The motor coil winding load includes an inductor L, a resistor R, and a motor back electromotive force e f ;

[0150] One end of the inductor L is connected to one end of the resistor R, the other end of the resistor R is connected to the positive pole of the motor back electromotive force e f , and the negative pole of the motor back electromotive force e f is grounded, and the other end of the inductor L serves as the input end of the motor coil winding load.

[0151] The filter circuit adopts an LCL filter circuit, and the LCL filter circuit includes filter capacitors C f1 , C f2 , filter inductor L f and the filtering effect of the load inductor L;

[0152] One end of the filter capacitor C f1 is connected to the positive terminal of the power supply, one end of the filter capacitor C f2 is connected to the negative terminal of the power supply, the other end of the filter capacitor C f1 is connected to the other end of the filter capacitor C f2 , and the connection point of the filter capacitor C f1 and the filter capacitor C f2 is connected to one end of the filter inductor L f , and the connection point of the filter capacitor C f1 and the filter capacitor C f2 serves as the output end of the filter circuit, and the other end of the filter inductor L f serves as the input end of the filter circuit.

[0153] A control method based on a flying capacitor seven-level soft-switching power amplifier as a topological structure is completed by the control method of the present invention. The soft-switching power amplifier is implemented based on the traditional seven-level soft-switching power amplifier ( Figure 2 ); its topological structure is that a resonant capacitor is connected in parallel at both ends of the MOS tube of the traditional seven-level soft-switching power amplifier;

[0154] Resonant inductor L f , inductor L, resistor R, constant current source ef are connected in series in turn, and the resonant inductor L f has one end connected to the output end of the traditional seven-level soft-switching power amplifier, and the constant current source e f has the other end grounded.

[0155] Explanation: Two-level modulation ( Figure 16 ) generally refers to a two-level modulation scheme. At this time, the filter inductor current adopts a triangular wave form, and the intermediate voltage is not actively used, as Figure 2 shown.

[0156] Three-level modulation refers to the three-level modulation scheme proposed by the present invention. The filter inductor current is trapezoidal. Compared with the triangular wave form, the peak value and root mean square of the trapezoidal filter current are smaller, resulting in lower conduction loss and switching loss. The principle of the present invention is to approximate the sine wave by using the average value of countless small switching cycles within a sine period. Since the output voltage is constantly changing, in order to make the average voltage approximate the sine wave, we select the intermediate level closest to the output voltage in each small switching cycle.

[0157] The present invention is illustrated by several specific embodiments. Those skilled in the art should understand that various transformations and equivalent substitutions can be made to the present invention without departing from the scope of the present invention. In addition, various modifications can be made to the present invention for specific situations or circumstances without departing from the scope of the present invention. Therefore, the present invention is not limited to the specific embodiments disclosed, but should include all embodiments falling within the scope of the claims of the present invention.

Claims

1. A control method for a flying capacitor seven-level soft-switching power amplifier, which is applicable to a topology-reconfigurable modular multilevel power amplifier. This method is described with a flying capacitor seven-level soft-switching power amplifier as the topology structure. It is characterized in that The method includes the following steps: Step 1: Apply a DC voltage source V dc , to obtain a trapezoidal filtered current shape, and the stage gradients include α, γ, and β; Step 2: Average current i for a single cycle set is compared with the output current i amplified by k times out and then amplified by C e times to obtain the average current set value i for a single cycle set * . The average current set value i for a single cycle set * is fed into the filtered current generation calculation model; The specific method for obtaining the filtering current generation calculation model in the second step is as follows: Step 2-1: According to the DC voltage source V dc and the output voltage V out obtained in Step 4, select the optimal intermediate level V sn from 2, 1, 0, -1, -2, and at the same time obtain the basis m [k] for selecting the calculation method; Step 22: Calculate the trapezoidal filter current slopes based on the optimized intermediate level V sn , the DC voltage source V dc and the output voltage V out to obtain the stage gradients α [k] , γ [k] , β [k] ; 1 ≤ k ≤ n, where n is an integer; Step 2-3: Select according to m based on the calculation method described in Step 2-1 [k] and the stage gradient α described in Step 2-2 [k] , γ [k] , β [k] Select the calculation method for generating a trapezoidal wave; The specific selection method for the calculation method of generating a trapezoidal wave in the 23rd step is: Step 231: Calculate the trapezoidal current generation calculation model within a single period by given charge and discharge times T1, T2 of the flying capacitor and the average current i in a single period set Calculate the trapezoidal current generation calculation model within a single period; Step 232: Since the dead time is much smaller than each switching mode time, the dead time is first ignored in the calculation process; Step 233: Although the charge amount of the flying capacitor charged and discharged in each period is obtained by calculation, it is difficult to keep equal within a sine period. Therefore, in order to maintain the voltage balance of the capacitors C q1 ~C q5 in the flying capacitor type power amplifier, the charging and discharging charges of the flying capacitors are kept equal within a single period, as shown in Equation (2): t1 represents the inductor current i Lf Starting from zero, switches S1…S6 are turned on, and S7…S 12 The duration of the off mode; t2 represents the switch tubes S7…S 12 conduct, and S1…S6 are turned off until the inductor current i Lf becomes zero and the duration size is maintained; t3 represents the inductor current i Lf Starting from zero, the switching transistors S7…S 12 conduct, and the duration of the off state of S1…S6; t4 represents the duration during which the switching transistors S1…S6 are turned on and S7…S 12 are turned off until the inductor current i Lf becomes zero; Step 234: Equation (2) is used to effectively ensure that the flying capacitor is maintained at a stable value in each period. However, since i1, i2, i3, and i4 are unknown at this time, additional conditions are required; Step 235: Assume that the seven-level soft-switching power amplifier outputs sinusoidal current and voltage, and the trapezoidal current equals a constant value after filtering, that is, the average current \(i\) required for a single period set , as shown in Equation (3), the product of the filtered average value of the trapezoidal current and the period is equal to the change in the amount of charge; i1 represents the inductor current i Lf Starting from zero, switches S1…S6 are turned on, and S7…S 12 The magnitude of the current at the end of the turn-off mode; i2 represents the inductor current i Lf When it is greater than zero, the switching transistors S1…S5, S7 are turned on, and S6, S8…S 12 The magnitude of the current at the end of the turn-off mode; i3 represents the current magnitude at the end of the conduction of switching transistors S7…S 12 and the turn-off of S1…S6; i4 represents the inductor current i Lf When it is less than zero, the switching transistors S1…S5, S7 are turned on, and S6, S8…S 12 The magnitude of the current at the end of the turn-off mode; Step 236: According to α, β, and γ in each time period in the filtering current generation calculation model, the relationships between i1, i2, i3, i4 and each time period are shown in Equations (4) and (5): i2 = i1 + γT, i3 = i4 - γT2 (5) Step 24: Under the joint action of establishing the upper and lower boundary constraint conditions of the trapezoidal filtering current and the capacitor voltage circulating current equalization method, the filtering current generation calculation model is obtained according to the calculation method of generating a trapezoidal wave selected in the 23rd step; Step 3: Obtain the calculated time t through calculation by the filtered current generation calculation model [k] and the filtered current i Lf[k] , and generate a PWM control signal with periodic variation through the calculated time t [k] and the filtered current i Lf[k] ; Step 4: Input the PWM control signal into the flying-capacitor seven-level soft-switching power amplifier to obtain the output voltage V out and the output current i out ; Step Five: Output current i out The output current set value i is obtained after being amplified by k times out * , and return to Step Two.

2. The control method of a flying capacitor seven-level soft-switching power amplifier according to claim 1, characterized in that The equation for calculating the slope of the trapezoidal filtering current in the 22nd step is as follows: L f represents a filter inductor.

3. The control method of a flying capacitor seven-level soft-switching power amplifier according to claim 1, characterized in that, The filtering current generation calculation model includes 10 switching modes, specifically as follows: Switching mode 1: During the time period from t0 to t1, switches S1…S6 are turned on, and S7…S 12 are turned off. The upper DC power supply V dc supplies energy to the load. The current flowing through inductor L f increases linearly with a slope of α. At this time, V sn = V dc ; Switching mode 2: During the time period from t1 to t2, the switching tubes S1…S5 are turned on and S6…S 12 are turned off. The upper DC power supply V dc charges the parallel capacitor of S7 reversely through the flying capacitor C q1 until the capacitance voltage of the parallel capacitor becomes zero; Switching mode 3: During the time period from t2 to t3, switches S1…S5, S7 are turned on, and S6, S8…S 12 are turned off. At this time, since the voltage of the parallel capacitor of S7 becomes zero, the current flows through the switching diode to the inductor L f with a slope of γ. At this time, V sn = 2 / 3V dc ; Switching mode 4: During the time period from t3 to t4, switch S7 is turned on, and switches S1…S6, S8…S 12 are turned off, and the inductor current i Lf charges the parallel capacitors of switches S8, S9, S 10 , S 11 , S 12 in reverse until the capacitor voltage of the parallel capacitors becomes zero; Switching mode 5: During the time period from t4 to t5, the switching transistors S7…S 12 conduct, the switching transistors S1…S6 turn off, and the inductor current i Lf flows through the freewheeling diodes of the switching transistors S7…S 12 and decreases continuously with a slope of β until the current finally becomes zero. At this time, V sn =-V dc ; Switching Mode 6: During the time period from t5 to t6, the state of the switch is the same as that in Mode 5, but at this time the inductor current reverses, and -V dc is applied to the inductor L f , and the inductor current increases linearly in the reverse direction with a slope of β. At this time, V sn = -V dc ; Switching mode 7: During the time period from t6 to t7, switch S7 is turned on, and switches S1...S6, S8...S are turned off. V is charged reversely through the flying capacitor C to the parallel capacitors of S1...S5 until the capacitor voltage of the parallel capacitors becomes zero. 12 is turned off, and V dc charges reversely through the flying capacitor C q1 to the parallel capacitors of S1...S5 until the capacitor voltage of the parallel capacitors becomes zero; Switching mode 8: During the time period from t7 to t8, the switching transistors S1…S5, S7 are conducting, and the switching transistors S6, S8…S 12 are turned off. At this time, since the voltage of the parallel capacitors of S1…S5 becomes zero, the current flows back to the power supply through the switching diodes with a slope of γ. At this time, V sn = 2 / 3V dc ; Switching mode 9: During the time period from t8 to t9, the switching transistors S1…S5 are turned on, and the switching transistors S6…S 12 are turned off. The current charges the parallel capacitor of S6 in the reverse direction until the capacitance voltage of the parallel capacitor becomes zero; Switching Mode 10: During the time period from t9 to t 10 the switching transistors S1…S6 are turned on, and the switching transistors S7…S 12 are turned off. The inductor current i Lf flows through the freewheeling diodes of the switching transistors S1…S6, and the inductor current decreases continuously with a slope of α. Eventually, the current becomes zero and returns to Switching Mode 1. At this time, V sn = V dc .

4. The control method of a flying-capacitor seven-level soft-switching power amplifier according to claim 1, characterized in that The specific selection method for the calculation method of generating a trapezoidal wave includes four types: First type: average current i of a single cycle set <0, V sn -V out > 0; Second type: average current i of a single cycle set > 0, V sn -V out > 0; Third type: average current i of a single cycle set <0, V sn -V out > 0; Fourth type: average current i of a single cycle set > 0, V sn -V out <0; According to the actual situation, select the above four types as the basis for subsequent simulation.

5. The control method of a flying-capacitor seven-level soft-switching power amplifier according to claim 1, characterized in that i set has both positive and negative values. Depending on the actual situation, the magnitudes of T1 and T2 are different; when i set > 0, T1 < T2; when i set < 0, T1 > T2.

6. The control method of a flying capacitor seven-level soft-switching power amplifier according to claim 1, characterized in that For the phase gradients α and β in the filtered current generation calculation model, select V dc and -V dc as the required distribution levels. For the levels V sn corresponding to the intermediate stage part of the trapezoidal filtered current, the selected levels depend on the magnitude of the output voltage V out .

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