Isolated DC Converter with High Gain and Wide Output Voltage Range and Its Control Method

Abstract

The present invention relates to an isolated DC converter with high gain and wide output voltage range and its control method. In this converter, one end of the equivalent primary leakage inductance with the same name terminals is connected in series with the primary side of the transformer, and the other end is connected to one ends of the first DC inductor, the third switch and the fourth switch. The first DC inductor is connected in series with the second switch. The second switch is respectively connected to one end of the first switch and the positive pole of the first DC power supply. The other end of the first switch is respectively connected to the third switch and the capacitor unit. The capacitor unit and the negative pole of the first DC power supply are connected to the other end of the fourth switch. The different name terminals of the primary side of the transformer are respectively connected to the fifth switch and the sixth switch. The fifth switch is connected to the other end of the third switch, and the sixth switch is connected to the other end of the fourth switch. The seventh switch and the eighth switch are connected in series. After being connected in series, one end is connected to the capacitor unit, and the other end is connected to the different name terminals of the primary side of the transformer. Compared with the prior art, the present invention has the advantages of reducing the switching loss of the converter and improving the conversion efficiency, etc.

Description

Technical Field

[0001] The present invention relates to the technical field of power electronic conversion, and particularly to an isolated DC converter with high gain and wide output voltage range and its control method. Background Technique

[0002] At present, high-gain wide-voltage-range DC converters have extensive and important applications in systems such as data centers, electric vehicles, and space power supplies. As an energy interaction interface between the high-voltage side and the low-voltage side, this DC converter is an indispensable part of the power conversion process. In recent years, the application of distributed generation and the popularization of electric vehicles have put forward new requirements for DC converters. The battery or supercapacitor charging of household electric vehicles and urban rail trains requires a wide range of output voltage changes. In distributed generation systems such as photovoltaic power generation or offshore wind power, due to the influence of natural conditions such as light and wind intensity on power generation, the voltage will fluctuate greatly. Therefore, it is of great significance to study the application of high-gain wide-output-voltage-range DC converters in the above power systems.

[0003] After retrieval, Chinese Patent CN117728690A discloses an isolated DC converter with wide voltage regulation range and low current ripple and its control method. The main circuit of the DC converter mainly consists of ten power switches, four capacitors, three inductors, two resonant elements, and two transformers, and can achieve high efficiency within a wide output voltage range and power range. However, this DC converter has two power channels. The main power channel adopts the topology of a dual-active-bridge DC converter, and the secondary power channel adopts the topology of a full-resonant DC converter. The control is complex, the power density is low, and all the switching tubes of this converter are hard-switched, with large switching losses. In addition, Chinese Patents CN117937936A and CN108718156A also have similar defects. Therefore, how to reduce switching losses and improve power conversion efficiency has become a problem to be solved in this field. Summary of the Invention

[0004] The purpose of the present invention is to provide an isolated DC converter with high gain and wide output voltage range and its control method to overcome the defects of large switching losses and low power density existing in the above-mentioned prior art.

[0005] The purpose of the present invention can be achieved by the following technical solutions:

[0006] According to a first aspect of the present invention, there is provided an isolated DC converter with high gain and wide output voltage range, comprising a first DC power supply, a first DC inductor, a first switch, a second switch, a third switch, a fourth switch, a fifth switch, a sixth switch, a seventh switch, an eighth switch, a capacitor unit, a transformer, an equivalent primary-side leakage inductance in series with the primary side of the transformer, and a secondary-side circuit of the transformer. The other end of the equivalent primary-side leakage inductance is respectively connected to the first DC inductor, one end of the third switch and one end of the fourth switch. The first DC inductor is connected in series with the second switch, and the second switch is respectively connected to one end of the first switch and the positive electrode of the first DC power supply. The other end of the first switch is respectively connected to the third switch and the capacitor unit. Both the capacitor unit and the negative electrode of the first DC power supply are connected to the other end of the fourth switch. The primary-side opposite ends of the transformer are respectively connected to the fifth switch and the sixth switch. The fifth switch is connected to the other end of the third switch, and the sixth switch is connected to the other end of the fourth switch. The seventh switch and the eighth switch are connected in series, and one end of the series connection is connected to the capacitor unit, and the other end is connected to the primary-side opposite ends of the transformer.

[0007] As a preferred technical solution, the capacitor unit includes a first capacitor and a second capacitor connected in series. The other end of the first capacitor is connected to the other end of the first switch, the other end of the second capacitor is connected to the other end of the fourth switch, and one end of the series connection of the seventh switch and the eighth switch is connected between the first capacitor and the second capacitor.

[0008] As a preferred technical solution, the secondary-side circuit of the transformer includes a second DC power supply, a ninth switch, a tenth switch, an eleventh switch, a twelfth switch, and a third capacitor. The primary-side same ends of the transformer are respectively connected to the ninth switch and the tenth switch, the primary-side opposite ends of the transformer are respectively connected to the eleventh switch and the twelfth switch. Both the ninth switch and the eleventh switch are connected to one end of the third capacitor, and both the tenth switch and the twelfth switch are connected to the other end of the third capacitor. The second DC power supply is connected in parallel across both ends of the third capacitor.

[0009] As a preferred technical solution, the first switch, the second switch, the third switch, the fourth switch, the fifth switch, the sixth switch, the seventh switch, the eighth switch, the ninth switch, the tenth switch, the eleventh switch, and the twelfth switch all include power switches with reverse-conduction characteristics.

[0010] As a preferred technical solution, the first switch includes a first switching transistor, a first diode, and a first parasitic capacitor. The first diode and the first parasitic capacitor are both anti-parallel to the first switching transistor. The positive electrode of the first diode is connected to the third switch, and the negative electrode is connected to the second switch.

[0011] As a preferred technical solution, the second switch includes a second switching transistor, a second diode, and a second parasitic capacitor. The second diode and the second parasitic capacitor are both anti-parallel to the second switching transistor. The positive electrode of the second diode is connected to the first DC inductor, and the negative electrode is connected to the first switch.

[0012] As a preferred technical solution, the turns ratio of the primary side to the secondary side of the transformer is N:1.

[0013] As a preferred technical solution, the calculation method of the turns ratio is as follows:

[0014]

[0015] In the formula, N 1 represents the turns ratio of the transformer when the voltage gain is less than 1, N 2 represents the turns ratio of the transformer when the voltage gain is greater than 1, P t is the output power of the converter, D ss represents the phase difference between the on-signals of the sixth switch and the ninth switch, D s0 is the phase difference between the on-signals of the ninth switch and the twelfth switch, D p0 represents the phase difference between the on-signals of the sixth switch and the third switch, L s represents the equivalent leakage inductance of the same-name terminals of the transformer, U 1 is the voltage of the first DC power supply, U 2 is the voltage of the second DC power supply, D p1 represents the duty cycle of the full-bridge ±U 1 level on the primary side of the converter, T h represents half of the switching period, and Tr(t) is the integral of the standard square-wave signal.

[0016] According to a second aspect of the present invention, there is provided a control method for the isolated DC converter described above, including the following steps: S1, based on a preselected transformer turns ratio, obtain an initial voltage gain; S2, compare the magnitude of the initial voltage gain with a preset threshold, and control the on / off states of the first switch and the second switch according to the comparison result; S3, obtain the current real-time voltage gain, and according to the real-time voltage gain, obtain a specific control parameter by using a preset duty ratio calculation method; S4, based on the specific control parameter, generate and send the current PI control instruction; S5, convert the current PI control instruction into a PWM drive signal; S6, based on the current PWM drive signal, control the on / off of the main switch tube, and return to S3, where the main switch tube includes the switches in the converter other than the first switch and the second switch; wherein, the specific control parameter includes the phase difference between the turn-on signals of the sixth switch and the third switch, the duty ratios of the positive and negative first DC power supply voltage levels on the primary side of the converter, the phase difference between the turn-on signals of the sixth switch and the ninth switch, and the phase difference between the turn-on signals of the ninth switch and the twelfth switch.

[0017] As a preferred technical solution, S2 specifically includes: when the voltage gain is less than 1, control the first switch to be closed and the second switch to be opened; when the voltage gain is greater than 1, control the first switch to be opened and the second switch to be closed.

[0018] Compared with the prior art, the present invention has the following beneficial effects:

[0019] 1. For the isolated DC converter provided by the present invention, the first switch and the second switch are provided, and it is possible to achieve a wide output voltage range by flexibly selecting the switch states of the first switch and the second switch without changing the topology structure, and at the same time, high conversion efficiency within a wide output voltage range can be ensured;

[0020] 2. In the isolated DC converter provided by the present invention, the third switch, the fourth switch, the fifth switch, the sixth switch, the seventh switch and the eighth switch can all achieve zero-voltage turn-on (i.e., ZVS turn-on), which can reduce switch losses and electromagnetic interference. At the same time, the ninth switch, the tenth switch, the eleventh switch and the twelfth switch can all achieve zero-current turn-off, which can reduce switch losses and electromagnetic interference;

[0021] 3. In the isolated DC converter of the present invention, the turns ratio of the primary and secondary sides of the transformer is N:1. By adjusting the turns ratio N of the transformer, the high voltage gain required by the converter can be obtained, thereby improving the applicability of the converter;

[0022] 4. Based on the proposed isolated DC converter, the present invention also provides a corresponding control method. This control method can, aiming at minimizing the effective value of the inductor current according to different working conditions, adopt voltage-loop PI control, and by selecting different duty-cycle calculation methods, obtain the minimum effective value of the inductor current of the converter, thereby reducing the conduction loss of the converter and improving the conversion efficiency.

[0023] 5. Through the provided control method of the isolated DC converter, based on selecting an appropriate transformer turns ratio according to the working ranges of the preset first DC voltage and the second DC voltage, this control method can determine the on-off states of the first switch and the second switch according to the magnitude of the initial voltage gain, control the current direction on the series equivalent leakage inductance with the same name ends on the primary side, and can achieve zero-voltage switching (i.e., ZVS switching) of the primary-side switching tube and zero-current switching (i.e., ZCS switching) of the secondary-side switching tube of the DC converter, which can further reduce the switching loss of the converter and improve the conversion efficiency. BRIEF DESCRIPTION OF THE DRAWINGS

[0024] Figure 1 is the circuit topology of the high-gain wide-output-voltage-range isolated DC converter in the embodiment of the present invention;

[0025] Figure 2 is the enlarged view of the primary-side circuit topology of the DC converter in the embodiment of the present invention;

[0026] Figure 3 is the enlarged view of the secondary-side circuit topology of the DC converter in the embodiment of the present invention;

[0027] Figure 4 is the voltage-current waveform diagram of the main components of the DC converter in the embodiment of the present invention;

[0028] Figure 5 is the circuit working state diagram of the DC converter in the embodiment of the present invention when M>1 and in working state a(t 0 -t 1 );

[0029] Figure 6 is the circuit working state diagram of the DC converter in the embodiment of the present invention when M>1 and in working state b(t 1 -t 2 );

[0030] Figure 7 is the circuit working state diagram of the DC converter in the embodiment of the present invention when M>1 and in working state c(t 2 -t 3 );

[0031] Figure 8 is the circuit working state diagram of the DC converter in the embodiment of the present invention when M>1 and in working state d(t 3 -t4 ) Circuit operating state diagram under

[0032] Figure 9 In the embodiment of the present invention, the DC converter is in M>1, operating state e(t 4 -t 5 ) Circuit operating state diagram under

[0033] Figure 10 In the embodiment of the present invention, the DC converter is in M>1, operating state f(t 5 -t 6 ) Circuit operating state diagram under

[0034] Figure 11 In the embodiment of the present invention, the DC converter is in M>1, operating state g(t 6 -t 7 ) Circuit operating state diagram under

[0035] Figure 12 In the embodiment of the present invention, the DC converter is in M>1, operating state h(t 7 -t 8 ) Circuit operating state diagram under

[0036] Figure 13 In the embodiment of the present invention, the DC converter is in M>1, operating state i(t 8 , t 9 ) Circuit operating state diagram under

[0037] Figure 14 Experimental control block diagram of the DC converter in the embodiment of the present invention;

[0038] Figure 15 Effect diagram of ZVS soft switching of the third switch 3 when the voltage gain M = 0.72, the input voltage Vin = 100V, the output voltage Vout = 24V, and the transmission power is 168W in the embodiment of the present invention;

[0039] Figure 16 Effect diagram of ZCS soft switching of the ninth switch 9 when the voltage gain M = 0.72, the input voltage Vin = 100V, the output voltage Vout = 24V, and the transmission power is 168W;

[0040] Figure 17 Effect diagram of ZVS soft switching of the third switch 3 when the voltage gain M = 1.65, the input voltage Vin = 100V, the output voltage Vout = 55V, and the transmission power is 275W;

[0041] Figure 18When the voltage gain M = 1.65, the input voltage Vin = 100V, the output voltage Vout = 55V, and the transmission power is 275W, the effect diagram of the ZCS soft switching of the ninth switch 9;

[0042] Where: 1. The first switch; 2. The second switch; 3. The third switch; 4. The fourth switch; 5. The fifth switch; 6. The sixth switch; 7. The seventh switch; 8. The eighth switch; 9. The ninth switch; 10. The tenth switch; 11. The eleventh switch; 12. The twelfth switch. Detailed implementation manners

[0043] The present invention will be described in detail below with reference to the accompanying drawings and specific embodiments. This embodiment is implemented on the premise of the technical solution of the present invention, and gives detailed implementation manners and specific operation processes, but the protection scope of the present invention is not limited to the following embodiments.

[0044] Embodiment

[0045] As Figure 1 shown, this embodiment provides an isolated DC converter with high gain and wide output voltage range. It includes a transformer T, an equivalent homonymous terminal leakage inductance L connected in series in the primary side of the transformer s , the primary circuit of the transformer T, and the secondary circuit of the transformer T. As Figure 2 shown, the primary circuit of the transformer T specifically includes: a first DC power supply U 1 , a first DC inductor L, a first switch 1, a second switch 2, a third switch 3, a fourth switch 4, a fifth switch 5, a sixth switch 6, a seventh switch 7, an eighth switch 8, and a capacitor unit. Among them, the capacitor unit includes a first capacitor C 1 connected in series with a second capacitor C 2 . As Figure 3 shown, the secondary circuit of the transformer T specifically includes: a second DC power supply U 2 , a ninth switch 9, a tenth switch 10, an eleventh switch 11, a twelfth switch 12, and a third capacitor C 3 .

[0046] The connection relationships of each component are as follows:

[0047] One end of the equivalent homonymous terminal leakage inductance L s is respectively connected to one end of the first DC inductor L, one end of the third switch 3, and one end of the fourth switch 4. The other end of the first DC inductor L is connected in series with one end of the second switch 2. The other end of the second switch 2 is respectively connected to one end of the first switch 1 and the positive electrode of the first DC power supply U 1 . The other end of the first switch 1 is respectively connected to the other end of the third switch 3 and the other end of the first capacitor C 1 . The other end of the second capacitor C 2 is connected to the first DC power supply U1 The negative electrodes are all connected to the other end of the fourth switch 4. The different-named ends of the primary side of the transformer T are respectively connected to one end of the fifth switch 5 and one end of the sixth switch 6. The other end of the fifth switch 5 is connected to the other end of the third switch 3. The other end of the sixth switch 6 is connected to the other end of the fourth switch 4. The seventh switch 7 and the eighth switch 8 are connected in series, and one end after the series connection is connected to the first capacitor C 1 and the second capacitor C 2 in between, and the other end is connected to the different-named ends of the primary side of the transformer T.

[0048] The same-named ends of the secondary side of the transformer T are respectively connected to one end of the ninth switch 9 and one end of the tenth switch 10. The different-named ends of the secondary side of the transformer T are respectively connected to one end of the eleventh switch 11 and one end of the twelfth switch 12. The other end of the ninth switch 9 and the other end of the eleventh switch 11 are both connected to one end of the third capacitor C 3 The other end of the tenth switch 10 and the other end of the twelfth switch 12 are both connected to the other end of the third capacitor C 3 The second DC power supply U 2 is connected in parallel across both ends of the third capacitor C 3 Its positive pole is connected to the other end of the eleventh switch 11, and its negative pole is connected to the other end of the twelfth switch 12.

[0049] Among them, the first switch 1, the second switch 2, the third switch 3, the fourth switch 4, the fifth switch 5, the sixth switch 6, the seventh switch 7, the eighth switch 8, the ninth switch 9, the tenth switch 10, the eleventh switch 11, and the twelfth switch 12 are all power switches with reverse-conduction characteristics. Specifically:

[0050] The first switch 1 includes the first switching tube Q -sa and its anti-parallel first diode D -sa and the first parasitic capacitor C -sa , the positive pole of the first diode D -sa is connected to the third switch 3, and the negative pole is connected to the second switch 2.

[0051] The second switch 2 includes the second switching tube Q -sb and its anti-parallel second diode D -sb and the second parasitic capacitor C -sb , the positive pole of the second diode D -sb is connected to the first DC inductor L, and the negative pole is connected to the first switch 1.

[0052] The third switch 3 includes the third switching tube Q -s1 and its anti-parallel third diode D -s1 and the third parasitic capacitor C -s1 , the positive pole of the third diode D -s1 is connected to the equivalent same-named end leakage inductance Ls , the negative electrodes are respectively connected to the first switch 1 and the fifth switch 5.

[0053] The fourth switch 4 includes a fourth switching transistor Q -s2 and its anti-parallel fourth diode D -s2 and a fourth parasitic capacitor C -s2 , the positive electrode of the fourth diode D -s2 is respectively connected to the negative electrode of the first DC power supply U 1 and the sixth switch 6, and the negative electrode is connected to the equivalent leakage inductance L of the same name terminal s .

[0054] The fifth switch 5 includes a fifth switching transistor Q -s3 and its anti-parallel fifth diode D -s3 and a fifth parasitic capacitor C -s3 , the positive electrode of the fifth diode D -s3 is connected to the opposite name terminal of the primary side of the transformer T, and the negative electrode is connected to the third switch 3.

[0055] The sixth switch 6 includes a sixth switching transistor Q -s4 and its anti-parallel sixth diode D -s4 and a sixth parasitic capacitor C -s4 , the positive electrode of the sixth diode D -s4 is connected to the opposite name terminal of the primary side of the transformer T, and the negative electrode is connected to the fourth switch 4.

[0056] The seventh switch 7 includes a seventh switching transistor Q -s5 and its anti-parallel seventh diode D -s5 and a seventh parasitic capacitor C -s5 , the positive electrode of the seventh diode D -s5 is connected to the eighth switch 8, and the negative electrode is connected to the opposite name terminal of the primary side of the transformer T.

[0057] The eighth switch 8 includes an eighth switching transistor Q -s6 and its anti-parallel eighth diode D -s6 and an eighth parasitic capacitor C -s6 , the positive electrode of the eighth diode D -s6 is connected to the seventh switch 7, and the negative electrode is connected between the first capacitor C 1 and the second capacitor C 2 .

[0058] The ninth switch 9 includes a ninth switching transistor Q -1 and its anti-parallel ninth diode D -1 and a ninth parasitic capacitor C -1 , the positive electrode of the ninth diode D -1 is connected to the same name terminal of the secondary side of the transformer T, and the negative electrode is connected to the third capacitor C 3One end of which, that is, the second DC power supply U 2 The positive electrode of.

[0059] The tenth switch 10 includes a tenth switching transistor Q -2 And its anti-parallel twelfth diode D -2 And the tenth parasitic capacitor C -2 , The negative electrode of the twelfth diode D -2 Is connected to the same-named terminal of the secondary side of the transformer T, and the positive electrode is connected to the other end of the third capacitor C 3 One end of which, that is, the second DC power supply U 2 The negative electrode of.

[0060] The eleventh switch 11 includes an eleventh switching transistor Q -3 And its anti-parallel eleventh diode D -3 And the eleventh parasitic capacitor C -3 , The positive electrode of the eleventh diode D -3 Is connected to the different-named terminal of the secondary side of the transformer T, and the negative electrode is connected to one end of the third capacitor C 3 One end of which, that is, the second DC power supply U 2 The positive electrode of.

[0061] The twelfth switch 12 includes a twelfth switching transistor Q -4 And its anti-parallel twelfth diode D -4 And the twelfth parasitic capacitor C -4 , The negative electrode of the twelfth diode D -4 Is connected to the different-named terminal of the secondary side of the transformer T, and the positive electrode is connected to the other end of the third capacitor C 3 One end of which, that is, the second DC power supply U 2 The negative electrode of.

[0062] In this embodiment, the turn ratio of the primary side to the secondary side of the transformer T is N:1. By adjusting the turn ratio N of the transformer T, the high voltage gain required by the converter can be obtained, thereby improving the applicability of the converter. Among them, when the voltage gain M < 1 (NU 2 / U 1 ), The transformer turn ratio N 1 As shown in Equation (1):

[0063]

[0064] When the voltage gain M > 1 (NU 2 / U 1 ), The transformer turn ratio N 2 As shown in Equation (2):

[0065]

[0066] In the formula: T h = 1 / (2fs ), P t is the output power of the converter, D ss represents the phase difference of the turn-on signals of the sixth switch and the ninth switch, D s0 is the phase difference of the turn-on signals of the ninth switch and the twelfth switch, D p0 represents the phase difference of the turn-on signals of the sixth switch and the third switch, D p1 represents the duty cycle of the full-bridge ±U 1 level on the primary side of the converter, T h represents half of a switching period, f s represents the switching frequency, and Tr(t) is the integral of the standard square wave.

[0067] Furthermore, this embodiment also provides a control method for controlling the aforementioned isolated DC converter to achieve power conversion. This control method aims to minimize the effective value of the inductor current, adopts voltage-loop PI control, and by selecting different duty cycle calculation methods, according to the preset U 1 and U 2 operating range, select an appropriate transformer turns ratio, obtain the initial voltage gain after determining the transformer turns ratio, and determine the on-off states of the first switch and the second switch according to the initial voltage gain, and control the current direction on the series equivalent leakage inductance L s of the primary side, so as to realize zero-voltage turn-on (i.e., ZVS turn-on) of the primary-side switch tube of the DC converter and zero-current turn-off (i.e., ZCS turn-off) of the secondary-side switch tube, which can further reduce the switching loss of the converter and improve the conversion efficiency. The specific steps include:

[0068] Step S1, based on the preselected transformer turns ratio, obtain the initial voltage gain. In practical applications, the transformer turns ratio is jointly determined by the required input voltage range and output voltage range during actual operation.

[0069] Step S2, determine whether the initial voltage gain is less than 1, and control the on-off states of the first switch 1 and the second switch 2 according to the comparison result. Specifically:

[0070] When the voltage gain M < 1 (NU 2 / U 1 ), control the first switch 1 to be normally closed and the second switch 2 to be normally open. At this time, high conversion efficiency within a small range of output voltage is achieved. The circuit situation of the converter at this time is as follows:

[0071] The first DC power supply U 1 forms a loop with the first switch 1, the third switch 3, the primary side of the transformer T, and the sixth switch 6; the first DC power supply U 1 forms a loop with the first switch 1, the fifth switch 5, the primary side of the transformer T, and the fourth switch 4; the first DC power supply U 1With the first switch 1 and the first capacitor C 1 , the eighth switch 8, the seventh switch 7, the primary side of the transformer T, and the fourth switch 4 form a loop; the first DC power supply U 1 is connected to the first switch 1, the third switch 3, the primary side of the transformer T, the seventh switch 7, the eighth switch 8, and the second capacitor C 2 to form a loop; the secondary side of the transformer T is connected to the ninth switch 9, the third capacitor C 3 , and the twelfth switch 12 to form a loop; the secondary side of the transformer T is connected to the tenth switch 10, the third capacitor C 3 , and the eleventh switch 11 to form a loop; the second DC power supply U 2 is connected to the third capacitor C 3 to form a loop.

[0072] When the voltage gain M > 1 (NU 2 / U 1 ), control the first switch 1 to be normally open and the second switch 2 to be normally closed. At this time, both the output voltage range is broadened and a high conversion efficiency within a wide output voltage range is achieved. The circuit conditions of the converter at this time are as follows:

[0073] The first DC power supply U 1 is connected to the second switch 2, the first DC inductor L, and the fourth switch 4 to form a loop; the fourth switch 4 is connected to the sixth switch 6 and the primary side of the transformer T to form a loop; the first DC power supply U 1 is connected to the second switch 2, the first DC inductor L, the third switch 3, the first capacitor C 1 , and the second capacitor C 2 to form a loop; the first capacitor C 1 is connected to the second capacitor C 2 , the sixth switch 6, the primary side of the transformer T, the third switch 3 to form a loop; the first capacitor C 1 is connected to the third switch 3, the primary side of the transformer T, the seventh switch 7, and the eighth switch 8 to form a loop; the first capacitor C 1 is connected to the fifth switch 5, the primary side of the transformer T, the fourth switch 4, and the second capacitor C 2 to form a loop; the second capacitor C 2 is connected to the eighth switch 8, the seventh switch 7, the primary side of the transformer T, and the fourth switch 4 to form a loop; the secondary side of the transformer T is connected to the ninth switch 9, the third capacitor C 3 , and the twelfth switch 12 to form a loop; the secondary side of the transformer T is connected to the tenth switch 10, the third capacitor C 3 , and the eleventh switch 11 to form a loop; the second DC power supply U 2 is connected to the third capacitor C 3Form a loop; the third switch 3 is connected in series with the fourth switch 4; the fifth switch 5 is connected in series with the sixth switch 6; the ninth switch 9 is connected in series with the tenth switch 10; the eleventh switch 11 is connected in series with the twelfth switch 12.

[0074] Step S3: Obtain the current real-time voltage gain, and based on this real-time voltage gain, use a preset duty cycle calculation method to obtain specific control parameters. Among them, the specific control parameters include the conduction signal phase difference D between the sixth switch 6 and the third switch 3 p0 , the duty cycle D of the full-bridge ±U 1 level on the primary side of the converter p1 , the conduction signal phase difference D between the sixth switch 6 and the ninth switch 9 ss and the conduction signal phase difference D between the ninth switch 9 and the twelfth switch 12 s0 .

[0075] Among them, the preset duty cycle calculation method is implemented according to the calculation process given in Table 1. The duty cycle calculation results obtained with the goal of minimizing the effective value of the inductor current for the DC converter provided in this embodiment are also shown in Table 1. In the table, ε represents the soft-switching margin factor to ensure the realization of soft-switching technology.

[0076] Table 1 Duty cycle calculation results of the converter under different working conditions

[0077]

[0078]

[0079]

[0080] Step S4: Based on the calculated various specific control parameters, generate and send the current PI control instruction.

[0081] Specifically, obtain the duty cycles of the third switch 3 to the twelfth switch 12 at the current moment according to the various specific control parameters, and then obtain the actual output voltage at the current moment. According to the deviation between the preset target output voltage and the actual output voltage at the current moment, the PI controller generates control instructions for each main switch tube and sends them to the PWM drive unit;

[0082] Steps S5 to S6: The PWM drive unit converts the received current PI control instruction into a PWM drive signal, and controls the on and off of each main switch tube according to this PWM drive signal to achieve power conversion control of the converter. Each main switch tube is the third switch 3 to the twelfth switch 12. Then return to step S3 and repeat the execution in sequence until the actual output voltage tends to be stable (approaches the target output voltage).

[0083] Among them, the relationship between each specific control parameter and the duty cycle of the third switch 3 to the twelfth switch 12 is specifically as follows: The duty cycles D of the third switch 3, the fourth switch 4, the ninth switch 9, the tenth switch 10, the eleventh switch 11, and the twelfth switch 12 are the same and equal to 0.5; The duty cycles D of the fifth switch 5 and the seventh switch 7 are the same and equal to (D p0 +D p1 ); The duty cycles D of the sixth switch 6 and the eighth switch 7 are the same and equal to (D p0 +D p1 ). The first switch 1 and the second switch 2, the third switch 3 and the fourth switch 4, the fifth switch 5 and the seventh switch 7, the sixth switch 6 and the eighth switch 8, the ninth switch 9 and the tenth switch 10, the eleventh switch 11 and the twelfth switch 12 are respectively complementary-conducted, and the conduction signal phases of the fifth switch 5 and the sixth switch 6, the seventh switch 7 and the eighth switch 8 differ by 180°, and the phase difference between the turn-on signals of the sixth switch 6 and the third switch 3 is the phase-shift angle D p0 and is between 0° and 180°, and the phase difference between the turn-on signals of the sixth switch 6 and the ninth switch 9 is the phase-shift angle D ss and is between 0° and 180°, and the phase difference between the turn-on signals of the ninth switch 9 and the twelfth switch 12 is the phase-shift angle D s0 and is between 0° and 180°.

[0084] Figure 4 shows the main element waveform diagram of the DC converter provided in this embodiment when operating stably under M>1. Next, in combination with Figures 5 to 13 the working state of the DC converter under M>1 will be analyzed in detail, Figures 5 to 13 is the circuit working state diagram of the implementation process of the DC converter within half a switching period when M>1 (one switching period includes 18 working processes). Considering the symmetry of the working process, only the working process of the first half of the working cycle (t 0 -t 9 ) will be analyzed:

[0085] Working state a (t 0 -t 1 ) is as Figure 5 shown: Before t 0 , the fourth switch tube Q -s2 , the seventh switch tube Q -s5 , the tenth switch tube Q -2 and the eleventh switch tube Q -3 are in the conducting state, and the sixth switch tube Q -s4 is in the off state. At the moment of t 0 , the sixth switch tube Q -s4 is in the conducting state, as long as it is ensured that i Ls (t 0) < 0, that is, before the sixth switching transistor Q - s4 is turned on, current flows through the body diode D of the sixth switching transistor Q -s4 to ensure zero - voltage turn - on of the sixth switching transistor Q -s4 . During this stage, the converter v -s4 = 0, v p = -V s . The current i out through the equivalent series primary - side leakage inductance L s of the transformer starts to increase at a constant slope, obtaining the following relationship: Ls where v

[0086]

[0087] is the AC voltage on the primary side of the transformer of the converter, v p is the AC voltage on the secondary side of the transformer of the converter, and V s is the output voltage of the converter. out

[0088] Operating state b(t 1 - t 2 ) is as shown in Figure 6 : At time t 1 , the fourth switching transistor Q -s2 turns off. At this time, the current on the primary side of the transformer T starts to discharge the junction capacitance of the third switching transistor Q -s1 and charge the junction capacitor C -s2 of the fourth switching transistor Q -s2 . Through reasonable design, when the drain - source voltage of the fourth switching transistor Q -s2 is charged to V a , the drain - source voltage of the third switching transistor Q -s1 is zero, and the body diode D -s1 of the third switching transistor Q -s1 will conduct, thus creating ZVS turn - on conditions (zero - voltage switching) for the third switching transistor Q -s1 , as follows:

[0089] i Ls (t 2 ) < 0 (4)

[0090] Operating state c(t 2 - t 3 ) is as shown in Figure 7 : At t 2 , the third switching transistor Q -s1 is turned on by ZVS. During this stage, the converter v p = V a , v s = -V out . The equivalent series primary - side leakage inductance L​s The current i on Ls Starts to increase at a constant slope, obtaining the following relationship:

[0091]

[0092] The working state d(t 3 -t 4 ) is as Figure 8 shown: At time t 3 , the tenth switching transistor Q -2 turns off. The current on the secondary side of the transformer T flows through the body diode D -1 of the ninth switching transistor Q -1 . As long as it is ensured that the current i s on the equivalent series primary-side leakage inductance L Ls (t 3 ) = 0, zero-current turn-off of the tenth switching transistor Q -2 can be achieved.

[0093] The working state e(t 4 -t 5 ) is as Figure 9 shown: At t 4 , the ninth switching transistor Q -1 is turned on. In this stage, the converter v p = V a , v s = 0. The current i s on the equivalent series primary-side leakage inductance L Ls starts to increase at a constant slope, obtaining the following relationship:

[0094]

[0095] The working state f(t 5 -t 6 ) is as Figure 10 shown: At time t 5 , the eleventh switching transistor Q -3 turns off. The current on the secondary side of the transformer T flows through the body diode D -4 of the twelfth switching transistor Q -4 . As long as it is ensured that i Ls (t 5 ) = 0, zero-current turn-off of the eleventh switching transistor Q -3 can be achieved.

[0096] The working state g(t 6 -t 7 ) is as Figure 11 shown: At t 6 , the twelfth switching transistor Q -4 is turned on. In this stage, the converter vp = V a , v s = V out . The current i on the equivalent primary series leakage inductance L of the same name terminals s starts to increase at a constant slope, and the following relationship is obtained: Ls The operating state h(t

[0097]

[0098] - t 7 - t 8 ) is as Figure 12 shown: At t 7 , the sixth switching transistor Q -s4 turns off. At this time, the primary side current of the transformer T starts to discharge the junction capacitors C -s3 and C -s6 of the fifth switching transistor Q -s3 and the eighth switching transistor Q -s6 , and charges the junction capacitor C -s4 of the sixth switching transistor Q -s4 . After reasonable design, after the charging process ends, the drain-source voltages of the fifth switching transistor Q -s3 and the sixth switching transistor Q -s4 are both V a / 2, the drain-source voltage of the eighth switching transistor Q -s6 reaches zero, and the body diode D -s6 turns on, creating ZVS conditions for the eighth switching transistor Q -s6 to turn on in the next stage. If the charging energy stored in these capacitors is ignored, the ZVS turn-on condition of the eighth switching transistor Q -s6 is as follows:

[0099] i Ls (t 8 ) < 0 (8)

[0100] The operating state i(t 8 , t 9 ) is as Figure 13 shown: At t 8 , the eighth switching transistor Q -s6 is turned on by ZVS. In this stage, the converter v p = V a / 2, v s = V out . The current i on the equivalent primary series leakage inductance L of the same name terminals s starts to increase at a constant slope, and the following relationship is obtained: Ls The following relationships are obtained:

[0101]

[0102] Since the operating state when M < 1 is similar to that when M > 1, it will not be described in detail.

[0103] This embodiment is verified by an experimental method, and the verification environment is as follows: the equivalent leakage inductance L s is 38.68 μH, the inductance L is 4 mH, and the clamping capacitors C 1 , C 2 and C 3 are all 470 μF, the switching frequency f s is 50 kHz, and the turns ratio N of the built-in high-frequency transformer is 3:1. When the voltage gain M < 1 (NU 2 / U 1 ), the first switch 1 is normally open and the second switch 2 is normally closed; when the voltage gain M > 1 (NU 2 / U 1 ), the first switch 1 is normally closed and the second switch 2 is normally open.

[0104] Figure 14 Fig. shows the experimental control block diagram of the DC converter provided in this embodiment, and the applicable power under different working conditions is shown as follows:

[0105]

[0106] This embodiment verifies the converter and the control strategy through an experimental method, and respectively verifies the transmission efficiency of the converter when the input voltage Vin = 100 V and the output voltages Vout = 15 V, 24 V, 48 V, and 55 V. The results are shown in Table 2.

[0107] Table 2 Transmission efficiency of the converter under different working conditions

[0108]

[0109] Figure 15 For the voltage gain M = 0.72, the input voltage V in = 100 V, the output voltage V out = 24 V, and the transmission power is 168 W, the effect diagram of the ZVS soft switching of the third switch 3; Figure 16 For the voltage gain M = 0.72, the input voltage V in = 100 V, the output voltage V out = 24 V, and the transmission power is 168 W, the effect diagram of the ZCS soft switching of the ninth switch 9; Figure 17 For the voltage gain M = 1.44, the input voltage V in = 100 V, the output voltage V out = 48 V, and the transmission power is 240 W, the effect diagram of the ZVS soft switching of the third switch 3; Figure 18 For the voltage gain M = 1.44, the input voltage Vin = 100V, output voltage V out = 48V, when the transmission power is 240W, the effect diagram of the ZCS soft switching of the ninth switch 9. Combining Table 2, it can be seen that at the input voltage V in = 100V, output voltage V out = 15V, 24V, 48V, 55V, the DC converter has a conversion efficiency of more than 90%, that is, the DC converter has a high conversion efficiency within a wide voltage output range. Figures 15 to 18 All can achieve the purpose of eliminating the overlapping area of the switch tube current and voltage, that is, to achieve the effect of ZVS turn-on of the primary switch tube of the DC converter and ZCS turn-off of the secondary switch tube.

[0110] The preferred specific embodiments of the present invention have been described in detail above. It should be understood that those of ordinary skill in the art can make many modifications and variations based on the concept of the present invention without creative work. Therefore, all technical solutions that can be obtained by those skilled in the art in the technical field of the present invention through logical analysis, reasoning or limited experiments based on the concept of the present invention on the basis of the prior art should be within the protection scope determined by the claims.

Claims

1. A high-gain, wide-output voltage range isolated DC converter, characterized in that: The invention comprises a first DC power supply, a first DC inductor, a first switch, a second switch, a third switch, a fourth switch, a fifth switch, a sixth switch, a seventh switch, an eighth switch, a capacitor unit, a transformer, an equivalent same-name leakage inductor connected in series with the primary side of the transformer, and a secondary circuit of the transformer, wherein the other end of the equivalent same-name leakage inductor is respectively connected to one end of the first DC inductor, the third switch and the fourth switch, the first DC inductor is connected in series with one end of the second switch, the other end of the second switch is respectively connected to one end of the first switch and the positive electrode of the first DC power supply, the other end of the first switch is connected to the other end of the third switch, the negative electrode of the first DC power supply is connected to the other end of the fourth switch, the primary opposite-name terminals of the transformer are respectively connected to one end of the fifth switch and the sixth switch, the other end of the fifth switch is connected to the other end of the third switch, the other end of the sixth switch is connected to the other end of the fourth switch, the seventh switch is connected in series with the eighth switch, and one end of the series connection is connected to the primary opposite-name terminal of the transformer; The capacitor unit includes a first capacitor and a second capacitor connected in series, the other end of the first capacitor is connected to the other end of the first switch, the other end of the second capacitor is connected to the other end of the fourth switch, and the other end of the seventh switch and the eighth switch connected in series is connected between the first capacitor and the second capacitor.

2. The high-gain, wide-output voltage range isolated DC converter according to claim 1, characterized in that: The secondary circuit of the transformer includes a second DC power supply, a ninth switch, a tenth switch, an eleventh switch, a twelfth switch and a third capacitor. The secondary like-named ends of the transformer are respectively connected to the ninth switch and the tenth switch, the secondary unlike-named ends of the transformer are respectively connected to the eleventh switch and the twelfth switch, the ninth switch and the eleventh switch are both connected to one end of the third capacitor, the tenth switch and the twelfth switch are both connected to the other end of the third capacitor, and the second DC power supply is connected in parallel to both ends of the third capacitor.

3. The high-gain, wide-output voltage range isolated DC converter according to claim 2, characterized in that: The first switch, the second switch, the third switch, the fourth switch, the fifth switch, the sixth switch, the seventh switch, the eighth switch, the ninth switch, the tenth switch, the eleventh switch, and the twelfth switch all include power switches with reverse conduction characteristics.

4. The high-gain, wide-output voltage range isolated DC converter according to claim 1, characterized in that: The first switch includes a first switch tube, a first diode and a first parasitic capacitor. The first diode and the first parasitic capacitor are both anti-parallel connected to the first switch tube. The positive electrode of the first diode is connected to the other end of the third switch, and the negative electrode is connected to the other end of the second switch.

5. The high-gain, wide-output voltage range isolated DC converter according to claim 4, characterized in that: The second switch includes a second switch tube, a second diode and a second parasitic capacitor. The second diode and the second parasitic capacitor are both anti-parallel connected to the second switch tube. The positive electrode of the second diode is connected to the first DC inductor, and the negative electrode is connected to one end of the first switch.

6. The high-gain, wide-output voltage range isolated DC converter according to claim 2, characterized in that: The turns ratio between the primary side and the secondary side of the transformer is N:

1.

7. The high-gain, wide-output voltage range isolated DC converter according to claim 6, characterized in that: The calculation method of the turns ratio is: In the formula, It represents the transformer turns ratio when the voltage gain is less than 1. It represents the transformer turns ratio when the voltage gain is greater than 1. is the converter output power, represents the phase difference between the turn-on signals of the sixth switch and the ninth switch, is the phase difference between the turn-on signals of the ninth switch and the twelfth switch, represents the phase difference between the turn-on signals of the sixth switch and the third switch, represents the equivalent leakage inductance of the transformer, is the first DC power supply voltage, is the second DC power supply voltage, Represents the primary full bridge of the converter The duty cycle of the level, represents half of a switching cycle, is the integral of the standard square wave signal.

8. A control method for the isolated DC converter according to any one of claims 2 to 7, characterized in that: The following steps are involved: S1, based on the preselected transformer turns ratio, obtain the initial voltage gain; S2, comparing the initial voltage gain with a preset threshold, and controlling the on / off status of the first switch and the second switch according to the comparison result; S3, obtaining the current real-time voltage gain, and obtaining a specific control parameter according to the real-time voltage gain using a preset duty cycle calculation method; S4, generating and sending a current PI control instruction based on the specific control parameter; S5, converting the current PI control instruction into a PWM drive signal; S6, based on the current PWM drive signal, controlling the on and off of the main switch tube, and returning to S3, the main switch tube includes switches in the converter except the first switch and the second switch; Among them, the specific control parameters include the phase difference of the turn-on signals of the sixth switch and the third switch, the duty cycle of the positive and negative first DC power supply voltage levels of the primary full bridge of the converter, the phase difference of the turn-on signals of the sixth switch and the ninth switch, and the phase difference of the turn-on signals of the ninth switch and the twelfth switch.

9. The control method of the isolated DC converter according to claim 8, characterized in that: The S2 specifically includes: When the voltage gain is less than 1, controlling the first switch to be closed and the second switch to be opened; When the voltage gain is greater than 1, the first switch is controlled to be opened and the second switch is controlled to be closed.

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