High-precision power amplifier based on resonance auxiliary commutation technology and parameter design method

Abstract

The invention discloses a high-precision power amplifier based on a resonance auxiliary commutation technology and a parameter design method, and belongs to the technical field of power amplifiers. The invention aims to solve the problems of current zero-crossing distortion and dead-zone effect of the existing switching power amplifier modulated by PWM (Pulse-Width Modulation). The power amplifier comprises a capacitor C1, a capacitor C2, four MOSFET switch tubes S1, S2, S3 and S4, an auxiliary inductor Ld1, an auxiliary inductor Ld2, a filter inductor Lo and a filter capacitor Co, wherein the switch tubes S1, S2, S3 and S4 form two bridge arms which are connected in parallel with a branch circuit formed by the capacitor C1 and the capacitor C2; an auxiliary inductor Ld1 and an auxiliary inductor Ld2 are connected in series between the two bridge arms; a connection point between the auxiliary inductor Ld1 and the auxiliary inductor Ld2 is used as a point C, and a filter inductor Lo and a filter capacitor Co are sequentially connected in series between the point C and the connection point of the capacitor C1 and the capacitor C2; and the load R is connected in parallel with the filter capacitor Co. According to the invention, the full-load range soft switching of the power amplifier can be realized.

Description

Technical Field

[0001] The invention relates to a high-precision power amplifier based on a resonance-assisted commutation technology and a parameter design method, and belongs to the technical field of power amplifiers. Background Art

[0002] The high-precision motion platform in semiconductor manufacturing equipment needs to achieve μm or even nm-level positioning accuracy, so the motor on the motion platform needs to be equipped with a high-precision power amplifier. In order to prevent the expansion and deformation of precision components due to heat, the temperature rise of the power amplifier must be limited. Traditional linear power amplifiers have high precision, but the loss is as high as 40%, resulting in serious heat dissipation, which is not suitable for high-precision motion platforms. Although the loss of hybrid power amplifiers is lower than that of linear power amplifiers, it still cannot meet the low heat dissipation requirements required by semiconductor manufacturing equipment. In addition, the hybrid structure of linear power amplifiers and switching power amplifiers increases the complexity of the circuit, and the series or parallel structure of the two may also have small signal stability problems, so hybrid power amplifiers are not recommended. The loss of switching power amplifiers can be as low as 5%, and the heat dissipation is much lower than that of linear power amplifiers and hybrid power amplifiers. It is the only suitable choice.

[0003] For switching amplifiers using PWM modulation, the disadvantages are that hard switching leads to large device losses, severe heat dissipation, current zero-crossing distortion and dead zone effect, and high output current distortion. However, its advantages are fixed-frequency switching and insensitivity to control system delays. For switching amplifiers using hysteresis control, the disadvantage is variable-frequency switching, which is sensitive to control system delays. However, its advantages are soft switching over the full load range, low device heat dissipation, no current zero-crossing distortion and dead zone effect, and low output current distortion. Summary of the invention

[0004] Aiming at the problems of current zero-crossing distortion and dead zone effect in the existing PWM modulated switching power amplifier, the present invention provides a high-precision power amplifier based on resonance-assisted commutation technology and a parameter design method.

[0005] The invention discloses a high-precision power amplifier based on resonance auxiliary commutation technology, comprising a capacitor C1, a capacitor C2, four MOSFET switch tubes S1, S2, S3, S4, an auxiliary inductor L d1 , auxiliary inductor L d2 , filter inductor L o And filter capacitor C o ,

[0006] Capacitor C1 and capacitor C2 are connected in series between the positive electrode of the power supply and the negative electrode of the power supply;

[0007] The positive electrode of the power supply is connected to the drain of the switch tube S1, the source of the switch tube S1 is connected to the drain of the switch tube S2, and the source of the switch tube S2 is connected to the negative electrode of the power supply; the positive electrode of the power supply is also connected to the drain of the switch tube S3, the source of the switch tube S3 is connected to the drain of the switch tube S4, and the source of the switch tube S4 is connected to the negative electrode of the power supply;

[0008] The connection point between the source of switch tube S1 and the drain of switch tube S2 is taken as point A, the connection point between the source of switch tube S3 and the drain of switch tube S4 is taken as point B, and an auxiliary inductor L is connected in series between point A and point B. d1 and auxiliary inductor L d2 ; Set the auxiliary inductor L d1 and auxiliary inductor L d2 The connection point between them is taken as point C, and the filter inductor L is connected in series between point C and the connection point of capacitor C1 and capacitor C2. o And filter capacitor C o ; Load R and filter capacitor C o in parallel.

[0009] According to the high-precision power amplifier based on the resonance auxiliary commutation technology of the present invention, the auxiliary inductor L d1 and auxiliary inductor L d2 The inductance values ​​are equal.

[0010] According to the high-precision power amplifier based on the resonance auxiliary commutation technology of the present invention, the auxiliary inductor L d1 and auxiliary inductor L d2 The two half-bridge arms formed by the switch tubes S1 to S4 are coupled so that the two half-bridge arms serve as resonant auxiliary commutation branches for each other.

[0011] According to the high-precision power amplifier based on the resonance auxiliary commutation technology of the present invention, SPWM modulation is used for the two half-bridge arms, so that the two half-bridge arms are staggered and operated in parallel to form a differential mode current, and the auxiliary inductor L d1 , auxiliary inductor L d2 It resonates with the junction capacitance of the four MOSFET switch tubes S1 to S4 to achieve zero voltage switching in continuous current mode.

[0012] The present invention also provides a parameter design method for a high-precision power amplifier based on the resonance-assisted commutation technology, and performs parameter design on the high-precision power amplifier based on the resonance-assisted commutation technology;

[0013] For the filter inductor L o And filter capacitor C o The filter consists of a filter inductor L o And filter capacitor C o Need to meet:

[0014]

[0015] 2π·f o,max ·L o,eff I o ≤k v ·U o ,

[0016] 2π·f o,max ·C o ·U o ≤k i I o ,

[0017] Where L o,eff is the filter inductor L o The equivalent value of k f is the cut-off frequency coefficient, f o,max is the maximum value of the actual output frequency of the power amplifier; I o is the effective value of the output current, k v is the maximum inductance voltage drop coefficient, U o is the effective value of the output voltage, k i is the maximum capacitance shunt factor.

[0018] According to the parameter design method of the high-precision power amplifier based on the resonance auxiliary commutation technology of the present invention, the auxiliary inductor L d1 and auxiliary inductor L d2 The design value must meet:

[0019] Flowing through the auxiliary inductor L d1 and auxiliary inductor L d2 The peak current is not greater than the soft switching current threshold of the switch tubes S1 to S4; the switch tubes S1 to S4 maintain soft switching at maximum load.

[0020] According to the parameter design method of the high-precision power amplifier based on the resonance auxiliary commutation technology of the present invention, the auxiliary inductor L d1 and auxiliary inductor L d2 The inductance value is expressed as L d , then L d The peak current ΔI dm The following conditions must be met:

[0021]

[0022] Where the peak current ΔI dm is the peak-to-peak value of the differential mode current, t is the time, U dc is the DC bus voltage between the positive and negative poles of the power supply, m is the modulation ratio, T s is the switching cycle.

[0023] According to the parameter design method of the high-precision power amplifier based on the resonance-assisted commutation technology of the present invention, the switch tubes S1 to S4 maintain soft switching at the maximum load, and L d The following conditions must be met

[0024]

[0025] i s_dx When the corresponding switch tube is turned off, the current flowing through the auxiliary inductor L d1 Or flows through the auxiliary inductor L d2 The current, C oss is the equivalent junction capacitance of the MOSFET switch tube, I dx,ZVS is the soft switching current threshold; Q oss C oss The switch tube withstands voltage U ds The amount of charge stored when ds is the positive bus voltage.

[0026] Beneficial effects of the present invention: The power amplifier of the present invention can be used in precision driving of voice coil motors. It is based on auxiliary resonant commutation (ACRP) technology, can achieve soft switching of the power amplifier in the full load range, eliminates current zero-crossing distortion and dead zone effect, and is insensitive to control system delay. BRIEF DESCRIPTION OF THE DRAWINGS

[0027] Figure 1 is a circuit diagram of a high-precision power amplifier based on resonance-assisted commutation technology according to the present invention; oss1 is the junction capacitance of switch tube S1, C oss2 is the junction capacitance of switch tube S2, C oss3 is the junction capacitance of switch tube S3, C oss4 is the junction capacitance of the switch tube S4, i d1 is the current flowing through the auxiliary inductor L d1 The current, i d2 is the current flowing through the auxiliary inductor L d2 The current, i Lo For the current filter inductor L o The current, i o is the load current;

[0028] Figure 2 This is a typical waveform diagram of the resonant assisted commutation soft switch of the power amplifier in the parameter design method of the high-precision power amplifier based on the resonant assisted commutation technology of the present invention; S1_d ~d S4_d is the driving signal of the switch tubes S1 to S4 including the dead zone, U A is the voltage at point A, UB is the voltage at point B;

[0029] Figure 3 d is a dead zone effect diagram of the power amplifier of the present invention using SPWM modulation; S1 to d S4 is the driving signal of the switch tubes S1 to S4 without dead zone, u′ A and u′ B is the ideal value of the voltage at points A and B without dead zone, u′ AR and u′ BR is the actual value of the voltage at points A and B when the dead zone is included, u' Ae and u' Be Represents the error between the ideal value and the actual value, that is, u' Ae =u' A -u' AR ,u' Be =u' B -u' BR ;U' eq is the average error voltage at point C;

[0030] Figure 4 yes Figure 3 The enlarged view of the dotted line in the middle; A1 is the positive voltage error, A2 is the negative voltage error, T r1 is the negative resonance time, that is, u' BR The fall time, T r2 is the forward resonance time, that is, u' BR Rise time;

[0031] Figure 5 The filter inductor L o And filter capacitor C o Schematic diagram of the phase relationship of the parameters of the filter composed at the base frequency; in the figure, u Lo is the filter inductor L o Voltage;

[0032] Figure 6 yes Figure 5 Equivalent circuit diagram of

[0033] Figure 7 is a schematic diagram of the design space of the filter; in the figure, f c is the filter cutoff frequency;

[0034] Figure 8 This is a schematic diagram of the fitting results of the junction capacitance when the switch tube model is C3M0060065K;

[0035] Fig. 9 It corresponds to Figure 8 Schematic diagram of charge calculation results; Q ossThe output charge of the switch junction capacitance;

[0036] Fig.10 This is a schematic diagram of the fitting results of the junction capacitance when the switch tube model is IDL08G65C5; D is the diode junction capacitance, U R is the reverse cut-off voltage of the diode;

[0037] Fig.11 It corresponds to Fig.10 Schematic diagram of charge calculation results; Q D is the output charge of the diode junction capacitance;

[0038] Fig.12 YesL d Schematic diagram of the design space; neg is the negative soft switching current under maximum load conditions;

[0039] Fig.13 1 is a current experimental waveform diagram of the load current and the three inductors in the embodiment; Lo1 For flow Fig.23 The current of the filter inductor Lo1 in the first module, i Lo2 For flow Fig.23 The current of the filter inductor Lo2 in the second module, i Lo3 For flow Fig.23 The current of the filter inductor Lo3 in the third module;

[0040] Fig.14 is a diagram showing the results of a soft switch experiment in the embodiment; ds2 is the drain-source voltage of switch tube S2, u gs2 is the gate-source voltage of the switch tube S2;

[0041] Fig.15 is a waveform diagram of the voltage error experiment in the embodiment; gs1 is the gate-source voltage of switch tube S1, u A Equivalent to U A ;

[0042] Fig.16 is the auxiliary inductor L in the embodiment d1 and auxiliary inductor L d2 The current self-balanced current experiment result diagram;

[0043] Fig.17 is the average error voltage U under different dead zones in the embodiment e Schematic diagram;

[0044] Fig.18 is a schematic diagram of average error voltage at different switching frequencies in the embodiment; sis the switching frequency;

[0045] Fig.19 is a schematic diagram of average error voltage under different loads in the embodiment;

[0046] Fig. 20 In the embodiment, the input voltage is 300V, the load winding is 30mH+4.8Ω, and f s =250kHz, f o =25Hz, T d = Schematic diagram of load current THD experimental results at 100ns and 150ns;

[0047] Fig.21 In the embodiment, the input voltage is 300V, the load winding is 30mH+4.8Ω, and f s =250kHz, f o =25Hz, T d = Load current THD at 100ns; 4.8mH+0.9Ω load winding, f s =250kHz, f o =250Hz, T d = Schematic diagram of load current THD experimental results when 100ns;

[0048] Fig. 22 In the embodiment, the input voltage is 100V, the load winding is 30mH+4.8Ω, T d =100ns, f o = 25Hz load current THD test results diagram; when f s =250kHz, L d =15mH; when f s =62.5kHz, L d =60mH;

[0049] Fig.23 It is a schematic diagram of connecting multiple modules of the high-precision power amplifier of the present invention in parallel;

[0050] Fig.24 It corresponds to Figure 2 Operation modal diagram of the power amplifier at each time point in the figure; (a) is the operation modal diagram of stage 1; (b) is the operation modal diagram of stage 2; (c) is the operation modal diagram of stage 3; (d) is the operation modal diagram of stage 4; (e) is the operation modal diagram of stage 5; (f) is the operation modal diagram of stage 6; (g) is the operation modal diagram of stage 7; (h) is the operation modal diagram of stage 8; (i) is the operation modal diagram of stage 9; (j) is the operation modal diagram of stage 10; (k) is the operation modal diagram of stage 11; (l) is the operation modal diagram of stage 12

[0051] Fig.25 is an equivalent circuit diagram of the power amplifier of the present invention in the resonant and diode clamped modes;

[0052] Fig.26 is the auxiliary inductor L d1 Auxiliary commutation soft switch power amplifier state plane trajectory diagram;

[0053] Fig. 27 is the auxiliary inductor L d2 State plane trajectory diagram of the auxiliary commutation soft switch power amplifier. DETAILED DESCRIPTION

[0054] The following will be combined with the drawings in the embodiments of the present invention to clearly and completely describe the technical solutions in the embodiments of the present invention. Obviously, the described embodiments are only part of the embodiments of the present invention, not all of the embodiments. Based on the embodiments of the present invention, all other embodiments obtained by ordinary technicians in this field without creative work are within the scope of protection of the present invention.

[0055] It should be noted that, in the absence of conflict, the embodiments of the present invention and the features in the embodiments may be combined with each other.

[0056] The present invention will be further described below in conjunction with the accompanying drawings and specific embodiments, but they are not intended to limit the present invention.

[0057] Specific implementation method 1. Combination Figure 1 As shown, the present invention provides a high-precision power amplifier based on resonance auxiliary commutation technology, including a capacitor C1, a capacitor C2, four MOSFET switch tubes S1, S2, S3, S4, an auxiliary inductor L d1 , auxiliary inductor L d2 , filter inductor L o And filter capacitor C o ,

[0058] Capacitor C1 and capacitor C2 are connected in series between the positive electrode of the power supply and the negative electrode of the power supply;

[0059] The positive electrode of the power supply is connected to the drain of the switch tube S1, the source of the switch tube S1 is connected to the drain of the switch tube S2, and the source of the switch tube S2 is connected to the negative electrode of the power supply; the positive electrode of the power supply is also connected to the drain of the switch tube S3, the source of the switch tube S3 is connected to the drain of the switch tube S4, and the source of the switch tube S4 is connected to the negative electrode of the power supply;

[0060] The connection point between the source of switch tube S1 and the drain of switch tube S2 is taken as point A, the connection point between the source of switch tube S3 and the drain of switch tube S4 is taken as point B, and an auxiliary inductor L is connected in series between point A and point B. d1 and auxiliary inductor L d2 ; Set the auxiliary inductor Ld1 and auxiliary inductor L d2 The connection point between them is taken as point C, and the filter inductor L is connected in series between point C and the connection point of capacitor C1 and capacitor C2. o And filter capacitor C o ; Load R and filter capacitor C o in parallel.

[0061] In this embodiment, the auxiliary inductor L d1 and auxiliary inductor L d2 The inductance values ​​are equal.

[0062] Furthermore, this embodiment adopts the resonance auxiliary commutation technology, by adding the auxiliary inductor L d1 and auxiliary inductor L d2 The two half-bridge arms formed by the switch tubes S1 to S4 are coupled so that the two half-bridge arms serve as resonant auxiliary commutation branches. Figure 1 shown.

[0063] Use SPWM modulation on the two half-bridge arms to make the two half-bridge arms run in parallel and staggered to form a differential mode current, and make the auxiliary inductor L d1 , auxiliary inductor L d2 It resonates with the junction capacitance of the four MOSFET switch tubes S1 to S4 to achieve zero voltage switching (ZVS-ON) in continuous current mode (CCM).

[0064] In this embodiment, the auxiliary filter inductor L d1 and L d2 The current has already passed zero before the switch tube (SiC MOSFET) is turned off, so the current will not be clamped to 0 by the anti-parallel diode of the switch tube, overcoming the current zero-crossing distortion of the traditional half-bridge amplifier. In addition, the resonant auxiliary commutation process eliminates the dead zone effect, and there is only a slight dead voltage error caused by the non-ideal rising and falling edges of the SiC MOSFET drain-source voltage, such as Figure 3 and Figure 4 shown.

[0065] The amplifier described in this embodiment is compatible with SPWM modulation, has a fixed switching frequency, and is insensitive to control system software and hardware delays. Therefore, a more complex control algorithm can be used, and a delta-sigma ADC sampling chip with high accuracy but large delay can also be used in conjunction.

[0066] Specific implementation method 2: Combination Figures 1 to 12 As shown, the present invention also provides a parameter design method for a high-precision power amplifier based on resonance-assisted commutation technology, and performs parameter design on the high-precision power amplifier based on resonance-assisted commutation technology described in the first specific implementation mode;

[0067] For the filter inductor L o And filter capacitor C o The filter composed of two parts needs to be designed to establish the filter design space. For the filter, its design needs to consider two requirements: one is to meet the bandwidth requirements of the power amplifier; the other is the filter inductor voltage drop u Lo and filter capacitor current i Co The value cannot be too large, that is, the reactive power needs to be limited.

[0068] Filter inductor L o And filter capacitor C o Need to meet:

[0069] The following restrictions should be imposed on bandwidth:

[0070]

[0071] right Figure 1 Do equivalent processing, then the filter equivalent input voltage is u sw,eff , the total input current is i sum . Figure 5 is the load current i o , load voltage u o with u sw,eff and i sum Schematic diagram of the phase relationship between them; Figure 6 It is the equivalent circuit of multiple modules staggered in parallel. When the output frequency increases, the reactive power absorbed by the filter capacitor and filter inductor will increase accordingly. If the parameter design is unreasonable, excessive reactive power may burn the switching device. Therefore, the filter parameter design needs to take into account the maximum output frequency i Co and u Lo , that is, the following two constraints need to be considered:

[0072] 2π·f o,max ·L o,eff I o ≤k v ·U o ,

[0073] 2π·f o,max ·C o ·U o ≤k i I o ,

[0074] Where L o,eff is the filter inductor L o The equivalent value of k f is the cut-off frequency coefficient, which is generally taken as an empirical value of 4; f o,max is the maximum value of the actual output frequency of the power amplifier; Io is the effective value of the output current, k v is the maximum inductance voltage drop coefficient, U o is the effective value of the output voltage, k i is the maximum capacitance shunt coefficient. k v and k i The possible values ​​are 15% and 33% respectively.

[0075] In summary, we can get Figure 7 The filter design space shown in Figure 1. The area that satisfies the bandwidth constraint is the lower left area of ​​the blue hyperbola, which satisfies the maximum u Lo The area of ​​the constraint should be to the left of the yellow straight line, satisfying the maximum i Co The area of ​​constraints should be the lower side of the orange straight line, and the final filter design space is the intersection of the three-phase constraints.

[0076] The ideal filter parameter value point should be located in the upper right part of the filter design space. At the same time, the influence of the device standard value should also be considered. For example, the capacitance value is usually 220nF or 330nF. Take the following indicators as an example: ① DC bus voltage 2U dc =300V;②Output current effective value I o The peak value reaches 10A (resistive load); ③ switching frequency 250kHz; ④ target bandwidth 20kHz. The final filter parameters are the point marked by the red five-pointed star (L o =10μH, C o =220nF).

[0077] Furthermore, the auxiliary inductor parameters are designed:

[0078] Auxiliary inductor L d1 and auxiliary inductor L d2 The design value must meet:

[0079] Under the premise of given input voltage level, switching frequency and filter inductance, the current flowing through the auxiliary inductor L d1 and auxiliary inductor L d2 The peak current is not greater than the soft switching current threshold of the switch tubes S1 to S4, otherwise the switch tubes are easily damaged; the switch tubes S1 to S4 still maintain soft switching at the maximum load.

[0080] For the first condition: Set the auxiliary inductor L d1 and auxiliary inductor L d2 The inductance value is expressed as L d , then L d The peak current ΔI dm The following conditions must be met:

[0081]

[0082] Where the peak current ΔI dm is the peak-to-peak value of the differential mode current, which must be within the safe operating area (SOA) of the SiC MOSFET; t is the time, U dc is the DC bus voltage between the positive and negative poles of the power supply, m is the modulation ratio, T s is the switching cycle.

[0083] For the second condition, the switches S1 to S4 maintain soft switching at maximum load, L d The following conditions must be met:

[0084]

[0085] i s_dx When the corresponding switch tube is turned off, the current flowing through the auxiliary inductor L d1 Or flows through the auxiliary inductor L d2 The current, C oss is the equivalent junction capacitance of the MOSFET switch tube, I dx,ZVS is the soft switching current threshold; Q oss C oss The switch tube withstands voltage U ds The amount of charge stored when ds is the positive bus voltage. oss It is the sum of the equivalent junction capacitance of SiC MOSFET and SiC SBD.

[0086] The following takes C3M0060065K and IDL08G65C5 as examples. Figure 8 and Fig.10 The fitting results of the junction capacitances of the two are given respectively. Fig. 9 and Fig.11 The calculation results of the output charge of the two are given respectively. The equivalent output capacitance values ​​of C3M0060065K and IDL08G65C5 at 300V can be calculated as 153.3nF and 46.7nF respectively by using the junction capacitance calculation formula.

[0087] Combine the following Figure 2 , Figure 24 to Figure 27 The working principle of the power amplifier of the present invention is described as follows:

[0088] Figure 2 and Fig.24 The typical waveforms of the power amplifier of the present invention and 12 specific operating modes within a switching cycle are shown. S1 to S4 are SiC MOSFETs, L d1 and L d2 is the auxiliary inductor, the value of the auxiliary inductor is equal, that is, Ld1 =L d2 =L d .L o is the filter inductor. C o is the filter capacitor, C oss1 ~C oss4 is the equivalent junction capacitance of SiCMOSFET, and C oss1 =C oss2 =C oss3 =C oss4 =C oss . The input voltage is ±U dc , the output voltage is u o .i d1 、i d2 、i Lo and i o They are two auxiliary inductor currents, filter inductor current and output load current. d1 Composed of a bridge arm unit, S3, S4 and L d2 The two bridge arms have the same parameters and serve as auxiliary commutation bridge arms for each other.

[0089] Among these 12 modes, there are 4 resonance auxiliary stages. The equivalent junction capacitance of SiCMOSFET involved in the resonance can be calculated as follows:

[0090]

[0091] Where Q oss The junction capacitance is the voltage U that the switch tube withstands ds The amount of charge stored when oss (U ds ) can be obtained from the data sheet.

[0092] During resonance, the characteristic impedance Z r and angular frequency ω r for:

[0093]

[0094] It should be emphasized here that in practical applications, SiC MOSFET is usually connected in parallel with SiC SBD, so C in formula (2) oss By default, the junction capacitance of SiC SBD is not included. The junction capacitance of SiC SBD is calculated by formula (1), but the signs of the variables are different, so it will not be repeated here.

[0095] The common mode voltage u output by the two bridge arms C for:

[0096]

[0097] Where u A is the voltage at point A, u B is the voltage at point B;

[0098] Current i d1 and i d2 It can be decomposed into the differential mode current i dm and the common mode current i cm :

[0099]

[0100] The voltage drop across the filter inductor is given by u C And the output voltage u o Together, according to Kirchhoff's current law, the filter inductor current i Lo with i d1 、i d2 The sum is expressed as:

[0101]

[0102] According to formulas (4) and (5), we can get i d1 、i d2 The current expression is:

[0103]

[0104] Next, the operating modes of the power amplifier are analyzed.

[0105] Phase 1: [t0, t1): Figure 2 and Fig.24 As shown in (a). At t0, S2 realizes ZVS-ON, after which S2 and S3 remain open, and the voltages at points A and B are pulled down and up to -U respectively. dc and +U dc .i d1 in-U dc The incentive continues to decline, i d2 in U dc The incentive for continued growth. cm In SiC MOSFET, L o and L d The voltage generated at both ends is related to U dc Together they form u o ,i dm In L d1 and L d2 The voltage generated on the dc The difference between the on-state voltage drop of SiC MOSFET and the on-state voltage drop of SiC MOSFET. In summary, the equation group is:

[0106]

[0107] Where R ds,on is the on-state resistance of the switch tube.

[0108] Solving the equations (6) and (7) yields the following equation: d1 、i d2 The expression is:

[0109]

[0110] Among them I cm,t0 and I dm,t0 is time t0 cm and i dm The initial value of .

[0111] Phase 2: [t1, t2): before t1, i d1 The current is reversed, S2 is turned off at t1, i d1 Start charging the equivalent junction capacitance of S1 and discharging the equivalent junction capacitance of S2. oss With L d +L d ||L o Series resonance occurs. Lo The amplitude change is much smaller than i d1 and i d2 , so for the convenience of calculation, assume that i Lo The current remains unchanged during the resonance period, and its current value is always equal to the initial value at time t1. The initial value of the circuit at this stage is u S1 (t1)=2U dc ,u S2 (t1) = 0, i d1 (t1) = I d1,t1 ,i Lo (t1) = I Lo,t1 The corresponding current path is Fig.24 As shown in (b), the equivalent circuit is Fig.25 (a). The circuit equations are as follows:

[0112]

[0113] Where u S1 is the drain-source voltage of switch tube S1, u S2 is the drain-source voltage of the switch tube S2.

[0114] Solve to get i d1 、i d2 、u S1 and u S2 :

[0115]

[0116] The trajectory diagram corresponding to the resonance process described by this mode is Fig.26 The radius of the arc is r1, and the expression of the radius is:

[0117] r1=Z r |i s_dx |(11),

[0118] According to the geometric relationship shown in the trajectory diagram, r1≥2U dc To ensure that you S2 From 0 to 2U dc , which in turn makes C oss1 At t2, it discharges to 0V, achieving ZVS-ON of S1. Otherwise, C oss1 The discharge will not be completed before S1 is turned on, resulting in partial loss of ZVS-ON. The minimum i required to meet full ZVS-ON is s dx The following conditions should be met:

[0119]

[0120] I dx,ZVS This is the soft switching current threshold.

[0121] Phase 3: [t2, t3): after time t2, i d1 Flows through diode D1 (anti-parallel diode of S1), u S1 and u S2 Clamped by D1 at 0 and 2U respectively dc , its equivalent circuit is Fig.25 (b). This ensures that S1 is ZVS-ON at time t3. Lo Start to rise.

[0122] Phase 4: [t3, t4): At t3, S1 achieves ZVS-ON, S1 and S3 are both in the on state, and the voltages at points A and B are both U dc ,i d1 and i d2 The direction is opposite. The circuit equations are as follows:

[0123]

[0124] Solving the equations (6) and (13) yields d1 and i d2 The expression is as follows:

[0125]

[0126] Among them I cm,t3 and I dm,t3It is the initial value of the common mode current and the differential mode current at time t3.

[0127] Phase 5: [t4, t5): S3 is turned off at t4, L d +L d ||L o The equivalent junction capacitance of S3 and S4 is 2C oss Series resonance occurs, and the resonant current is oss3 Charging, for C oss4 Discharge. The initial value of the circuit at this stage is u S3 (t4) = 0, u S4 (t4) = 2U dc ,i d2 (t4) = I d2,t4 ,i Lo (t4) = I Lo,t4 At this stage, the current path is as follows Fig.24 (e) shows that the corresponding equivalent circuit is as follows: Fig.25 (c). The circuit equations are as follows:

[0128]

[0129] Solve for i d1 、i d2 、u S3 and u S4 The expression is:

[0130]

[0131] Where I d2,t4 is time t4 d2 The initial value of u S3 is the drain-source voltage of switch tube S3, u S4 is the drain-source voltage of the switch tube S4.

[0132] The trajectory diagram corresponding to this mode is Fig. 27 The radius of the arc is r4, and the expression of the radius is:

[0133]

[0134] When the resonance process is over, C oss4 Discharge to 0V.

[0135] Phase 6: [t5, t6): After t5, i d2 Flows through diode D4 (anti-parallel diode of S4), u S3 and u S4 Clamped in 2U dc and 0, its equivalent circuit is Fig.25(d). This ensures that S4 achieves ZVS-ON at time t6. Lo Start to descend.

[0136] Phase 7: [t6, t7): At t6, S4 realizes ZVS-ON. Since the voltages at points A and B are U dc and -U dc , then the voltage across the auxiliary inductor is 2U dc make i d1 Rise, i d2 Decline, and before t7 i d2 The direction changes, so that the charging and discharging of the equivalent junction capacitance of S1 and S2 are completed in the resonance process after t7, realizing soft switching. d1 、i d2 and i Lo The change process of the three can be described by the following equations:

[0137]

[0138] Solving the equations shown in formulas (6) and (18) yields d1 、i d2 The expression is:

[0139]

[0140] Among them I cm,t6 and I dm,t6 It is the initial value of the common mode current and differential mode current at time t6.

[0141] Phase 8: [t7, t8): S4 is turned off at t7, L d +L d ||L o The equivalent junction capacitance of S3 and S4 is 2C qoss Series resonance occurs, and the resonant current is oss3 Discharge, to C oss4 Charging. The initial value of the circuit is u S3 (t7) = 2U dc ,u S4 (t7) = 0, i d2 (t7) = I d2,t7 ,i Lo (t7) = I Lo,t7 At this stage, the current path is as follows Fig.24 (h) shows that the corresponding equivalent circuit is Fig.25 (e) is shown. The circuit equations are as follows:

[0142]

[0143] Solve for i d1、i d2 、u S3 and u S4 The expression is as follows:

[0144]

[0145] The trajectory corresponding to this stage is Fig. 27 The radius of the arc with radius r3 is calculated the same as in stage 2.

[0146] Stage 9: [t8, t9): After t8, i d2 Flows through diode D3 (anti-parallel diode of S3), u S3 and u S4 are clamped at 0 and 2U respectively dc , its equivalent circuit is Fig.25 (f). This ensures that S3 achieves ZVS-ON at time t9. Lo Start to rise.

[0147] Phase 10: [t9, t 10 ): At t9, S3 realizes ZVS-ON, S1 and S3 are both in the on state, and the voltages at points A and B are both U dc .i d1 and i d2 The direction is opposite. The circuit equation is:

[0148]

[0149] Solve the system of equations shown in formulas (6) and (22):

[0150]

[0151] Among them I cm,t9 and I dm,t9 It is the initial value of the common mode current and differential mode current at time t9.

[0152] Phase 11: 10 , t 11 ):t 10 At the moment S1 is turned off, L d +L d ||L o The sum of the equivalent junction capacitance of S1 and S2 is 2C oss Series resonance occurs, and the resonant current is oss1 Charging, for C oss2 Discharge. The initial value of the circuit at this stage is u S1 (t 10 )=0,u S2 (t 10 )=2U dc ,id1 (t 10 )=I d1,t10 ,i Lo (t 10 )=I Lo,t10 At this stage, the current path is as follows Fig.24 (k) shows that the corresponding equivalent circuit is Fig.25 (g). The circuit equations are as follows:

[0153]

[0154] Solve for i d1 、i d2 、u S1 and u S2 The expression is:

[0155]

[0156] The trajectory corresponding to this stage is Fig.26 The radius of the arc with radius r2 is calculated the same as in stage 5.

[0157] Phase 12: 11 , t 12 ): t 11 After the moment d1 Flows through diode D2 (anti-parallel diode of S2), u S1 and u S2 Clamped in 2U dc and 0, its equivalent circuit is Fig.25 (h). This ensures that S2 achieves ZVS-ON at time t0 in the next cycle. 11 After the moment Lo Start to descend.

[0158] Thus, the complete operation mode of the power amplifier is obtained. In this embodiment, the number of bridge arms staggered in parallel can be two or more. The specific number of bridge arms staggered in parallel is selected according to actual use requirements.

[0159] Embodiment: The effect of the present invention is described below by embodiments:

[0160] In this embodiment, capacitor C1, capacitor C2, four MOSFET switch tubes S1, S2, S3, S4, auxiliary inductor L d1 , auxiliary inductor L d2 and filter inductor L o The structure is a module. In actual use, multiple modules can be used in parallel according to the use requirements to meet the soft switching current threshold and safety current threshold requirements.

[0161] Take the following indicators as an example: ① DC bus voltage 2U dc =300V;②Output current I o Reach peak value 10A (resistive load); ③ switching frequency 250kHz; ④ target bandwidth 20kHz. The above conditions ① and ② correspond to Fig.12 The area to the right of the green vertical line and the area below the purple curve, where the green vertical line corresponds to 10μH. The three red lines with gradient colors are different L values ​​under 1 to 3 modules. d The maximum soft switching current at this time can be calculated by the following formula; Obviously, at least two modules need to be staggered in parallel to meet the soft switching current threshold and safety current threshold requirements at the same time. Finally, the number of staggered parallel modules is selected as 3, as shown in Fig.23 As shown. Finally, the auxiliary inductor L is selected d =15μH.

[0162]

[0163] Where i neg is the negative soft switching current, N is the total number of bridge arms, and R is the load.

[0164] Fig.13 The 20kHz load current and filter inductor current experimental waveforms show the actual load current i of the prototype. o and the three filter inductor currents i Lo1 ,i Lo2 and i Lo3 The peak load current of the prototype is 10A, and the bandwidth is 20kHz. There is no current attenuation at this bandwidth, so 20kHz is the available bandwidth rather than the bandwidth defined by -3dB. The switching frequency is 250kHz (three modules are staggered in parallel, and the equivalent switching frequency is 1.5MHz), and there is no obvious distortion of the sinusoidal current.

[0165] Fig.14 The soft switching waveform of S2 under maximum load condition is shown. ds2 After falling to zero and remaining there for a short time, u gs2 That is, the drain-source voltage drop process of S2 ends before its drain current rises, and S2 achieves ZVS.

[0166] Fig.15 The experimental waveform of shows the slight voltage error caused by the non-ideal voltage rising and falling edges. A with u gs1 and u gs2 For example, u A The falling edge of u gs1 The falling edge of u A The rising edge of u gs2The falling edge of u A The edge of the drain-source voltage is synchronized with the gate signal, and the voltage error caused by the dead zone is eliminated. Only a small voltage error caused by the non-ideal rising and falling edges of the drain-source voltage exists.

[0167] Fig.16 The experimental results of the auxiliary inductor current self-balanced current are shown. The input voltage of this experiment is 2U dc =100V, the switching frequency is 250kHz, and the load current frequency is 500Hz. Initially, the S1 turn-off time is delayed by 10ns to simulate the duty cycle d A (t) fluctuation. It can be seen that the 10ns turn-off delay causes i d1 The peak value rises to 3.8A, i d2 The peak value of the current drops to 2.9A, resulting in a current deviation of about 15%. When the 10ns S1 turn-off delay is cancelled, the current self-balancing is achieved within one modulation wave cycle without any delay or oscillation.

[0168] Fig.17 , Fig.18 and Fig.19 The average error voltage Fourier series (taking the first 100 harmonics) of the traditional staggered parallel half-bridge and the power amplifier proposed in the present invention under different dead zones, switching frequencies and load currents. Specifically, Fig.17 This indicates that the average error voltage of the power amplifier proposed by the present invention has nothing to do with the dead zone, fundamentally avoiding the harmonic distortion caused by the dead zone. The average error voltage of the power amplifier proposed is only about 3% of that of the traditional solution.

[0169] Fig.18 It shows that although the average error voltage of the power amplifier of the present invention increases with the increase of the switching frequency, its amplitude is still about 30 times lower than that of the traditional solution. Therefore, the power amplifier of this embodiment can withstand a higher switching frequency while maintaining a relatively low THD.

[0170] Depend on Fig.19 It can be seen that the average error voltage of the power amplifier of the present invention increases by about 12 times in the load current range of 3.3 to 10 A, but is still much lower than that of the traditional solution, showing a huge advantage.

[0171] Fig. 20 Shows different dead zones T d The load current THD measured data under o,peak is the peak value of the load current. Although THD increases with the increase of load, it is generally at a low level between 0.3% and 0.8%. From the experimental results, it can be seen that the load current THD of the power amplifier proposed by the present invention is not affected by T d Influence, no need to pursue too small T d, thereby reducing the complexity of hardware circuit design.

[0172] Fig.21 The measured data of load current THD at different output frequencies are shown. When the output frequency is 250Hz, the output current THD is about 0.2% to 0.5% higher than that at 25Hz. This is due to the decrease in open-loop gain when the output frequency is high. In addition, f o =250Hz, I o,peak = 9A, the THD inflection point appears, this is because the modulation ratio decreases at light load, which causes THD to increase.

[0173] Fig. 22 The THD measurement results are shown at 100V input voltage and switching frequencies of 250kHz and 62.5kHz. Since the modulation ratio remains unchanged, the output current peak value I o,peak To ensure that the auxiliary inductor current is the same at 250kHz and 62.5kHz switching frequencies, L d Take 15μH; when the switching frequency is 62.5kHz, L d Take 60μH. The experimental results show that the switching frequency is positively correlated with THD. Fig.16 Another conclusion can be drawn that the THD at 100V input voltage is much lower than that at 300V input. The final THD is as low as 0.063%.

[0174] In practical applications, there is no theoretical limit to the number of bridge arms that are staggered and operated in parallel, but the number of bridge arms must be kept to an even number to eliminate harmonics at multiples of the switching frequency. Fig.23 Theoretically, there is no limit to the number of modules connected in parallel. In practical applications, the number of modules connected in parallel can be determined according to the load size.

[0175] Although the present invention is described herein with reference to specific embodiments, it should be understood that these embodiments are merely examples of the principles and applications of the present invention. It should therefore be understood that many modifications may be made to the exemplary embodiments and that other arrangements may be devised without departing from the spirit and scope of the present invention as defined by the appended claims. It should be understood that the various dependent claims and features described herein may be combined in a manner different from that described in the original claims. It will also be understood that the features described in conjunction with a single embodiment may be used in other described embodiments.

Claims

1. A high-precision power amplifier based on resonance-assisted commutation technology, characterized in that: Including capacitor C1, capacitor C2, four MOSFET switch tubes S1, S2, S3, S4, auxiliary inductor L d1 , auxiliary inductor L d2 、Filter inductor L o And filter capacitor C o , Capacitor C1 and capacitor C2 are connected in series between the positive electrode of the power supply and the negative electrode of the power supply; The positive electrode of the power supply is connected to the drain of the switch tube S1, the source of the switch tube S1 is connected to the drain of the switch tube S2, and the source of the switch tube S2 is connected to the negative electrode of the power supply; the positive electrode of the power supply is also connected to the drain of the switch tube S3, the source of the switch tube S3 is connected to the drain of the switch tube S4, and the source of the switch tube S4 is connected to the negative electrode of the power supply; The connection point between the source of switch tube S1 and the drain of switch tube S2 is taken as point A, the connection point between the source of switch tube S3 and the drain of switch tube S4 is taken as point B, and an auxiliary inductor L is connected in series between point A and point B. d1 and auxiliary inductor L d2 ; Set the auxiliary inductor L d1 and auxiliary inductor L d2 The connection point between them is taken as point C, and the filter inductor L is connected in series between point C and the connection point of capacitor C1 and capacitor C2. o And filter capacitor C o ; Load R and filter capacitor C o in parallel.

2. The high-precision power amplifier based on resonance-assisted commutation technology according to claim 1, characterized in that: Auxiliary inductor L d1 and auxiliary inductor L d2 The inductance values ​​are equal.

3. The high-precision power amplifier based on resonance-assisted commutation technology according to claim 2, characterized in that: The auxiliary inductor L d1 and auxiliary inductor L d2 The two half-bridge arms formed by the switch tubes S1 to S4 are coupled so that the two half-bridge arms serve as resonant auxiliary commutation branches for each other.

4. The high-precision power amplifier based on resonance-assisted commutation technology according to claim 3, characterized in that: Use SPWM modulation on the two half-bridge arms to make the two half-bridge arms run in parallel and staggered to form a differential mode current, and make the auxiliary inductor L d1 , auxiliary inductor L d2 It resonates with the junction capacitance of the four MOSFET switch tubes S1 to S4 to achieve zero voltage switching in continuous current mode.

5. A method for designing parameters of a high-precision power amplifier based on resonance-assisted commutation technology, which performs parameter design on the high-precision power amplifier based on resonance-assisted commutation technology as claimed in claim 4; characterized in that: For the filter inductor L o And filter capacitor C o The filter consists of a filter inductor L o And filter capacitor C o Need to meet: 2π·f o,max ·L o,eff ·I o ≤k v ·U o , 2π·f o,max ·C o ·U o ≤k i ·I o , Where L o,eff is the filter inductor L o The equivalent value of k f is the cut-off frequency coefficient, f o,max is the maximum value of the actual output frequency of the power amplifier; I o is the effective value of the output current, k v is the maximum inductance voltage drop coefficient, U o is the effective value of the output voltage, k i is the maximum capacitance shunt factor.

6. The parameter design method of a high-precision power amplifier based on resonance-assisted commutation technology according to claim 5, characterized in that: Auxiliary inductor L d1 and auxiliary inductor L d2 The design value must meet the following requirements: Flowing through the auxiliary inductor L d1 and auxiliary inductor L d2 The peak current is not greater than the soft switching current threshold of the switch tubes S1 to S4; the switch tubes S1 to S4 maintain soft switching at maximum load.

7. The parameter design method of a high-precision power amplifier based on resonance-assisted commutation technology according to claim 6, characterized in that: The auxiliary inductor L d1 and auxiliary inductor L d2 The inductance value is expressed as L d , then L d The peak current ΔI dm The following conditions must be met: Where the peak current ΔI dm is the peak-to-peak value of the differential mode current, t is the time, U dc is the DC bus voltage between the positive and negative poles of the power supply, m is the modulation ratio, T s is the switching cycle.

8. The parameter design method of a high-precision power amplifier based on resonance-assisted commutation technology according to claim 7, characterized in that: At maximum load, switches S1 to S4 maintain soft switching, L d The following conditions must be met i s_dx When the corresponding switch tube is turned off, the current flowing through the auxiliary inductor L d1 Or flows through the auxiliary inductor L d2 The current, C oss is the equivalent junction capacitance of the MOSFET switch tube, I dx,ZVS is the soft switching current threshold; Q oss C oss The switch tube withstands voltage U ds The amount of charge stored when ds is the positive bus voltage.

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