Bidirectional isolated AC-DC converter

Abstract

A bidirectional high-voltage AC-DC converter includes a primary three-phase three-level T-type power factor correction circuit, a full-bridge switching circuit connected to an output of the primary three-phase three-level T-type power factor correction circuit, an isolated capacitor-inductor-inductor-capacitor (CLLC) resonant tank connected to the full-bridge switching circuit, and an isolation capacitor-inductor-inductor-capacitor (CLLC) resonant tank connected to the full-bridge switching circuit. A T-type full bridge output topology circuit connected to the CLLC resonant circuit; and a controller controlling one or more of the components of the bidirectional isolated high voltage AC-DC converter. The primary three-phase three-level T-type power factor correction circuit and the secondary three-level circuit include split DC link buses having respective voltages actively controlled by a controller. The CLLC resonant converter circuit includes an isolation transformer that electrically isolates the primary side from the secondary side of the bi-directional isolated high voltage AC-DC converter. The control method includes synthesizing one or more output voltages of a front-end three-phase three-level T-type PFC circuit via a modified space vector pulse width modulation (SVPWM) scheme or a variable amplitude dual carrier sinusoidal triangular PWM with capacitor bank neutral point voltage control, where the modified SVPWM scheme is based at least in part on a latest four-vector technique.

Description

Technical Field

[0001] The present application relates generally to bidirectional AC-DC converters, and more particularly to bidirectional isolated high voltage AC-DC converters and methods of controlling bidirectional AC-DC converters. Background Art

[0002] As many transportation vehicles, including aircraft or ships, turn to electric motors for propulsion or other power needs, DC or AC voltage buses that operate at significantly higher voltage levels than today's conventional are emerging to handle the increasing power levels. For example, traditional transportation vehicle AC distribution networks typically use transformer rectifier units (TRUs) for AC-DC conversion, which are large, cumbersome and have low efficiency. Some traditional transportation vehicle AC distribution networks utilize autotransformer rectifier units (ATRUs) instead of TRUs, which have higher power density and higher efficiency than traditional TRUs. However, one disadvantage associated with ATRUs is the loss of valuable electrical isolation.

[0003] With the latest advancements in power electronics, especially the adoption of wide bandgap power devices (SiC and GaN), ATRU solutions can only be classified as "stopgap" or "make do" because the sacrifice of electrical isolation is high and the gain in power density and efficiency is low. The latest power electronics technology requires the transformation of this traditional function to have the required power density, efficiency and intelligent fault handling capabilities without sacrificing the isolation benefits.

[0004] Conventional flight actuation controllers typically utilize braking resistors internally to dissipate regenerative energy. Unfortunately, this reduces system efficiency and results in higher operating temperatures. With the advent of next generation MEA microgrid power systems, the constraint prohibiting any regenerative energy from flowing back into the AC system may be lifted, at least for low duty cycle energy flows such as those that occur with motor-driven actuation of transportation vehicles. Designing for bidirectional power flow capability will become a new design paradigm to meet the goals of higher energy efficiency and reduced environmental impact. Summary of the invention

[0005] The present application describes a bidirectional high voltage AC-DC converter. The bidirectional high voltage AC-DC converter utilizes the latest power electronics technology to provide a high density bidirectional isolated power processing unit, enabling standard vehicle actuation products to be used on vehicles with an AC power bus power system, while additional features benefit the system level design of the actuation system.

[0006] Embodiments of the present application utilize an innovative AC / DC conversion topology enabled by wide bandgap GaN devices to implement bidirectional power conversion between an AC power source and a split DC bus output, which can be referenced to the vehicle frame with the necessary electrical isolation. Bidirectional power flow enables regenerative energy from flight actuators to flow back to the AC power source, improving overall system efficiency. An advantageous feature of utilizing a split DC bus for partial discharge hazard management is provided to enable optimized electric drive actuation systems, as well as other power requirements for vehicle applications where high-altitude operations would pose significant challenges to the high voltage power bus.

[0007] In an exemplary embodiment, a bidirectional high-voltage AC-DC converter includes a primary three-phase three-level T-type power factor correction circuit, a switch circuit connected to the primary three-phase three-level T-type power factor correction circuit, an isolated capacitor-inductor-inductor-capacitor (CLLC) resonant converter circuit connected to the switch circuit, a T-type full-bridge output topology circuit connected to the CLLC resonant converter circuit, and a controller for controlling one or more of the components of the bidirectional isolated high-voltage AC-DC converter. The primary three-phase three-level T-type power factor correction circuit and the secondary three-level circuit include a split DC link bus having a corresponding voltage actively controlled by the controller. The CLLC resonant converter circuit includes an isolation transformer that electrically separates the primary side of the bidirectional isolated high-voltage AC-DC converter from the secondary side.

[0008] The primary three-phase three-level T-type power factor correction circuit interfaces with the high voltage AC power bus, and the secondary three-level circuit interfaces with a split DC link bus having a regulated symmetrical inner DC bus connected to the rack as a common reference. For example, the regulated symmetrical inner DC bus can be configured to ±135VDC or ±270VDC. With this AC / DC converter, an advanced motor controller design with soft start and optimal inrush control and excellent disconnection function in the event of motor drive inverter failure is achieved.

[0009] The bidirectional isolated high voltage AC-DC converter of the present disclosure provides an optimized solution for system-level partial discharge risk management using an optimized actuator design. It also enables standard actuation products (e.g., ±135VDC or ±270VDC flight control actuation systems) to interface with the high voltage AC bus.

[0010] Therefore, an aspect of the present invention is a bidirectional AC-DC converter circuit. In an exemplary embodiment, the bidirectional AC-DC converter includes: a three-phase three-level T-type power factor correction (PFC) circuit including a primary split DC link bus; a switching circuit connected to the three-phase three-level T-type power factor correction circuit; a capacitor-inductor-inductor-capacitor (CLLC) resonant converter circuit connected to the switching circuit, the CLLC resonant converter circuit including an isolation transformer; and a three-level T-type full-bridge output circuit connected to the CLLC resonant converter circuit; wherein the isolation transformer isolates the switching circuit from the three-level T-type full-bridge output circuit.

[0011] In an exemplary embodiment, the bidirectional AC-DC converter further includes a controller for controlling a front-end three-phase three-level T-type power factor correction (PFC) circuit. In an exemplary embodiment, a control method includes synthesizing a rectified voltage of the front-end three-phase three-level T-type PFC circuit via a modified space vector pulse width modulation (SVPWM) scheme; wherein the modified SVPWM scheme is based on a nearest-four-vectors technique.

[0012] In an exemplary embodiment, the method for controlling a front-end three-phase three-level T-type power factor correction (PFC) circuit further includes: determining a closest first vector, a closest second vector, a closest third vector, and a closest fourth vector, wherein two of the closest first, second, third, or fourth vectors produce output currents i having opposite signs. o , wherein two closest vectors that produce currents with opposite signs are used to calculate a combined closest vector, and wherein a nearest three vector synthesis is performed on the combined closest vector and the other two closest vectors.

[0013] These and other features of the present invention will be apparent with reference to the following description and accompanying drawings. In the specification and accompanying drawings, specific embodiments of the present invention have been disclosed in detail as indicating some of the ways in which the principles of the present invention may be used, but it is understood that the scope of the present invention is not correspondingly limited. On the contrary, the present invention includes all changes, modifications and equivalents falling within the spirit and terms of the appended claims. The features described and / or illustrated with respect to one embodiment may be used in one or more other embodiments in the same manner or in a similar manner and / or combined with or replace the features of other embodiments. BRIEF DESCRIPTION OF THE DRAWINGS

[0014] Figure 1 is a drawing illustrating a schematic block diagram of an exemplary bidirectional isolated high voltage AC-DC converter.

[0015] Figure 2 is a drawing showing an exemplary circuit diagram of a bidirectional isolated high voltage AC-DC converter.

[0016] Figure 3 is a plot illustrating an exemplary analytical model of a bidirectional isolated high voltage AC-DC converter.

[0017] Figure 4 is a drawing showing the two-dimensional space vector diagram of the primary side three-level full-bridge converter.

[0018] Figure 5 is a drawing showing a two-dimensional space vector diagram illustrating the voltage space vector synthesis of a primary side three-level full-bridge converter.

[0019] Figure 6 is a drawing illustrating a PFC stage control loop structure with an exemplary NFV SVPWM.

[0020] Figure 7 is a graph illustrating a variable amplitude dual carrier sine-triangle PWM implementation of an exemplary NFV SVPWM of a PFC stage.

[0021] Figure 8 is a drawing illustrating an exemplary flow chart of a method for controlling bidirectional conversion of a high AC voltage to a DC voltage. DETAILED DESCRIPTION

[0022] Embodiments of the present application will now be described with reference to the accompanying drawings, wherein like reference numerals are used to refer to like elements throughout. It will be understood that these drawings are not necessarily to scale.

[0023] Figure 1 A schematic block diagram of an exemplary bidirectional isolated AC-DC converter 100 is illustrated. Figure 2 is a more detailed circuit diagram showing Figure 1 Components of a bidirectional isolated AC-DC converter 100. Figure 1 and Figure 2 In the example, the bidirectional isolated AC-DC converter 100 includes a front-end three-phase three-level T-type power factor correction (PFC) circuit 110, which includes a primary split DC link bus 160, a switching circuit 120 connected to the primary split DC link bus, a capacitor-inductor-inductor-capacitor (CLLC) resonant converter circuit 130 connected to the switching circuit 120, and a three-level T-shaped full-bridge output circuit 140 connected to the CLLC resonant converter circuit 130.

[0024] Specific reference Figure 2, the primary split DC link bus 160 includes a primary positive DC bus 201, a primary midpoint bus 202, and a primary negative bus 203. The CLLC resonant converter circuit 130 includes an isolation transformer 215 that electrically isolates the switching circuit from the three-level T-type full-bridge output circuit. The three-level T-type full-bridge output circuit includes a secondary split DC bus 170. The secondary split DC bus 170 includes a secondary positive DC bus 212, a secondary midpoint bus 213, and a secondary negative DC bus 214. In the depicted example, the secondary positive DC bus 212 is +135 VDC, the secondary midpoint bus 213 is grounded, and the secondary negative DC bus 213 is -135 VDC. The secondary positive DC bus 212 and the secondary negative DC bus 214 are regulated and symmetrical. The ground can be referenced to an electronics chassis (such as, for example, a chassis of a transportation vehicle).

[0025] One or more controllers 150 are used to control the front-end three-phase three-level T-type PFC circuit 110, the switch circuit 120 and / or the three-level T-type full-bridge output circuit 140. For example, one or more controllers can be used for the active split secondary split DC bus 170 to achieve optimized high voltage management of partial discharge hazards and optimized actuator design using the secondary positive DC bus 212 and the secondary negative DC bus 214. Equipped with the AC / DC converter 100, an advanced motor controller design with soft start technology can improve the inrush current control and disconnection function when the motor drive inverter fails. In addition, a modified space vector pulse width modulation (SVPWM) scheme is used to operate the front-end three-phase three-level T-type PFC circuit. Although the bidirectional isolated AC-DC converter is described as using a three-level three-phase topology, the bidirectional isolated AC-DC converter can also be used as a single-phase AC-DC power converter.

[0026] The primary three-phase three-level T-type power factor correction circuit 110 includes a primary AC bus 204, three inductors L1, L2 and L3, a first primary phase leg 205, a second primary phase leg 206, a third primary phase leg 207 and a primary split DC link bus 160. The primary AC bus 204 includes three AC voltage sources V1a, V1b and V1c.

[0027] The first primary phase leg 205 includes switching devices Q1R, Q2R, Q1RN and Q2RN. The second primary phase leg 206 includes switching devices Q3R, Q4R, Q3RN and Q4RN. The third primary phase leg 207 includes switching devices Q5R, Q6R, Q5RN and Q6RN. The first primary phase leg 205, the second primary phase leg 206 and the third primary phase leg 207 are connected to the primary capacitor midpoint of the primary split DC link bus 160 through Q1RN / Q2RN, Q3RN / Q4RN and Q5RN / Q6RN, respectively. The switching devices Q1R, Q2R, Q1RN and Q2RN of the first primary phase leg 205, the switching devices Q3R, Q4R, Q3RN and Q4RN of the second primary phase leg 206 and the switching devices Q5R, Q6R, Q5RN and Q6RN of the third primary phase leg 207 can be controlled to achieve different switching levels, as described more fully below.

[0028] The primary split DC link bus 160 includes capacitors C1P and C2P, a primary capacitor midpoint 202, and a primary DC negative reference voltage 203. The voltage of the primary capacitor midpoint 202 is actively controlled via a controller, such as by applying a PWM control signal.

[0029] The switch circuit 120 includes a first switch branch 208 and a second switch branch 209. The first switch branch 208 includes switch devices Q3P and Q4P. The second switch branch 209 includes switch devices Q1P and Q2P.

[0030] The CLLC resonant converter circuit 130 includes resonant capacitors Crp and Crs, resonant inductors Lrp and Lrs, and an isolation transformer T2 / 215. The resonant capacitor Crp and the resonant inductor Lrp are connected in series with each other, and the resonant capacitor Crs and the resonant inductor Lrs are connected in series with each other. The isolation transformer T2 / 215 includes a primary winding and a secondary winding. The primary winding includes terminals 1 and 2, and the secondary winding includes terminals 3 and 4. The resonant inductor Lrp is connected to terminal 1 of the primary winding, and the resonant inductor Lrs is connected to terminal 3 of the secondary winding. The resonant capacitor Crp is connected to the second switch branch 209 of the switching circuit at a point between the switching devices Q1P and Q2P. Terminal 2 of the primary winding is connected to the first switch branch 208 at a point between the switching devices Q3P and Q4P. The resonant inductor Lrp, the resonant inductor Lrs, and the isolation transformer T2 / 215 can be integrated into a single physical device. The CLLC resonant converter circuit operates to provide electrical isolation, voltage gain or reduction, and energy transfer. With the correct parameters, the CLLC resonant converter circuit can also achieve zero voltage switching of the power switch in the primary phase leg under forward power flow conditions and zero voltage switching of the power switch in the secondary phase leg under reverse power flow conditions.

[0031] The secondary T-type three-level circuit 140 includes a secondary split DC link bus 170, a first secondary phase leg 210, and a second secondary phase leg 211. The secondary split DC link bus 170 includes capacitors C1S and C2S, a secondary positive DC bus 212, a secondary capacitor midpoint 213, and a secondary negative DC bus 214. Figure 2 In the example of , the secondary positive DC bus 212 is associated with a positive low voltage DC power source (e.g., +135 VDC), the secondary capacitor midpoint 213 voltage is a secondary reference and is associated with the secondary capacitor midpoint 213, and the secondary negative DC bus 214 is associated with a negative low voltage DC power source (e.g., -135 VDC). The voltage of the secondary capacitor midpoint 213 is actively controlled via the controller 150, for example, by applying a PWM control signal.

[0032] The first secondary phase leg 210 includes switching devices Q1S, Q2S, Q1SN and Q2SN. The second secondary phase leg 211 includes switching devices Q3S, Q4S, Q3SN and Q4SN. The first secondary phase leg 210 and the second secondary phase leg 211 are connected to the secondary capacitor midpoint 213 of the secondary split DC link bus 170 through Q1SN / Q2SN and Q3S / Q4SN, respectively. The switching devices Q1S, Q2S, Q1SN and Q2SN of the first secondary phase leg 210 and the switching devices Q3S, Q4S, Q3SN and Q4SN of the second secondary phase leg 211 can be controlled to achieve different switching stages, as described more fully below.

[0033] When operating in the forward direction, the primary three-phase three-level T-type power factor correction circuit 110 receives an input voltage from the AC power bus 204, such as 115 Vrms, from the power bus of the vehicle. The controller 150 applies a control signal (e.g., a PWM control signal) to the switching devices Q1R-Q6R and Q1RN-Q6RN to apply power factor correction, producing a high-quality regulated DC voltage. The primary capacitor midpoint voltage regulation function (e.g., active midpoint primary voltage control) can be integrated into the PWM control signal, which determines the pulse duration and sequence of the phase terminals at each possible voltage level. The controller 150 applies a control signal (e.g., a PWM control signal) to the switching devices Q1P-Q4P to modulate the DC voltage into a high-frequency AC square wave. The isolation transformer 215 of the CLLC resonant converter circuit 130 reduces the high-frequency alternating voltage. The controller 150 applies a control signal (such as a PWM control signal) to the switching devices Q1S-Q4S and Q1SN-Q4SN of the secondary T-type three-level output circuit 140 to rectify the high-frequency alternating voltage into a relatively low VDC output voltage (such as, for example, ±135VDC or ±270VDC voltage) to power the low-voltage load. Under the forward power flow condition, the switching devices Q1P-Q4P perform zero-voltage switching through the CLLC resonant conversion circuit. Similarly, under the reverse power flow condition, the switching devices Q1S-Q4S perform zero-voltage switching.

[0034] The voltage regulation function of the secondary capacitor midpoint 213 (such as active midpoint secondary voltage control) can be integrated into the PWM control signal. Since the isolation transformer 215 is present in the resonant conversion process, the secondary capacitor midpoint 213 can be connected to the ground referenced to the electronics chassis. In this example, when the secondary capacitor midpoint 213 is referenced to the electronics chassis, the result is an internal power bus of ±135VDC.

[0035] When operating in reverse, the secondary T-type three-level circuit 140 output receives an input VDC voltage from a low voltage source, such as, for example, a ±135 VDC voltage. The controller 150 applies a control signal (such as a PWM control signal) to the switching devices Q1S-Q4S and Q1SN-Q4SN to modulate the ±135 VDC voltage into a high-frequency AC square wave. The isolation transformer 215 of the CLLC resonant converter circuit steps up the high-frequency alternating voltage. The controller applies a control signal (such as a PWM control signal) to the switching devices Q1P-Q4P of the switching circuit 120 to rectify the high-frequency alternating voltage into a relatively high VDC output voltage. The controller 150 applies a control signal (such as a PWM control signal) to the switching devices Q1R-Q6R and Q1RN-Q6RN to invert the DC voltage into an AC output voltage, thereby providing power to the AC power bus 204.

[0036] Figure 3An exemplary analytical model of a front-end three-phase three-level T-type PFC circuit with circuit variable definitions is illustrated. Figure 3 The switch variable S shown in a , S b , S c The definition is as follows:

[0037]

[0038] make Then and

[0039]

[0040] One or more controllers utilize SVPWM switches of active power devices (e.g., gallium nitride high electron mobility transistors (GaN HEMTs)) of the front-end three-phase three-level T-type PFC circuit to synthesize VR according to the current regulation requirements. a 、vr b 、vr c The synthesis process can be mathematically represented in space vector format. That is, the rectifier voltage space vector It can be defined by the following equation:

[0041] in and

[0042]

[0043] Utilize the base Will Converting to unit voltage gives:

[0044] in

[0045] Compared with the conventional space vector representation, the embodiment of the present application uses a modified space vector representation which takes into account the capacitor midpoint deviation variable v or vpu, that is, the DC voltage distribution between the capacitors is taken into account. The resulting switching space vector diagram is shown in Figure 4 As shown in , where it is assumed that v ≥ 0 and all vectors are composed of triples [S a S b S c ] mark. Information about the current flowing into the midpoint of the capacitor is also marked on Figure 4 switch in space vector illustration.

[0046] Compared with traditional space vector diagrams, Figure 4The intermediate vector of the switching space vector diagram is dispersed due to the uneven distribution of capacitor voltage. Therefore, in order to synthesize the required voltage vector, the nearest four vector (NFV) technology is generally used. The described configuration structure thus performs the modified SVPWM scheme as a simplified NFV technology and NTV technology. For example, the exemplary modified SVPWM scheme can be performed as described below.

[0047] Assumptions Figure 3 The two capacitors C1 and C2 in have equal capacitance, i.e. C1 = C2 = C. Then And by applying sliding mode control using the sliding surface {v=0}, or by direct inspection based on the Lyapunov stability criterion, the middle two vectors aligned in the same direction are weighted differently so that i0 has a sign that favors adjusting v to zero.

[0048] Figure 5 The diagram shows the use of Figure 4 An exemplary voltage space vector synthesis of a modified SVPWM scheme for a high-fidelity switching space vector diagram. In order to synthesize the voltage vector NFV is identified as First, find where k is the design freedom allocated to the capacitor midpoint voltage regulator. This is possible because and causes i0 to have the opposite sign. Once is determined, the remaining tasks are based on Implementing NTV synthesis:

[0049]

[0050]

[0051] In the front-end three-phase three-level T-type PFC circuit, the AC phase current is normally aligned with the voltage vector, i.e., a displacement angle of 0 degrees for positive power flow, or a displacement angle of 180 degrees for negative power flow. Therefore, the role of the intermediate space vector in the capacitor midpoint voltage control process is clear.

[0052] Figure 6 The AC / DC front-end PFC stage control loop structure using an exemplary NFV SVPWM is summarized. A dual-carrier based PWM with a variable carrier amplitude to reflect the actual voltage profile on the output capacitor can be used to implement an equivalent SVPWM algorithm of the sine-triangle PWM approach. Figure 7 The variable amplitude dual carrier sine triangle PWM details are shown and the capacitor bank midpoint voltage deviation (in V) is visualized. pu≤0 is an example) on the pulse width timing parameters. PWM control of the DC / DC converter stage is described in the applicant's international patent application PCT / US2023 / 64008, which is incorporated herein by reference.

[0053] Accordingly, the bidirectional isolated AC-DC converter uses a three-level output topology with active midpoint voltage control and an isolation transformer to split the output into symmetrical dual supplies with a common reference connected to the chassis. This arrangement produces an optimized solution for system-level partial discharge hazard management and enables standard ±135VDC or ±270VDC flight control actuation systems to interface to aircraft and other vehicle applications. Figure 4 A high-fidelity switching space vector diagram, a modified SVPWM scheme is used to operate the front-end three-phase three-level T-type PFC circuit to eliminate the input phase current distortion caused by the capacitor midpoint voltage deviation. The isolated topology of the CLLC resonant converter circuit allows the output to be soft-started with the required inrush current control during the sudden application of the input voltage. In some embodiments, the bidirectional isolated AC-DC converter is implemented with advanced health monitoring functions, and if necessary, the bidirectional isolated AC-DC converter can safely disconnect the motor drive inverter from the relevant power bus when the motor drive inverter fails. In addition, in the event of one of the phases being lost, the bidirectional isolated AC-DC converter can operate based on a single-phase AC-DC algorithm. In addition, the bidirectional isolated AC-DC converter can operate standard flight control actuators without regenerative energy being dissipated in the box, that is, the regenerative energy can be sent back to the source. An exemplary transistor that can be used by the bidirectional isolated AC-DC converter includes a GaN HEMT, which provides relatively high power density and high efficiency.

[0054] Figure 8 An exemplary flow chart of a method 800 for synthesizing one or more rectified voltages of a front-end three-phase three-level T-type PFC circuit via a modified space vector pulse width modulation (SVPWM) scheme is illustrated. At step 801 and step 802, the voltage value at each of the three branches a, b and c is At step 803, the modified SVPWM switching space vector diagram is used to find the voltage vector The four closest vectors to and At step 804 and step 805, the parameter k is calculated based on the requirement of the medium voltage controller of the PFC capacitor bank. At step 806, based on and A similar NTV synthesis is implemented. A similar algorithm flow chart can also be drawn for a variable amplitude dual carrier sine triangle PWM implementation of an exemplary three-level PFC stage control. In an exemplary embodiment, the rectifier voltage space vector is obtained by means of an inverse Clarke transformation and by means of a common mode voltage containing a control parameter k to achieve the primary capacitor bank midpoint voltage. Generate phase branch rectifier voltage command or

[0055] Although the present invention has been shown and described for a specific one or more embodiments, it is obvious that, after reading and understanding this specification and the accompanying drawings, other technical personnel in the field will think of equivalent changes and modifications. In particular, with respect to the various functions implemented by the above-mentioned elements (components, assemblies, devices, compositions, etc.), unless otherwise specified, the terms used to describe such elements (including references to "means / apparatus") are intended to correspond to any element that implements the specified function of the element (i.e., functionally equivalent), even if the structure is not equivalent to the disclosed structure that implements the function in one or more exemplary embodiments of the present invention illustrated in this article. In addition, although the specific features of the present invention may be described above for only one or more of the illustrated embodiments, such features may also be combined with one or more other features of other embodiments, such as for any given or specific application It would be desirable and advantageous.

Claims

1. A bidirectional AC-DC converter, comprising: A three-phase power factor correction (PFC) circuit including a primary split DC link bus; a switching circuit connected to a primary split DC link bus, wherein the three-phase power factor correction circuit and the switching circuit constitute a primary side of the bidirectional AC-DC converter; a capacitor-inductor-inductor-capacitor (CLLC) resonant converter circuit connected to the switching circuit, the capacitor-inductor-inductor-capacitor resonant converter circuit including an isolation transformer, and a three-level T-type full-bridge output circuit connected to the capacitor-inductor-inductor-capacitor resonant converter circuit on the secondary side of the bidirectional AC-DC converter, the three-level T-type full-bridge output circuit including a secondary split DC link bus; The isolation transformer electrically isolates the primary side of the bidirectional AC-DC converter from the three-level T-type full-bridge output circuit on the secondary side bidirectional AC-DC converter.

2. The bidirectional AC-DC converter according to claim 1, wherein: The primary split DC link bus includes a primary midpoint bus associated with a midpoint voltage; and The bidirectional AC-DC converter also includes a controller configured to actively control a midpoint voltage associated with the primary midpoint bus.

3. The bidirectional AC-DC converter according to any one of claims 1 to 2, wherein: The primary split DC link bus includes a primary positive DC bus and a primary negative DC bus that are symmetrical to each other.

4. The bidirectional AC-DC converter according to claim 2-3, wherein: The secondary split DC link bus includes a secondary midpoint bus associated with a secondary midpoint voltage; and The controller is also configured to actively control a secondary midpoint voltage associated with the secondary midpoint bus.

5. The bidirectional AC-DC converter according to any one of claims 1 to 4, wherein: The secondary split DC link bus includes a secondary positive DC bus and a secondary negative DC bus that are symmetrical to each other.

6. The bidirectional AC-DC converter according to any one of claims 4 to 5, wherein: The secondary midpoint bus is connected to ground referenced to the electronics chassis.

7. The bidirectional AC-DC converter according to any one of claims 1 to 6, wherein: The switching circuit includes a plurality of transistors arranged in a full-bridge configuration.

8. The bidirectional AC-DC converter according to any one of claims 1 to 7, wherein: The three-phase power factor correction (PFC) circuit includes three inductors and a three-level T-type phase leg topology capable of bidirectional power flow.

9. The bidirectional AC-DC converter according to any one of claims 1 to 8, wherein: The T-type full-bridge output circuit includes a plurality of transistors configured to switch between three output levels.

10. The bidirectional AC-DC converter according to any one of claims 1 to 9, wherein: The T-type full-bridge output circuit interfaces with standard ±135VDC or ±270VDC flight control actuation systems for aircraft AC power systems.

11. The bidirectional AC-DC converter according to any one of claims 1 to 10, wherein: The three-phase PFC circuit includes a plurality of gallium nitride-based transistors configured to implement power factor correction.

12. The bidirectional AC-DC converter according to any one of claims 1 to 11, wherein: The three-level T-type full-bridge output circuit includes a plurality of gallium nitride-based transistors.

13. The bidirectional AC-DC converter of any one of claims 1-12, further comprising a controller configured to implement a modified space vector pulse width modulation (SVPWM) scheme to control the three-phase PFC circuit, the modified space vector pulse width modulation scheme being at least partially based on a nearest four vector technique.

14. The bidirectional AC-DC converter according to any one of claims 1 to 13, A controller is included that is configured to safely disconnect the motor drive inverter from an associated power bus upon a motor drive inverter failure.

15. The bidirectional AC-DC converter of any one of claims 1-14, comprising a controller configured to operate according to a single-phase AC-DC algorithm in the event of a loss of one of the three phases.

16. The bidirectional AC-DC converter according to any one of claims 1 to 15, wherein: The three-level T-type full-bridge output circuit performs soft start using inrush current control during sudden application of input voltage.

17. The bidirectional AC-DC converter according to any one of claims 1 to 16, wherein: The bidirectional isolated AC-DC converter operates standard flight control actuators without dissipating regenerative energy.

18. A method for controlling a front-end three-phase three-level T-type power factor correction (PFC) circuit, comprising: The rectified voltage of the front-end three-phase three-level T-type PFC circuit is synthesized via a modified space vector pulse width modulation (SVPWM) scheme; Therein, the modified space vector pulse width modulation scheme is at least partially based on the nearest four vector technique.

19. The method for controlling a front-end three-phase three-level T-type power factor correction (PFC) circuit according to claim 18, further comprising controlling a PWM switching function of the phase branch to have a first state, a second state and a third state.

20. The method for controlling a front-end three-phase three-level T-type power factor correction (PFC) circuit according to claim 19, wherein: The primary split DC link bus includes a primary positive DC bus, a midpoint bus, and a primary negative DC bus that are symmetrical to each other; Therein, in the first state, the phase leg is connected to the positive DC bus.

21. A method for controlling a front-end three-phase three-level T-type power factor correction (PFC) circuit according to any one of claims 19-20, in, In the second state, the phase legs are connected to the midpoint bus.

22. A method for controlling a front-end three-phase three-level T-type power factor correction (PFC) circuit according to any one of claims 19 to 21, in, In the third state, the phase leg is connected to the negative DC bus.

23. A method for controlling a front-end three-phase three-level T-type power factor correction (PFC) circuit according to any one of claims 18 to 22, in, The front-end three-phase three-level T-type power factor correction (PFC) circuit has three branches, wherein each of the three branches has a different branch voltage.

24. The method for controlling a front-end three-phase three-level T-type power factor correction (PFC) circuit according to claim 23, in, The rectifier voltage space vector is calculated using three switching functions associated with the three branches, the DC bus voltage and the capacitor midpoint deviation voltage, and is generated from all switching function combinations collectively from the switching space vector diagram of the front end PFC.

25. The method for controlling a front-end three-phase three-level T-type power factor correction (PFC) circuit according to claim 24, in, The rectifier voltage space vector is synthesized by means of the modified space vector pulse width modulation by switching the space vector diagram with the primary capacitor bank midpoint voltage control.

26. A method for controlling a front-end three-phase three-level T-type power factor correction (PFC) circuit according to any one of claims 24-25, Also included is determining a closest first vector, a closest second vector, a closest third vector, and a closest fourth vector, wherein: Two of the closest first, second, third or fourth vectors having intermediate lengths in the switching space vector diagram cause primary capacitor bank midpoint charging currents i having opposite signs. o .

27. The method for controlling a front-end three-phase three-level T-type power factor correction (PFC) circuit according to claim 26, in, The two closest vectors that result in currents having opposite signs are used to calculate the combined closest vector.

28. The method for controlling a front-end three-phase three-level T-type power factor correction (PFC) circuit according to claim 27, in, A standard nearest three-vector PWM synthesis is performed on the combined closest vector and the other two closest vectors.

29. A method for controlling a front-end three-phase three-level T-type power factor correction (PFC) circuit according to any one of claims 24 to 28, in, The rectifier voltage space vector is synthesized by a variable amplitude dual carrier sine-triangle PWM with primary capacitor bank midpoint voltage control.

30. The method for controlling a front-end three-phase three-level T-type power factor correction (PFC) circuit according to claim 29, Also includes the voltage distribution difference of the primary capacitor bank by modulation The carrier amplitude is used to generate multiple variable amplitude dual-carrier waveforms. in, The positive carrier and load waves have amplitudes that reflect the ongoing DC bus voltage distribution between the primary high-side capacitor and the primary low-side capacitor.

31. A method for controlling a front-end three-phase three-level T-type power factor correction (PFC) circuit according to any one of claims 18 to 30, It also includes the modified space vector pulse width modulation and the associated variable amplitude dual carrier sine triangle PWM to operate the front end three-phase three-level T-type power factor correction circuit so that the input phase current distortion caused by the capacitor midpoint voltage deviation is eliminated.

32. A method for controlling a front-end three-phase three-level T-type power factor correction (PFC) circuit according to any one of claims 18 to 31, The invention also includes the step of converting the rectifier voltage space vector by means of an inverse Clarke transformation and by means of a common mode voltage including a control parameter k for realizing the midpoint voltage of the primary capacitor bank. Generate phase branch rectifier voltage command or 33. The method for controlling a front-end three-phase three-level T-type power factor correction (PFC) circuit according to claim 32, The method further includes generating a plurality of phase branch switching functions by combining the phase branch rectified voltage commands The three-level sine-triangle PWM process is performed by comparison with the dual carrier waveform.

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