SIR-based radio frequency microstrip filter with independently reconfigurable order and frequency

Abstract

The invention relates to the technical field of reconfigurable filters, in particular to an order and frequency independent reconfigurable radio frequency microstrip filter based on an SIR, a band-pass filter of which the center frequency and the order can be independently adjusted is designed based on an asymmetric structure of the SIR, and a variable capacitance diode and an adjustable coupling structure are designed and loaded by analyzing the coupling structure; then, a coupling structure of the feed and the input / output resonator is adjusted, such as tap feed, slot feed and feed position, in addition, adjustment can be carried out by changing the capacitance value of the variable capacitance diode, so that a proper external quality factor value is obtained, and the feed design is completed; and finally, through continuous simulation optimization of electromagnetic simulation software such as electromagnetic simulation software and circuit simulation software, the design of the radio frequency microstrip filter with independently reconfigurable order and frequency based on the SIR is realized.

Description

Technical Field

[0001] The invention relates to the technical field of reconfigurable filters, and in particular to a radio frequency microstrip filter whose order and frequency are independently reconfigurable based on SIR. Background Art

[0002] The research on RF reconfigurable filters at home and abroad mainly focuses on frequency reconfiguration, bandwidth reconfiguration and simultaneous frequency and bandwidth reconfiguration, and is developing towards the direction of reconfiguration of both order and frequency.

[0003] At present, the common research on filter order and frequency reconfiguration includes: using more lumped components or combining lumped and distributed resonators to achieve the reconstruction function, etc. Using more lumped components will lead to large insertion loss, large volume, and poor harmonic suppression effect.

[0004] Due to the above reasons, the existing order and frequency reconfigurable filters are large in size, have large insertion loss and return loss, and have poor harmonic suppression effects. Summary of the invention

[0005] The purpose of the present invention is to provide a radio frequency microstrip filter whose order and frequency are independently reconfigurable based on SIR, aiming to provide a microstrip filter whose order and frequency are independently reconfigurable and which has small size, is easy to process, has small insertion loss and return loss, and has certain suppression of harmonics.

[0006] To achieve the above-mentioned purpose, the present invention provides a SIR-based order and frequency independently reconfigurable radio frequency microstrip filter, including four 1 / 4 wavelength SIR resonators, each of which is composed of a short-circuited grounded section and an open-circuited section of transmission line, with a certain coupling gap spacing between each other; the four resonators respectively use SIRs with different impedance ratios, and when the four resonators have the same fundamental frequency, the spurious frequencies are different and suppress each other.

[0007] There are two independent signal paths, and the signal transmission path is changed by adjusting the coupling between the resonators to achieve order reconstruction.

[0008] The coupling between the resonators is adjusted using a loaded varactor diode, the high impedance end of the SIR resonator is loaded with a varactor diode, and the boundary with the strongest magnetic field of the resonator is loaded with two varactor diodes.

[0009] Among them, different center frequencies can be adjusted by adjusting the capacitance of the varactor diode.

[0010] The invention provides a radio frequency microstrip filter whose order and frequency are independently reconfigurable based on SIR, and a bandpass filter whose center frequency and order are independently adjustable based on SIR asymmetric structure. Firstly, a loading varactor diode and an adjustable coupling structure are designed by analyzing the coupling structure, and then the coupling structure of the feeding and input / output resonator is adjusted, such as tap feeding, slot feeding and the position of the feeding. In addition, the varactor diode capacitance can be changed for further adjustment to obtain a suitable external quality factor value, thereby completing the feeding design. Finally, electromagnetic simulation software, circuit simulation software and other electromagnetic simulation software are used for continuous simulation optimization to realize the design of the radio frequency microstrip filter whose order and frequency are independently reconfigurable based on SIR. BRIEF DESCRIPTION OF THE DRAWINGS

[0011] In order to more clearly illustrate the embodiments of the present invention or the technical solutions in the prior art, the drawings required for use in the embodiments or the description of the prior art will be briefly introduced below. Obviously, the drawings described below are only some embodiments of the present invention. For ordinary technicians in this field, other drawings can be obtained based on these drawings without paying creative work.

[0012] Figure 1 It is a schematic diagram of the SIR filter coupling structure of the present invention.

[0013] Figure 2 It is a structural schematic diagram of a SIR-based order and frequency-independently reconfigurable radio frequency microstrip filter of the present invention.

[0014] Figure 3 It is a schematic diagram of the 1 / 4 wavelength SIR structure.

[0015] Figure 4 It is a schematic diagram of the relationship between the impedance ratio and the normalized stray resonant frequency according to a specific embodiment of the present invention.

[0016] Figure 5 It is a schematic diagram of a curve showing a change in the resonant frequency of a resonator along with a varactor diode according to a specific embodiment of the present invention.

[0017] Figure 6 It is a schematic diagram of the distribution of the resonator coupled electromagnetic field at 2.5 GHz in a specific embodiment of the present invention.

[0018] 6(a) Schematic diagram of the electric field distribution between resonator 1 and resonator 2.

[0019] 6(b) Schematic diagram of the magnetic field distribution between resonator 1 and resonator 4.

[0020] 6(c) Schematic diagram of the electric field distribution between resonator 1 and resonator 4.

[0021] 6(d) Schematic diagram of the magnetic field distribution between resonator 1 and resonator 2.

[0022] Figure 7 It is a schematic diagram of a curve showing a change in the resonator coupling coefficient as a function of the coupling gap and the capacitance of the varactor diode according to a specific embodiment of the present invention.

[0023] 7(a) Schematic diagram of the relationship between the resonator coupling coefficient and the coupling gap.

[0024] 7(b) Schematic diagram of the variation of coupling coefficient with the size of the varactor diode (S12=0.45mm, S34=0.45mm, S14=1.9mm).

[0025] Figure 8 It is a schematic diagram of a coupling structure and a curve showing a change in coupling strength with different varactor loading positions according to a specific embodiment of the present invention.

[0026] 8(a) Schematic diagram of the coupling structure of resonator 1 and resonator 2.

[0027] 8(b) Schematic diagram of the variation of the coupling strength between resonator 1 and resonator 2 with the loading position of the varactor.

[0028] 8(c) Schematic diagram of the coupling structure of resonator 3 and resonator 4.

[0029] 8(d) is a graph showing the variation of the coupling strength between resonators 3 and 4 with the loading position of the varactor.

[0030] Fig. 9 It is a schematic diagram of the change of group delay with the capacitance of the varactor diode in a specific embodiment of the present invention.

[0031] Fig.10 It is a simulation model diagram of the filter according to a specific embodiment of the present invention.

[0032] Fig.11 It is a schematic diagram of S parameter response curves of SIR-based and UIR-based filters at a frequency of 2.5 GHz according to a specific embodiment of the present invention.

[0033] Fig.12 Schematic diagram of the second-order simulated transmission response of the filter in a specific embodiment of the present invention, (a) insertion loss; (b) return loss.

[0034] Fig.13 It is a schematic diagram of S parameters of the fourth-order simulated transmission response of the filter in a specific embodiment of the present invention.

[0035] Fig.14 1 is a schematic diagram comparing the second-order simulation and test transmission responses of the filter according to a specific embodiment of the present invention, (a) insertion loss; (b) return loss.

[0036] Fig.15 1 is a schematic diagram comparing the fourth-order simulation and test transmission responses of the filter according to a specific embodiment of the present invention, (a) insertion loss; (b) return loss. DETAILED DESCRIPTION

[0037] Embodiments of the present invention are described in detail below, examples of which are shown in the accompanying drawings, wherein the same or similar reference numerals throughout represent the same or similar elements or elements having the same or similar functions. The embodiments described below with reference to the accompanying drawings are exemplary and are intended to be used to explain the present invention, and should not be construed as limiting the present invention.

[0038] The following is an explanation of some of the English term abbreviations that appear in this application:

[0039] SIR: stepped-impedance resonator, stepped impedance resonator.

[0040] See also Figures 1 to 3 The present invention provides a SIR-based order and frequency independently reconfigurable radio frequency microstrip filter, including four 1 / 4 wavelength SIR resonators, each of which is composed of a short-circuited grounded and open-circuited transmission line, with a certain coupling gap spacing between each other; the four resonators respectively use SIRs with different impedance ratios, and when the four resonators have the same fundamental frequency, the spurious frequencies are different and suppress each other.

[0041] Two independent signal paths are set up through two and four resonators.

[0042] The coupling between the resonators is adjusted using loaded varactor diodes. The high impedance end of the SIR resonator is loaded with a varactor diode, and the strongest magnetic field boundary of the resonator is loaded with two varactor diodes.

[0043] Different center frequencies can be adjusted by adjusting the capacitance of the varactor diode.

[0044] The present invention is further described below in combination with the design ideas and the structure of the specific embodiments:

[0045] The SIR filter coupling topology of the present invention is as follows Figure 1 As shown, the solid line is magnetic coupling and the dotted line is electric coupling.

[0046] For the second-order filter response, two resonators are used for signal transmission, and the coupling between resonators 1 and 2 and resonators 3 and 4 should be zero. Since these couplings cannot be perfectly tuned to zero, the resonant frequencies of resonators 2 and 3 should be tuned out of the center frequency of the filter to minimize the impact on the second-order response. The coupling matrix of the second-order response is shown in Equation 1:

[0047]

[0048] For a second-order response, the coupling (∈) between resonators 1 and 2 and between resonators 3 and 4 should be as small as possible. and can be larger or smaller than the center frequency so that resonators 2 and 3 do not affect the second-order response. That is, resonators 2 and 3 should be detuned. In addition, it is preferred that and have different signs. As long as ∈ is small and and They are all very big. The value of can be changed within a certain range with little effect on the filter response.

[0049] For the 4th-order Chebyshev response, the coupling between resonators 1 and 4 should be zero. Although the coupling coefficient of the proposed tunable coupling structure varies greatly, the coupling coefficient cannot be reduced to zero. Here, the cross-coupling between resonator 1 and resonator 4 can be used to form transmission zeros on both sides of the passband. Since the proposed filter structure is an asymmetric structure, there is also a certain cross-coupling between resonator 1 and resonator 3. The reconfigurability of the filter coupling mechanism can be used to adjust other couplings to compensate for non-zero cross-coupling and form an equal ripple response. There is cross-coupling between resonator 1 and resonator 4. Zero points are formed on both sides of the filter passband, and its coupling matrix is ​​formula 2:

[0050]

[0051] Actual coupling coefficient (m ij ) and the normalized coupling coefficient are as shown in formula 3:

[0052]

[0053] Where BW is the design bandwidth and f0 is the center frequency. In the subsequent actual design of the filter, the actual coupling matrix can be calculated from the normalized coupling matrix according to the filter index to be designed, and the corresponding design optimization can be performed according to the actual coupling matrix. Because there are sign differences in different electromagnetic couplings, the designed SIR order and frequency independent reconfigurable filter is an asymmetric structure, as shown in the following figure. Figure 2 shown.

[0054] Furthermore, the quarter-wavelength SIR has a smaller size, which is suitable for compact design, and its second harmonic is located near the third harmonic, which helps to improve the out-of-band suppression capability. The resonance characteristics and spurious frequencies of the λg / 4 type SIR are analyzed in detail. Figure 3As shown in Figure 1, a quarter-wavelength SIR consists of a short-circuited and grounded transmission line and an open-circuited transmission line, where the characteristic impedances of the short-circuited end and the open-circuited end are recorded as Z1 and Z2, respectively, and the equivalent electrical lengths are θ1 and θ2, respectively. A core parameter of the SIR is the ratio of the characteristic impedances of the two transmission lines, that is, the ratio of Z2 to Z1, which is defined as follows (Formula 4):

[0055]

[0056] The step characteristics between the two transmission lines and the edge discontinuity of the open end may cause parasitic effects, however, these effects can usually be ignored during analysis. Observed from the open end, the input impedance is Equation 5:

[0057]

[0058] When resonant, Z in tends to ∞, so we can get:

[0059] Z2-Z1 tanθ1tanθ2

[0060] Thus we can get:

[0061]

[0062] The above formula shows that the resonance condition of SIR is not only affected by the electrical length, but also related to its characteristic impedance, which provides additional adjustment freedom for the design. The harmonic resonance frequency of SIR is also the result of the combined effect of the length of the transmission line and the impedance ratio K, and these harmonic frequencies have a direct impact on the stopband performance of the filter.

[0063] Assuming the fundamental resonant frequency is f0, for a quarter-wavelength SIR, the first harmonic frequency is denoted by f s1 , and its corresponding electrical length is denoted as θ s1 Usually, the electrical lengths of the two transmission lines are made the same in the design, that is, θ1 = θ2 = θ0. The first stray resonance frequency f can be obtained by formula 6: s1 The relationship (Formula 7):

[0064]

[0065] Therefore, further deduction can be obtained that the first stray resonant frequency is (Equation 8):

[0066]

[0067] Figure 4 is the distribution of the normalized stray resonant frequency relative to f0 under different impedance ratios K. In the design process, it is usually chosen to make f s1As far away from the fundamental frequency as possible. The four resonators in the present invention respectively use SIRs with different impedance ratios. When the fundamental frequencies of the four resonators are the same, the spurious frequencies will be different. The spurious frequencies of the four resonators suppress each other, so that the harmonics of the filter are suppressed.

[0068] In order to realize the frequency reconfiguration of the filter, the present invention loads a varactor diode at the high impedance end of the SIR. Adjusting the capacitance of the varactor diode can change the electrical length of the resonator, thereby changing the resonant frequency of the resonator. Figure 5 As shown, as the capacitance of the varactor increases, the resonant frequency of the resonator decreases.

[0069] Then comes the design of the adjustable coupling structure. The comprehensive reconfigurable coupling matrix shows that the main coupling coefficients are all positive. Therefore, when selecting the coupling structure between the two resonators, electrical coupling or magnetic coupling can be used. In the present invention, Figure 6 (a) and Figure 6 (b) shows the electric field distribution and magnetic field distribution between resonator 1 and resonator 2. Because all four resonators use SIR, the electromagnetic field strength distribution between other adjacent resonators is consistent. Two varactor diodes are loaded at the strongest magnetic field boundary of the resonator to adjust the coupling strength. For the cross coupling between resonator 1 and resonator 4, electrical coupling is realized, and its electromagnetic field distribution is shown in Figure 6 (c) and Figure 6 (d) As shown. At the boundary where the electric field strength of the two resonators is the highest, two varactor diodes are loaded to adjust the coupling strength. By adjusting the capacitance of the varactor diode, the electrical coupling between the two resonators can be adjusted, thereby achieving a change in the coupling coefficient.

[0070] According to the mapping principle between the main coupling coefficient and the tuning element, in order to construct the adjustable main coupling coefficient described in the reconfigurable coupling matrix in the actual physical structure, an intermediate variable, namely the coupling coefficient between the resonators, must be determined. This coupling coefficient serves as a bridge between the physical entity and the theoretical coupling matrix, laying a theoretical foundation for the further design of adjustable bandpass filters.

[0071] pass Figure 7 From the observation of (a), it can be seen that the coupling strength between resonators decreases as the coupling gap increases. If the coupling gap is fixed and the capacitance of the variable capacitance diode loaded between the resonators is changed, the coupling strength between the resonators will also change. Figure 7(b), as the capacitance values ​​of the varactor diodes C12 and C34 increase, the electrical coupling strength between the two resonators gradually increases. When the magnetic coupling strength remains unchanged, the total coupling coefficient decreases. As the capacitance value of the varactor diode C14 increases, the electrical coupling strength between resonator 1 and resonator 4 gradually increases. Since the cross coupling between resonator 1 and resonator 4 is electrical coupling, the total coupling coefficient increases. It should be noted that the coupling coefficient between the resonators is different when the varactor diode is loaded. This is because the field distribution is different at different positions of the resonator, so the energy coupling between the resonators will also be different. Figure 8 It can be seen that as the position of the varactor changes, the coupling coefficient between the two resonators also changes.

[0072] According to the design requirements, at different target center frequencies, the corresponding varactor diode capacitance can be obtained by directly viewing the chart, which provides convenience and accuracy in the subsequent tuning of the coupling strength of the filter.

[0073] The input / output coupling describes the interconnection between the first and last resonators of the filter and the external circuit. The tightness of the coupling can be measured by the external quality factor Q ext If Q ext If the value of is small, the filter will be undercoupled; on the contrary, when the external quality factor Q ext When Q is too high, the filter appears over-coupled, which affects the efficient transfer of energy. Improper coupling structure can cause energy reflection. Optimizing Q ext The value helps to achieve reflection-free energy transfer, which manifests itself in filter performance as passband flatness, low insertion loss, and low return loss.

[0074] Q ext The expression of is shown in formula 9:

[0075]

[0076] BW represents the 3dB bandwidth of the filter, and g0 and g1 are the component values ​​after frequency conversion. From Equation 9, we can see that Q ext It is proportional to the center frequency of the filter and inversely proportional to the 3dB bandwidth. The center frequency and order of the filter designed by the present invention are reconfigurable, so f0 is a variable. During the center frequency reconstruction process, the value on the right side of equation 9 will change. Given that the input / output coupling structure is constant, the corresponding Q ext Therefore, in Equation 9, the equations on both sides are no longer balanced, which may cause Q ext The mismatch causes the filter to be insufficiently coupled or over-coupled.

[0077] In order to maintain the validity of equation 9 in the case of center frequency reconstruction, the present invention loads a varactor diode in the feed line of the input / output coupling structure so that Q ext becomes an adjustable parameter.

[0078] The input / output coupling coefficient can be calculated using the frequency domain method and the group delay method. The present invention uses the group delay method. The following describes the process of calculating the input / output coupling coefficient using the group delay method. You can use:

[0079]

[0080] The group delay τ is expressed as:

[0081]

[0082] At resonance (ω=ω0), the group delay in equation 11 reaches a maximum value:

[0083]

[0084] The optimal input / output coupling configuration can be determined by using the group delay technique based on the reflection parameter S11. Group delay and external quality factor Q ext The group delay reaches its maximum at the resonant frequency. By simulating the S11 parameters of a single resonator with a varactor diode using electromagnetic simulation software, the characteristics of the group delay can be obtained. The simulation results are shown in Fig. 9 middle.

[0085] The results show that integrating capacitive coupling into the input / output coupling structure can significantly improve the external quality factor Q of the resonator. ext Fine-tune. By adjusting the variable capacitance diode C ext The capacitance value can maintain a suitable Q during the adjustment of the center frequency and bandwidth of the filter. ext The required external quality factor can be adjusted according to different center frequencies.

[0086] Finally, a three-dimensional model of the tunable filter was created in the electromagnetic simulation software (the model structure is as follows Fig.10), and performed parameter sweep optimization to achieve the best performance. Then, the part representing the lumped element was removed from the 3D model in the electromagnetic simulation software, and the remaining ports were defined as lumped ports. An electromagnetic simulation software model with n ports was generated and exported as a SnP file, which was then imported into the circuit simulation software. In the circuit simulation software environment, capacitors and inductors were connected at each port, and their values ​​were set as adjustable parameters for optimization. Finally, in the circuit simulation software, the previously constructed varactor diode model was used to replace the capacitor, and the filter center frequency was successfully adjusted by changing the voltage across the varactor diode.

[0087] For further information, see Figures 11 to 15 Parameters and simulation comparison description:

[0088] Fig.11 It is a schematic diagram of S parameter response curves of SIR-based and UIR-based filters at a frequency of 2.5 GHz according to a specific embodiment of the present invention.

[0089] Fig.12 The figure is a schematic diagram of the second-order simulated transmission response of the filter in a specific embodiment of the present invention, (a) insertion loss; (b) return loss. It can be seen from the simulation results that the second-order reconstruction frequency range of the filter is 2.4GHz-2.6GHz, the bandwidth is 200MHz, the insertion loss is better than 1.1dB, and the return loss is better than 14dB.

[0090] Fig.13 It is a schematic diagram of S parameters of the fourth-order simulated transmission response of the filter in a specific embodiment of the present invention.

[0091] Fig.14 1 is a schematic diagram comparing the second-order simulation and test transmission responses of the filter according to a specific embodiment of the present invention, (a) insertion loss; (b) return loss.

[0092] Fig.15 1 is a schematic diagram comparing the fourth-order simulation and test transmission responses of the filter according to a specific embodiment of the present invention, (a) insertion loss; (b) return loss.

[0093] The filter of the present invention is based on SIR and can suppress the harmonics of the filter. The simulation results show that by properly adjusting the sizes of C1, C2, C3 and C4 as well as the coupling varactor and the feeding varactor diode, the fourth-order passband center frequency of the filter can be tuned within the range of 2.1-2.7 GHz, and the second-order passband center frequency can be tuned within the range of 2.4-2.6 GHz while maintaining the passband quality.

[0094] Through repeated simulation verification, the actual size of the final product obtained in this specific embodiment is as follows: the overall size of the filter is 20mm×25mm. A RogersRO4350 substrate is used, with a thickness of h=0.508mm, a relative dielectric constant ∈r=3.66, a loss tangent angle tanδ=0.004, and a surface metal thickness of 35μm. The corresponding physical size parameters are: l1=4.14mm, l2=13.3mm, l3=14mm, l4=6.47mm, l5=15.11mm, l6=4.1mm, l7=14.35mm, l8=6.8mm, l9=10mm, w1=1.93mm, w2=0.5mm, w3=1mm, w4=2.5mm, w5=1.1mm, w6=1.1mm, s12=0.35mm, s23=0.5mm, s34=0.71mm, s14=1.9mm.

[0095] In summary, the beneficial effects of the present invention are as follows: when designing a filter with reconfigurable order and frequency, an asymmetric SIR resonator structure is introduced, and four 1 / 4 wavelength SIR resonators are composed of a short-circuited grounded section and an open-circuited section of transmission line, and there is a certain coupling gap spacing between the resonators; the four resonators respectively use SIRs with different impedance ratios to suppress the harmonics of the filter. The coupling between the resonators is adjusted using a loaded varactor diode, thereby changing the signal transmission path and realizing order reconstruction. The capacitance of the varactor diode is adjusted to adjust different center frequencies to achieve frequency reconstruction.

[0096] The test results show that the second-order response of the reconfigurable filter based on SIR frequency and order can be reconstructed in the range of 2.4GHz-2.6GHz, the insertion loss varies between 1.95dB-2.65dB, and the return loss varies between 13.79dB-18.97dB. The fourth-order response can be reconstructed in the range of 2.1GHz-2.7GHz, the insertion loss varies between 2.52dB-5.78dB, and the return loss varies between 10.32dB-20.41dB.

[0097] What is disclosed above is only a preferred embodiment of the present invention, and it certainly cannot be used to limit the scope of rights of the present invention. Ordinary technicians in this field can understand that all or part of the processes of the above embodiment and equivalent changes made according to the claims of the present invention still fall within the scope of the invention.

Claims

1. A SIR-based order and frequency-independent reconfigurable radio frequency microstrip filter, characterized in that: It includes 4 1 / 4 wavelength SIR resonators, each of which is composed of a short-circuited grounded transmission line and an open-circuited transmission line, with a certain coupling gap interval between them; the four resonators respectively use SIRs with different impedance ratios, and when the four resonators have the same fundamental frequency, the spurious frequencies are different and suppress each other.

2. The SIR-based order and frequency-independent reconfigurable RF microstrip filter according to claim 1, characterized in that: It has two independent signal paths, and the signal transmission path is changed by adjusting the coupling between the resonators to achieve order reconstruction.

3. The SIR-based order and frequency-independent reconfigurable RF microstrip filter as claimed in claim 2, characterized in that: The coupling between the resonators is adjusted using loaded varactor diodes. The high impedance end of the SIR resonator is loaded with a varactor diode, and the strongest magnetic field boundary of the resonator is loaded with two varactor diodes.

4. The SIR-based order and frequency-independent reconfigurable RF microstrip filter as claimed in claim 3, characterized in that: Different center frequencies can be adjusted by adjusting the capacitance of the varactor diode.

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