A single-stage current-fed bidirectional DC-AC converter and a fixed-frequency control method thereof

Abstract

The application belongs to the technical field of DC-AC converters, and relates to a single-stage current-fed bidirectional DC-AC converter and a fixed-frequency control method thereof. m The power transmission characteristics of the circuit are described, and a method for realizing the feedforward control of the AC side current is described; the application can realize fixed-frequency linear phase-shifting control in a super-wide DC side voltage range and can realize bidirectional real-time power flow control in a wide DC side voltage range; the application does not need a complex control loop design theory; therefore, the application effectively reduces the cost and power consumption of the controller, and has a higher power density upper limit and a higher efficiency upper limit.

Description

TECHNICAL FIELD

[0001] The present application belongs to the technical field of DC-AC converter, and relates to a single-stage current-fed bidirectional DC-AC converter and a fixed-frequency control method thereof. BACKGROUND

[0002] Small DC-AC converters provide a connection mode from low-voltage DC voltage (level between 20V-60V) to grid voltage (220VAC), which is widely demanded in photovoltaic micro-inverters, vehicle-mounted battery chargers (below 60V), UPS (uninterruptible) power supply systems and household and outdoor emergency power supplies and various occasions. In order to provide sufficient safety for individual users, electrical isolation needs to be performed in this type of DC-AC converter. Since the DC-AC converter needs to have both step-up and DC-AC conversion functions, a two-stage conversion architecture is mostly used in early schemes. This architecture is similar to the non-isolated string inverter scheme, except that the non-isolated Boost converter is replaced by an isolated DC-DC converter (such as LLC, DAB) converter. The architecture scheme classification of the DC-AC converter in the prior art is shown in Figure 1 .

[0003] The quasi-single-stage architecture is to realize DC-AC conversion without high-voltage DC bus through the sinusoidal output voltage control of the isolated DC-DC. The single-stage architecture is the current research focus. The flyback single-stage DC / AC topology is widely used in early micro-inverter schemes. However, because the transformer utilization rate is low, the power level cannot meet the demand of the existing single photovoltaic panel (400W or more). In addition, the household distributed energy storage system and the vehicle-mounted charger need to realize bidirectional power transmission from the battery to the grid and from the grid to the battery. Therefore, the single-stage high-frequency AC chain scheme with bidirectional power flow capability has become the mainstream research and product selection scheme. Its feature is to transfer energy through the phase difference of the high-frequency AC voltage square wave on the high-frequency transformer. Specifically, it is realized by using bidirectional switches on the AC side full-bridge based on the DAB converter for AC working condition application.

[0004] Under low-voltage DC input, the AC side structure is simplified to the DAB phase-shifted scheme of half-bridge, which has higher cost and performance ratio. It is similar to the core control idea of the traditional DAB type DC-DC converter, which directly realizes the control of flowing power by using phase-shifted modulation.

[0005] However, the DC-AC converter in the prior art has the following defects:

[0006] 1. The two-stage scheme has the disadvantages of complex circuit structure, high cost, and lower upper limit of power density and efficiency, although it is mature in technology and simple in control.

[0007] 2. The quasi-single-stage solution offers some optimization in efficiency and cost compared to the former due to the reduction in the number of high-frequency switching devices and decoupling capacitors. However, the unchanged number of devices means that the power density of the converter does not have a significant advantage over the two-stage solution. Moreover, the same wide voltage gain operating range as the single-stage solution makes its implementation significantly more difficult than the two-stage solution.

[0008] 3. The single-stage solution (flyback type), although simple in structure and low in cost, is limited by its asymmetrical transformer structure and significant high voltage and current stress, resulting in power ratings that cannot meet current requirements. Furthermore, bidirectional power flow cannot be achieved in this technology, meaning its application is limited to unidirectional DC-AC conversion scenarios such as photovoltaic micro-inverters.

[0009] 4. High-frequency AC chain-type solutions characterized by 12 components (including a full-bridge AC converter) have too many components, facing the same dilemma as quasi-single-stage solutions. Furthermore, for low-voltage applications, the cost is too high and the control is complex.

[0010] 5. The dual active bridge (DAB) scheme for real-time power control using phase-shift modulation (total 8 devices) currently mainly uses a voltage-fed DAB circuit topology. However, the periodic variation of the DAB voltage gain from 0°C at the power frequency makes converter control difficult. Furthermore, the voltage-fed DAB scheme has discontinuous input current and a narrow voltage range on the DC side—a disadvantage when connected to photovoltaic panels or cells.

[0011] Therefore, a DC-AC converter with a simple circuit topology, a simple control scheme, and a wider soft-switching operating range, and its control method are needed to solve the above-mentioned technical problems. Summary of the Invention

[0012] The purpose of this invention is to address the problems of existing technologies by proposing a current-fed DAB (Digital-Area Bridge) solution. The topology in this invention uses two independent half-bridges instead of a full-bridge on the DC side to avoid circulating current on the DC input side under the proposed fixed-frequency high-linearity control strategy. This invention achieves an extremely wide DC-side voltage application range, twice that of voltage-fed solutions.

[0013] The technical solution adopted by this invention to solve the technical problem is: a single-stage current-fed bidirectional DC-AC converter, comprising:

[0014] On the DC side, the input terminal is connected to the DC bus V. dc The DC side consists of two inductor-equipped Boost half-bridges; the Boost half-bridge includes: a first DC-side half-bridge and a second DC-side half-bridge;

[0015] On the AC side, the output terminal of the AC side is connected to the AC bus V.ac ; the AC side is composed of an AC side half-bridge and an AC side filter; the AC side half-bridge comprises: an AC side bidirectional half-bridge;

[0016] a transformer, a primary side of the transformer being connected to an output end of the DC side, a secondary side of the transformer being connected to an input end of the AC side; the transformer is a high-frequency transformer, and a transformation ratio of the transformer is 1:n;

[0017] a control device, the control device being used to control the DC side and the AC side to cross a conductance G m a method for realizing control of the AC side current in a feedforward manner by describing power transmission characteristics of the circuit; the control sampling signals of the control device comprise: a DC side port voltage, a DC side inductor current, an AC side port current and a port voltage.

[0018] Preferably, the DC side comprises: switching transistors Q1-Q4, inductors L dc1 and L dc2 , output capacitors C dc1 and C dc2 , the inductor L dc1 is a first path DC side filter inductor, the inductor L dc2 is a second path DC side filter inductor, the first path DC side half-bridge is composed of the switching transistors Q1 and Q2, the second path DC side half-bridge is composed of the switching transistors Q3 and Q4, the output capacitor C dc1 is a first path DC side clamping capacitor, the output capacitor C dc2 is a second path DC side clamping capacitor, and the connection mode of the DC side is that: a positive input terminal of a DC bus v dc is connected to input ends of the inductors L dc1 and L dc2 , an output end of the inductor L dc1 is connected to one end of the primary side of the transformer, the other end of the primary side of the transformer is connected to an output end of the inductor L dc2 , the switching transistors Q1 and Q2 are connected to two ends of the output capacitor C dc1 in series, and a connection node of the switching transistors Q1 and Q2 is connected to the output end of the inductor L dc1 , the switching transistors Q3 and Q4 are connected to two ends of the output capacitor C dc2 in series, and a connection node of the switching transistors Q3 and Q4 is connected to the output end of the inductor L dc2 , and negative poles of the output capacitors C dc1 and C dc2 are connected to a negative pole of the DC bus v dc .

[0019] the AC side comprises: switching transistors S1-S4, inductors L ac and L lk , series capacitors C ac1 and Cac2 Inductor L ac For AC side filter inductance, inductance L lk For transformer cascade inductance, capacitor C ac1 and C ac2 The AC side is connected in series with a capacitor, and the AC side bidirectional half-bridge consists of switching transistors S1 to S4; the AC side connection method is: AC bus v ac The positive input terminals are connected in series with inductor L. ac Series capacitor C ac1 Series capacitor C ac2 Connected to AC bus V ac The negative input terminal; the two ends of the switching transistors S1 to S4 are connected in series and then connected to the inductor L respectively. ac With series capacitor C ac1 Connection nodes, AC bus V ac The negative input terminal; the series inductance L at the connection node between switching transistors S2 and S4. lk The other end of the transformer secondary side is connected to one end of the transformer secondary side, and the other end of the transformer secondary side is connected to the series capacitor C. ac1 and C ac2 Connection nodes; mixed frequency band filter circuits filter out DC, 100Hz and frequency components above 200kHz.

[0020] This invention also discloses a fixed-frequency control method for a single-stage current-fed bidirectional DC-AC converter. This control method is used to control the aforementioned DC-AC converter. The control method includes: dividing the converter into eight regions based on the DC-side duty cycle D and phase shift ratio φ, and primarily operating it on the linear positive maximum power output trajectory located in regions II and IV; the control quantity relationship is expressed as:

[0021]

[0022] In equation (4), D represents the duty cycle of the DC-side device and φ represents the phase shift angle ratio.

[0023] Preferably, the converter is divided into 8 regions based on the DC-side duty cycle D and the phase shift ratio φ, and operates completely on a continuous bidirectional trajectory located in regions II, III, and IV, based on the minimum duty cycle D. min The phase-shift control mode serves as a replacement for the PWM mode during low-power transmission, improving the accuracy of low-power control.

[0024] More preferably, the control method uses transconductance G m The complete and unique control trajectory described over the wide DC-side voltage range is expressed as follows:

[0025]

[0026] In formula (6), k m represents the characteristic coefficient of the circuit.

[0027] More preferably, in the control method, the converter is controlled by G m The control trajectory is the core of the AC current control loop combining feedforward and feedback, and specifically includes the following steps:

[0028] First, the phase reference of the grid voltage is obtained by the PLL phase-locked loop, and then the control set current reference i ac,ref is generated; after a slight correction by the PR controller, G m is obtained, which is the current reference value required by the controller;

[0029] Second, the real-time control transconductance G m value is obtained according to the input current reference value and the sampled DC side voltage value; the polarity of the value is determined by the polarity of the grid voltage v ac ;

[0030] Third, the control amount D and φ are obtained, and the driving PWM of Q1 to Q4 and S1 to S4 is generated.

[0031] Preferably, the control method adopts a proportional control link for the DC side current i Ldc effective in a specific frequency band to suppress possible LC resonance.

[0032] The beneficial effects of the present application are:

[0033] 1. The present application can perform fixed-frequency linear phase-shift control in a wide DC side voltage range; the present application can realize bidirectional real-time power flow control in a wide DC side voltage range. The voltage range (20-60V) applied to the DC side can be about twice that of the traditional voltage feeding scheme (30-50V). However, the control method does not become complex because of the wide DC side voltage range. The only and fixed transconductance G m control trajectory is applied to feedforward predictive control, which only needs a slight PR feedback for correction, so it does not need complex control loop design theory. At the same time, the linearization degree of the control trajectory is as high as 98.7%, and it is applied to fixed switching frequency, so the controller does not need strong computing power and memory space, thus the present application effectively reduces the cost and power consumption on the controller.

[0034] 2. The present application has a higher and fixed switching frequency level; the present application works at a fixed switching frequency of 100kHz or more. The magnetic components and capacitors of the converter can be designed at a switching frequency of 100kHz or more; therefore, the present application has a higher upper limit of power density.

[0035] 3, The present application has smaller high-frequency current ripple level at both DC side and AC side; the present application can achieve bidirectional real-time power flow control in a wider DC side voltage range. This real-time power control can bring smaller high-frequency current ripple at the AC side, and the interleaved input inductance connection mode at the DC side also makes the high-frequency current ripple of the DC side current i Ldc significantly smaller than the voltage-fed scheme. Therefore, the continuous DC side current of the present application can be applied to the charge and discharge management of the battery or the MPPT technology of the photovoltaic module.

[0036] 4, The present application has a single-stage few-switch device structure and a wide ZVS operating range; the present application is a single-stage structure, and the number of switching devices is only 8, which is simple in structure; it has high realizability under the control scheme of simple fixed frequency and high linearity; fewer switching devices and wider soft switching range make the present application have higher upper limit of efficiency. BRIEF DESCRIPTION OF DRAWINGS

[0037] Figure 1 is a prior art isolated DC / AC converter technical scheme diagram;

[0038] Figure 2 is a single-stage current-fed bidirectional DC-AC converter and its fixed-frequency control method of the present application;

[0039] Figure 3 is a specific cross-conduct-based current controller implementation block diagram of the present application;

[0040] Figure 4 is a key voltage and current waveform diagram of the present application in 8 different operating regions;

[0041] Figure 5 is a G m control trajectory diagram of the present application, wherein Figure 5 (a) is a complete control trajectory represented in 8 different regions, Figure 5 (b) is the corresponding G m (from -0.1S to 0.1S) and control quantity correspondence diagram;

[0042] Figure 6 is a time ratio k zvs analysis diagram of the present application under rated voltage soft switching (ZVS), wherein Figure 6 (a) corresponds to Q1 and Q3, Figure 6 (b) corresponds to Q2 and Q4, Figure 6 (c) corresponds to S1 and S4;

[0043] Figure 7 is a simulation waveform diagram of the key voltage and current of the converter under rated operating condition of the present application (bidirectional 400W);

[0044] Figure 8 is the effect verification comparison chart of the DC side current control under the rated operating condition of the present application (forward 400W);

[0045] Figure 9 is the experimental verification chart of the zero voltage turn-on (ZVS) of the switching device Q2 under the rated operating condition of the present application, wherein Figure 9 (a) forward 400W, Figure 9 (b) reverse 400W;

[0046] Figure 10 is the complete control block diagram of the present application under the other two use methods, wherein Figure 10 (a) off-grid inverter mode, Figure 10 (b) PFC rectifier mode.

[0047] In the figure, 11, inductance L dc1 ; 12, inductance L dc2 ; 21, first DC side half-bridge; 22, second DC side half-bridge; 31, capacitor C dc1 ; 32, capacitor C dc1 ; 4, transformer; 5, inductance L lk ; 61, capacitor C ac1 ; 62, capacitor C ac2 ; 7, AC side bidirectional half-bridge; 8, inductance L ac ; 9, hybrid frequency band filter circuit. DETAILED DESCRIPTION

[0048] The related technologies in the present application will be described clearly and completely below in combination with the drawings in the embodiments of the present application. Obviously, the described embodiments are only some of the embodiments of the present application, but not all the embodiments. Based on the embodiments in the present application, all the other embodiments obtained by those skilled in the art without creative labor fall within the scope of protection of the present application.

[0049] With reference to Figures 1 to 10 , in the present embodiment, Figure 2 a single-stage current-fed bidirectional DC-AC converter topology and its control method are shown. The circuit realizes electrical isolation by a high-frequency transformer. The DC side is composed of two identical boost half-bridges (Q1, Q2 and Q3, Q4), whose input inductances L dc1 and L dc2 are connected in parallel, and output capacitors C dc1 and C dc2 are independent. The two half-bridges work staggered by 180 degrees, and the switching nodes a and b are connected to the two ports of the transformer primary side. The AC side is composed of a half-bridge (S1 to S4) applying bidirectional switches and a series capacitor Cac1 and C ac2 The composition, whose connection node is connected to two ports of the transformer secondary side. Finally, it is connected to the AC power grid through the AC filter inductor L ac and the power grid (AC power supply).

[0050] In terms of control, in one aspect, the present application adopts a hybrid constant frequency real-time AC current control method based on PWM control and phase-shift control. Two control variables are the DC side device opening duty ratio D (lower tube Q2 and Q4) and the phase-shift angle ratio φ (from Q1 to S2), respectively. On the AC side, the switching devices exhibit the characteristics of high-frequency operation for half of the power frequency cycle and direct through for the other half of the cycle. When the grid voltage v ac is greater than 0, S2 and S3 are complementary high-frequency switching actions with a duty ratio of 50%, while S4 and S1 remain open. And v ac is less than 0, S4 and S1 high-frequency switch while S2 and S3 remain open. It is worth noting that: S4 and S2, S3 and S1 work symmetrically.

[0051] The controller determines the phase of the grid voltage through the phase-locked loop (PLL) link, and then generates the current reference i ac,ref . After a small correction, the equivalent transconductance G m required by the converter is determined using this reference value and the DC side voltage v dc , and then the corresponding control quantities D and φ are generated. Finally, 4 groups of PWM drive signals corresponding to the 8 switching devices are generated. m G

[0052]

[0053] where k m is the characteristic coefficient of the circuit, defined as n / (4L lk f s ).

[0054] Through G m , the output control of the wide range of DC side voltage v dc input AC side current i ac can be realized. Positive values represent forward power transmission under DC-AC conversion, and negative values represent reverse power transmission under AC-DC conversion.

[0055] On the other hand, in addition to the current control loop based on G m , the DC side current loop in a specific frequency band is also used to suppress the LC resonance phenomenon on the DC side. The two Boost units on the DC side will have the risk of LC resonance, and the resonance center frequency f r is:

[0056]

[0057] where L dc is the total DC side inductance in μH; C dc is the total DC side capacitance in μF.

[0058] Table 1: Reference key design parameters (for simulation verification)

[0059] Circuit parameter Value Operating parameter Value Direct current side filter inductance L dc1 and L dc2 ]]> 18 μH RMS value of AC measurement voltage 220V rms ]]> Direct current side capacitor C dc1 and C dc2 ]] 30 μF AC side voltage frequency 50 Hz Transformer ratio n 4 DC side voltage range 20-60V Transformer inductance L 1k ]]> 85 μH Rated power 400W AC side filter inductance L ac ]] 18 μH Switching frequency 100 kHz

[0060] Under the reference design parameters given in Table 1, the resonance occurs in the frequency range of 2-6 kHz. Therefore, the total DC side inductance current i Ldc may contain the frequency components: DC component, 100 Hz, 2-6 kHz resonance component and high frequency component above 200 kHz. The present application uses a hybrid frequency band filter to extract the resonance component in the DC side current based on this, to form a DC side current proportional control loop that is only valid for the resonance frequency band. The DC side input current has very small high frequency current ripple, and the calculation formula is:

[0061]

[0062] Figure 3 The specific block diagram of the AC current control loop based on G m is shown. The current control is mainly feedforward and supplemented by feedback. The initial current reference value is slightly corrected by a PR (or PI) controller to generate the current real-time reference value, and then the absolute value of G m is calculated according to formula (1). The polarity of G m is determined by the polarity of the grid side voltage v ac . The value of G m is set by f(D, φ) to control the corresponding trajectory point set with an accuracy of 0.0001 S. In the present application, the range of G m is set to -0.1 S to 0.1 S. Among the corresponding 2000 trajectory points, the points that can be directly calculated according to the linear relationship account for 98.7% of the total number. Therefore, only 13 trajectory points need to be stored in the processor memory in advance.

[0063] Difference explanation from existing similar topologies (for circuit topology)

[0064] The present application replaces the full bridge in the traditional DAB with a double-path input interleaved parallel boost circuit in the DC side. The DC input current is continuous and low ripple by L dc1 and L dc2 , which reduces the demand for DC side filters. The characteristics of the present application are:

[0065] 1) In the present application, the DC side only has one external interface, without high voltage DC portFigure 2 Mid C dc1 and C dc2 During operation, the present invention exhibits varying C dc1 and C dc2 voltage.

[0066] 2) The DC side in the present invention is composed of two independent half-bridges, and the outputs are not in parallel. The technical purpose of this change in the present invention is to avoid unnecessary power frequency circulating current (which will increase the loss) at the DC side under the control strategy proposed in the present invention.

[0067] Basic principles of control operation

[0068] Table 2: Power transmission partition and calculation

[0069]

[0070] Figure 4 The key voltage and current waveforms of all 8 working states of the converter in the present invention on the switching frequency time scale are shown. The duty cycles of the two Boost half-bridges on the DC side are exactly the same, so their voltage and current only differ in phase. The working states are divided into A and B classes according to the current waveforms of the DC side inductors L dc1 , L dc2 . And they are divided into I-IV types according to the current waveforms of the transformer cascade inductors L lk . The power transmission characteristics of different working states are shown in Table 2.

[0071] The fixed frequency high linearity control scheme used in the present invention makes it run in the IIIA region, the IIA region and the IIB region (DC-AC conversion) or the IIIA region, the IVA region and the IVB region (AC-DC conversion). The main control method is PWM control, which satisfies the following formula:

[0072]

[0073] It shows that the PWM control in the present invention is carried out at the maximum phase-shifted power transmission point under a fixed duty cycle. The linear relationship between the AC output current and the control quantity can be obtained:

[0074]

[0075] The linear control relationship allows the current controller to achieve relatively accurate output through feedforward control, which effectively reduces the design pressure of the feedback controller. However, when the output current is too low (including the zero-crossing commutation process), the power transmission cannot be effectively controlled through the duty cycle D due to the existence of dead time. At the same time, the current change rate is the largest in the entire cycle at this time, so the commutation process often appears waveform distortion. From positive power flow to negative power flow, the Gm The trajectory also shows a discontinuous change in control variable at 0, which is an unstable factor for energy control.

[0076] Figure 5 (a) shows the complete bidirectional current control trajectory after adding the transition section based on phase-shift control. The phase-shift control section is at the minimum duty ratio D min The PWM linear control section connecting the IIA and IVA regions is performed on the circuit design, and the bootstrap is used to reduce the power supply of the two isolated drives. The final full control trajectory can be divided into five sections, G m The expression is expressed in (6):

[0077]

[0078] A. Section 1 - PWM linear control section (positive);

[0079] B. Section 2 - Phase-shift nonlinear control section (positive);

[0080] C. Section 3 - Phase-shift linear control section (including power transmission zero point);

[0081] D. Section 4 - Phase-shift nonlinear control section (negative);

[0082] E. Section 5 - PWM linear control section (negative).

[0083] Among them, sections 1 and 5 are PWM control modes, and 2-4 are phase-shift control modes.

[0084] Reference Figure 5 (b), the equivalent G m The corresponding relationship between the control variable D and φ is shown, ranging from -0.1S to 0.1S. Among them, only sections 2 and 4 are nonlinear control sections, and the duty ratio is only 1.3%. Therefore, the control method adopted by the present application has a linear degree of 98.7%, greatly reducing the design difficulty of the control loop and the requirement of the microprocessor chip on the hardware.

[0085] The upper and lower limits of the nonlinear section are defined as G m,12 and G m,23 :

[0086]

[0087] The present application has a wide ZVS region under the fixed-frequency high-linearity control method. The ZVS judgment formula of Q1 and Q3, Q2 and Q4, and AC side devices S1-S4 is equivalent, and the judgment formula is as follows:

[0088]

[0089] In the formula ton,1 the turn-on time of Q1, t on,2 the turn-on time of Q2, t on,s the turn-on time of S2, t

[0090] Figure 6 The ZVS effect diagram in the full DC side voltage range and the full power output range is given. Here k zvs represents the ZVS time proportion in one power frequency cycle, 1 represents full time period soft switching. The results show that the DC side device is more than 75% of the full time period full soft switching working condition range, and the AC side reaches 100%.

[0091] To verify the technical effect of the scheme, simulation and experiment are carried out according to the parameters given in table 1.

[0092] Figure 7 The input voltage v rms , input current i dc , capacitor voltage v Ldc , inductor current i cdc1 and AC side voltage and current waveforms v Llk and i ac of the application under rated voltage (DC side 40V, AC side 220V ac and rated power (400W) are shown. It can be verified that it has very small high-frequency current ripple on the DC side and low-voltage side, and the THD of the AC side current is only 2.45% at rated full load.

[0093] Figure 8 The two-port voltage and current waveforms before and after enabling the DC side current (in a specific frequency range) control of the application are shown. It can be seen that before the DC side control is enabled, the LC oscillation described in formula (2) occurs on the DC side, and the frequency is between 2-6kHz. It is reflected in the inductor current i Ldc , capacitor voltage v cdc1 and v cdc2 . After enabling the DC side control, the resonance is suppressed. This will eliminate the audible noise caused by the inductor on the DC side, and at the same time improve the THD level of the AC side current.

[0094] Figure 9 The experimental waveform measurement results of the gate-source voltage v gs and the drain-source voltage v ds of the device Q2 of the application under rated full load working condition are shown. The results show that it has zero voltage turn-on (ZVS) effect in half of the power frequency cycle, which is consistent with the theoretical analysis.

[0095] Figure 10 The complete control structure of the connection method of the converter applied to off-grid inverters and PFC rectifiers is shown.Figure 2 For grid-connected bidirectional DC / AC, only current control is involved.

[0096] (1) Off-grid inverter - active support of AC voltage

[0097] To supply single-phase AC load with DC voltage source (20V-60V), the converter actively constructs 220V AC grid through AC voltage outer loop, and the internal current control is still the control method proposed in this invention.

[0098] (2) Rectifier (with active power factor correction function, i.e. PFC)

[0099] To supply DC load with wide voltage (20V-60V) from grid, the converter controls DC side voltage through DC voltage outer loop, and the internal current control is still the control method proposed in this invention. Under this rectification operation, the input current and input voltage are in phase, so it belongs to active power factor correction circuit (PFC).

[0100] Fixed-frequency linear phase-shifted control in super-wide DC side voltage range

[0101] The current-fed DAB converter adopted in this embodiment can achieve bidirectional real-time power flow control in a very wide DC side voltage range under the proposed high-linearity transconductance control strategy. The voltage range applied to the DC side (20-60V) can be twice as high as that of the traditional voltage-fed scheme (30-50V). However, the control method does not become complex because of the wide DC side voltage range.

[0102] The unique and fixed transconductance Gm control trajectory correspondence is applied to feedforward predictive control, which only needs a small PR feedback to correct, so it does not need complex control loop design theory. At the same time, the linearization degree of the control trajectory is as high as 98.7%, and it is applied to fixed switching frequency, so the controller does not need strong computing power and memory space - which can effectively reduce the cost and power consumption on the controller.

[0103] This embodiment has a higher and fixed switching frequency level

[0104] The control method adopted in this embodiment makes the bidirectional DC / AC converter work at a fixed switching frequency of 100kHz or higher. The magnetic components and capacitors of the converter can be designed at a switching frequency of 100kHz or higher. For example, under the reference design parameters in Table 1, the design frequency of the transformer and the cascaded inductance Llk is 100kHz, and the rest of the inductance and capacitance is 200kHz. Therefore, the design scheme of the bidirectional DC / AC converter proposed in this embodiment has a higher upper limit of power density.

[0105] The embodiment has a smaller high-frequency current ripple level (both DC side and AC side)

[0106] The current-fed DAB converter adopted by the embodiment can realize bidirectional real-time power flow control in an extremely wide DC side voltage range under the proposed fixed-frequency high-linearity cross-regulation control strategy. The real-time power control can bring a smaller high-frequency current ripple on the AC side, and simulation and experiments show that the THD under full load is below 3%. The input inductance connection mode of the DC side interleave parallel connection also significantly reduces the high-frequency current ripple of the DC side current i Ldc Compared with the voltage-fed scheme. Continuous DC side current can be applied to battery charge and discharge management or photovoltaic module MPPT technology.

[0107] The embodiment has a single-stage few-switch device structure and a wide ZVS operating range

[0108] The current-fed bidirectional DC-AC converter adopted by the embodiment is a single-stage structure, and the number of switching devices is only 8, which is simple. Under the simple fixed-frequency and high-linearity control scheme, it has high realizability. Fewer switching devices and wider soft switching range make the converter have a higher efficiency upper limit.

[0109] In summary, the application can perform fixed-frequency linear phase-shift control in an extremely wide DC side voltage range, and can realize bidirectional real-time power flow control in a wide DC side voltage range. The application does not require complex control loop design theory. Therefore, the application effectively reduces the cost and power consumption of the controller, and has a higher power density upper limit and a higher efficiency upper limit.

[0110] It should be emphasized that: the above is only the preferred embodiment of the application, not any form of limitation on the application, any simple modification, equivalent change and modification of the above embodiment according to the technical essence of the application still belongs to the scope of the technical solution of the application.

Claims

1. A single-stage current-fed bidirectional DC-AC converter, characterized by, Comprise: a direct current side, an input end of the direct current side being connected to a direct current bus v dc ; The DC side consists of two inductive Boost half-bridges, the Boost half-bridge comprises: the first DC side half-bridge (21), the second DC side half-bridge (22); on the AC side, the output end of the AC side is connected to an AC bus v ac ; the AC side is composed of an AC side half-bridge and an AC side filter, the AC side half-bridge comprises: an AC side bidirectional half-bridge (7); The transformer (4), the primary side of the transformer (4) is connected to the output end of the DC side, the secondary side of the transformer (4) is connected to the input end of the AC side; a control device for controlling the direct current side and the alternating current side to cross the conductance G m The power transmission characteristics of the circuit are described, and a method for realizing the control of the alternating current side current in a feedforward manner is implemented; the control sampling signals of the control device include: the direct current side port voltage, the direct current side inductance current, the alternating current side port current, and the port voltage; The DC side includes: switching transistors Q1 to Q4, and inductor L. dc1 (11) and L dc2 (12) Output capacitor C dc1 (31) and C dc2 (32) The connection method of the DC side is: DC bus v dc The positive input terminal is connected to the inductor L dc1 (11) and L dc2 (12) input terminal, inductor L dc1 The output terminal of (11) is connected to one end of the primary side of the transformer (4), and the other end of the primary side of the transformer (4) is connected to the inductor L. dc2 (12) Output terminal; Switching transistors Q1 and Q2 are connected in series and then connected to the output capacitor C respectively. dc1 (31) The connection nodes of switching transistors Q1 and Q2 are connected to inductor L. dc1 (11) Output terminal; Switching transistors Q3 and Q4 are connected in series and then connected to the output capacitor C respectively. dc2 (32) The connection nodes of switching transistors Q3 and Q4 are connected to inductor L. dc2 (12) Output terminal; Output capacitor C dc1 (31) and C dc2 (32) The negative terminal is connected to the DC bus v dc The negative electrode; The AC side includes: switching transistors S1-S4, inductors L ac (8) and L lk (5), capacitors C ac1 (61) and C ac2 (62), and the AC side is connected in the following manner: the positive input terminal of the AC bus v ac is connected in series with the inductor L ac (8), the capacitor C ac1 (61), the capacitor C ac2 (62), and then connected to the negative input terminal of the AC bus v ac ; the switching transistors S1-S4 are connected in series, and then connected to the connection node of the inductor L ac (8) and the capacitor C ac1 (61), the negative input terminal of the AC bus v ac ; the connection node of the switching transistor S2 and the switching transistor S3 is connected in series with the inductor L lk (5), and then connected to one end of the secondary side of the transformer (4), and the other end of the secondary side of the transformer (4) is connected to the connection node of the capacitor C ac1 (61) and C ac2 (62).

2. A method for controlling a single-stage current-fed bidirectional DC-AC converter at a fixed frequency, characterized in that, The fixed frequency control method is used to control the DC-AC converter of claim 1, the control method comprises: dividing the converter into 8 regions with DC side duty ratio D and phase shift ratio φ, and mainly operating on the linear forward maximum power output track located in the II region and the IV region; The control quantity relationship is expressed as: In formula (4), D represents the on-duty ratio of the DC side device, and φ represents the phase shift angle ratio.

3. The method according to claim 2, wherein the method is a fixed frequency control method of a single-stage current-fed bidirectional DC-AC converter, characterized in that, The converter is divided into 8 regions with DC side duty ratio D and phase shift ratio φ, and is completely operated on the continuous bidirectional trajectory located in regions II, III and IV, to take the minimum duty ratio D min based phase shift control mode as the alternative of PWM mode at low power transmission, to improve the low power control accuracy.

4. The method according to claim 3, wherein the method is a fixed frequency control method of a single-stage current-fed bidirectional DC-AC converter, characterized in that, The control method uses a transconductance G m The complete and unique control trajectory over the wide DC-side voltage range is described as: In formula (6), k m represents a characteristic coefficient of the circuit.

5. The method according to claim 4, wherein the method is a fixed frequency control method of a single-stage current-fed bidirectional DC-AC converter, characterized in that, In the control method, the converter is in G m The control trajectory is the core to realize the combination of feedforward and feedback AC current control loop, and specifically includes the following steps: First step, get the phase reference of grid voltage by PLL, then generate the control setting current reference i ac,ref ; get G by implementing a small correction through PR controller m current reference value required by the controller; Secondly, the real-time control transconductance G is obtained according to the input current reference value and the sampled DC side voltage value; the polarity of the value is determined by the polarity of the grid voltage v m . ac . Third, according to the obtained control quantity D and φ, the driving PWM of Q1 to Q4 and S1 to S4 is generated.

6. The method according to claim 2, wherein The control method employs a proportional control element for the direct current i Ldc at the direct current side in the specific frequency range, in order to suppress possible LC resonance phenomena.

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