Magnetic balance control method and device of full-bridge DC-DC converter, storage medium and equipment

Abstract

The invention relates to the technical field of power electronics, in particular to a magnetic balance control method and device for a full-bridge DC-DC converter, a storage medium and equipment, and the method comprises the steps: triggering the sampling of the primary side current of a transformer at the midpoint moment of the conduction time of a switching tube Q1 and a switching tube Q4, and obtaining a current sampling value I1; at the midpoint moment of the conduction time of the switch tube Q2 and the switch tube Q3, triggering to sample the primary side current of the transformer to obtain a current sampling value I2; constructing a compensation amount according to the current sampling value I1 and the current sampling value I2; and introducing the compensation amount into a single voltage loop to control PWM output. Output voltage single voltage loop control is used, primary side transformer current is introduced for compensation, control is simple, parameters are easy to set, the problem of magnetic flux imbalance is solved, meanwhile, power supply cost and size are not increased, and actual application requirements can be better met.

Description

Technical Field

[0001] The present invention relates to the field of power electronics technology, and in particular to a magnetic balance control method, device, storage medium and equipment for a full-bridge DC-DC converter. Background Art

[0002] The statements in this section merely provide background information related to the present invention and do not necessarily constitute prior art.

[0003] In order to ensure the flux balance of the transformer, a traditional full-bridge DCDC converter generally connects a DC blocking capacitor in series on the primary side of the converter to suppress the DC voltage and current components from entering the transformer and causing the transformer core to saturate.

[0004] In addition to connecting a DC blocking capacitor in series, the prior art generally also adopts a peak current control method, and the primary current and the output voltage are controlled by a voltage and current dual loop, with the voltage loop as the outer loop and the current loop as the inner loop. The compensator can adopt a PI compensation method.

[0005] Regarding the above-mentioned series connection of a DC blocking capacitor or the use of voltage and current dual-loop control, the former will increase the cost and size of the power supply, increase losses, and reduce efficiency, while the latter introduces dual-loop control, which is complex and requires more parameters to be adjusted. Summary of the invention

[0006] In order to overcome the shortcomings of the background technology, the present invention provides a magnetic balance control method, device, storage medium and equipment for a full-bridge DC-DC converter, which uses a single voltage loop control of the output voltage and introduces the primary transformer current for compensation. The control is simple and the parameters are easy to adjust. While solving the problem of magnetic flux imbalance, it does not increase the power supply cost and volume, and can better meet the actual application needs.

[0007] To achieve the above object, the present invention provides the following technical solutions:

[0008] In a first aspect, a magnetic balance control method of a full-bridge DC-DC converter is provided, wherein the full-bridge DC-DC converter includes a transformer and a full-bridge inverter module arranged on the primary side of the transformer, wherein the full-bridge inverter module includes four switch tubes Q1, Q2, Q3, and Q4, and the method includes:

[0009] At the midpoint of the conduction time of the switch tube Q1 and the switch tube Q4, the primary current of the transformer is sampled to obtain a current sampling value I1;

[0010] At the midpoint of the conduction time of the switch tube Q2 and the switch tube Q3, the primary current of the transformer is sampled to obtain a current sampling value I2;

[0011] Constructing a compensation amount according to the current sampling value I1 and the current sampling value I2;

[0012] The compensation amount is introduced into the single voltage loop to control the PWM output.

[0013] Further,

[0014] The midpoint time of the on-time is determined by counting the counter.

[0015] Further,

[0016] The method of constructing a compensation amount according to the current sampling value I1 and the current sampling value I2 includes:

[0017] The current sampling value I1 and the current sampling value I2 are summed and then multiplied by the compensation coefficient K to obtain the compensation amount K*(I1+I2).

[0018] Further,

[0019] The absolute value of the compensation amount K*(I1+I2) is proportional to the degree of imbalance.

[0020] Further,

[0021] The compensation amount K*(I1+I2) is 0.1 times the output amount of the PI controller in the output voltage single voltage loop.

[0022] Further,

[0023] The step of introducing the compensation amount into a single voltage loop to control PWM output comprises:

[0024] The compensation amount is introduced after the PI controller in the single voltage loop.

[0025] Further,

[0026] After introducing the compensation amount into the PI controller in the single voltage loop, the method comprises:

[0027] In the positive half cycle when the switch tube Q1 and the switch tube Q4 are turned on, the output of the PI controller is subtracted from the compensation amount, and then sent to the PWM generator after being limited by the limiter to generate a PWM wave with a duty cycle D1 to control the switch tube Q1 and the switch tube Q4;

[0028] In the negative half cycle when the switch tubes Q2 and Q3 are turned on, the output of the PI controller is summed with the compensation amount, and then sent to the PWM generator after being limited by the limiter to generate a PWM wave with a duty cycle D2 to control the switch tubes Q2 and Q3.

[0029] In a second aspect, a magnetic balance control device for a full-bridge DC-DC converter is also provided, wherein the full-bridge DC-DC converter includes a transformer and a full-bridge inverter module arranged on the primary side of the transformer, wherein the full-bridge inverter module includes four switch tubes Q1, Q2, Q3, and Q4, and the device includes:

[0030] The current sampling module is used to trigger the sampling of the primary current of the transformer to obtain the current sampling value I1 at the midpoint of the conduction time when the switch tubes Q1 and Q4 are turned on; and is also used to trigger the sampling of the primary current of the transformer to obtain the current sampling value I2 at the midpoint of the conduction time when the switch tubes Q2 and Q3 are turned on;

[0031] A compensation amount building module, used for building a compensation amount according to the current sampling value I1 and the current sampling value I2;

[0032] The magnetic balance control module is used to introduce the compensation amount into the single voltage loop to control the PWM output.

[0033] Based on the same inventive concept, the present invention also provides a computer-readable storage medium storing one or more programs, which, when executed, can implement the aforementioned magnetic balance control method of the full-bridge DC-DC converter.

[0034] Based on the same inventive concept, the present invention also provides an electronic device, including a processor, a communication interface, a computer-readable storage medium as described above, and a communication bus; wherein the processor, the communication interface, and the computer-readable storage medium communicate with each other via the communication bus; and the processor is used to execute the program stored in the computer-readable storage medium.

[0035] Compared with the prior art, the present invention has the following beneficial effects:

[0036] The output voltage is controlled by a single voltage loop, and the primary transformer current is introduced for compensation, that is, a single voltage loop + primary positive and negative current difference compensation control is adopted. The control is simple, and the parameters are easy to adjust. It can suppress the magnetic flux imbalance problem caused by inconsistent device parameters or loop impedance. While solving the magnetic flux imbalance problem, it will not increase the power supply cost and volume, and can better meet the actual application needs.

[0037] Other features and advantages of the present invention will be described in the following description, and partly become apparent from the description, or understood by practicing the present invention. The purpose and other advantages of the present invention can be realized and obtained by the structures pointed out in the description, claims and drawings.

[0038] The present invention will be further described below in conjunction with the accompanying drawings. BRIEF DESCRIPTION OF THE DRAWINGS

[0039] In order to more clearly illustrate the embodiments of the present invention or the technical solutions in the prior art, the drawings required for use in the embodiments or the description of the prior art will be briefly introduced below. Obviously, the drawings described below are only some embodiments of the present invention. For ordinary technicians in this field, other drawings can be obtained based on these drawings without paying creative labor.

[0040] Figure 1 A schematic flow chart of a magnetic balance control method for a full-bridge DC-DC converter according to an embodiment of the present invention;

[0041] Figure 2 A topological diagram of a full-bridge DC-DC converter according to an embodiment of the present invention;

[0042] Figure 3 for Figure 2 The local topology diagram of the primary side of the transformer;

[0043] Figure 4 This is the traditional single voltage loop control block diagram of a full-bridge DC-DC converter;

[0044] Figure 5 A schematic diagram of the primary current of a transformer according to an embodiment of the present invention;

[0045] Figure 6 A single voltage loop control block diagram that introduces a compensation value K*(I1+I2) according to an embodiment of the present invention;

[0046] Figure 7 A schematic diagram of simulation results using a traditional control method;

[0047] Figure 8 A schematic diagram of simulation results of a single voltage loop control method using the introduction of compensation K*(I1+I2);

[0048] Fig. 9 A schematic structural diagram of a magnetic balance control device for a full-bridge DC-DC converter according to an embodiment of the present invention;

[0049] Fig.10 The figure is a schematic diagram of the structure of an electronic device according to an embodiment of the present invention. DETAILED DESCRIPTION

[0050] In order to make the purpose, technical solution and advantages of the implementation of the present invention clearer, the technical solution in the embodiment of the present invention will be clearly and completely described below in conjunction with the drawings in the embodiment of the present invention. Obviously, the described embodiment is only a part of the embodiment of the present invention, not all the embodiments. Based on the embodiments of the present invention, all other embodiments obtained by ordinary technicians in this field without creative work are within the scope of protection of the present invention.

[0051] like Figure 1 to Figure 2 As shown, the first embodiment of the present invention provides a magnetic balance control method for a full-bridge DC-DC converter, wherein the full-bridge DC-DC converter includes a transformer and a full-bridge inverter module arranged on the primary side of the transformer, wherein the full-bridge inverter module includes four switch tubes Q1, Q2, Q3, and Q4, and the method includes the following steps:

[0052] At the midpoint of the conduction time of the switch tube Q1 and the switch tube Q4, the primary current of the transformer is sampled to obtain a current sampling value I1;

[0053] At the midpoint of the conduction time of the switch tube Q2 and the switch tube Q3, the primary current of the transformer is sampled to obtain a current sampling value I2;

[0054] Constructing a compensation amount according to the current sampling value I1 and the current sampling value I2;

[0055] The compensation amount is introduced into the single voltage loop to control the PWM output.

[0056] In the above technical solution, sampling the primary current of the transformer can facilitate the construction of the compensation amount, and then introducing the constructed compensation amount into the output voltage single voltage loop for control, which can well realize the magnetic balance control of the full-bridge DC-DC converter. This solution uses the output voltage single voltage loop control and introduces the primary transformer current for compensation. The control is simple and the parameters are easy to adjust. While solving the problem of magnetic flux imbalance, it will not increase the power supply cost and volume, and can better meet the actual application needs.

[0057] The embodiment of the present invention provides a solution to the problem of flux imbalance in a DC-DC transformer (i.e., a full-bridge DC-DC converter of the present embodiment). Conventional full-bridge converters generally control the output voltage by controlling the duty cycle, such as Figure 2 and Figure 4 As shown, by sampling the output voltage and the given voltage, the difference is made, and then compensated by the PI controller and limited by the limiter, and then given to the PWM generator, a set of duty cycles with symmetry in the positive and negative half cycles and a phase difference of 180° is output to control the power device (that is, the output uses the PWM wave of the duty cycle to control the power device). Among them, the role of the limiter is to limit the maximum duty cycle. Due to the discreteness of the power device parameters, the aging drift of the resistor and capacitor parameters, and the inconsistent loop impedance, the positive and negative half-cycle conduction will be inconsistent, which will cause the magnetization curve of the transformer core to shift in one direction, eventually leading to magnetic saturation and power supply failure.

[0058] The above-mentioned full-bridge DC-DC converter belongs to the prior art and is only briefly described here. Figure 2 , Figure 3In the figure, the primary side of the transformer includes four switching tubes Q1, Q2, Q3, and Q4, which are used for inversion. T is a transformer, which is used for power transmission and voltage isolation. D1 and D2 are secondary side rectifier tubes, Lo is a filter inductor, and Co is a filter capacitor.

[0059] In some embodiments, the midpoint of the conduction time is determined by counting with a counter. In this embodiment, the midpoint of the conduction time of the corresponding switch tube can be conveniently and effectively determined by counting with a counter.

[0060] The above current sampling is further explained below in conjunction with a specific current waveform.

[0061] like Figure 5 As shown, Figure 5 The trapezoidal wave in the middle is the transformer primary current waveform (the current waveform here corresponds to Figure 2 The current Ir collected in the sample is positive and negative, and it is positive after sampling signal processing and AD conversion), where:

[0062] When Q1 and Q4 are turned on, the current increases linearly in the positive direction. When the counter counts to the midpoint of the on-time (the position indicated by the arrow above in the figure), the trigger current sampling is I1; here, the counter counts to the midpoint of the on-time, which means the midpoint of the on-time of Q1 and Q4. The trigger current sampling 11 here is to take the Ir value at that moment, and sampling is performed in each switching cycle.

[0063] When Q2 and Q3 are turned on, the current increases linearly in the negative direction. Similarly, when the counter counts to the midpoint of the on-time (the position indicated by the lower arrow in the figure), the trigger current sampling is I2; here, the counter counts to the midpoint of the on-time, which means the midpoint of the on-time of Q2 and Q3. The trigger current sampling I2 here is to take the Ir value at that moment, and sampling is performed in each switching cycle.

[0064] If there is no DC component at this time, I1+I2=0; if I1+I2≠0, it means that there is a DC component, and long-term accumulation will cause the transformer core to saturate.

[0065] In this embodiment, the current sampling is performed at the midpoint because the midpoint position avoids the noise of the switch tube turning on and off. If the start or end time of the conduction is taken, switching noise may be introduced and the collected current signal will be disturbed. The above current sampling value I1 and current sampling value I2 are both signed, where I1 is a positive value and I2 is a negative value.

[0066] As a preferred technical solution, constructing the compensation amount according to the current sampling value I1 and the current sampling value I2 includes: summing the current sampling value I1 and the current sampling value I2 and multiplying the sum by the compensation coefficient K to obtain the compensation amount K*(I1+I2).

[0067] In this embodiment, the compensation amount is designed to be K*(I1+I2), and the parameter K value is easy to adjust.

[0068] Furthermore, the absolute value of the compensation amount K*(I1+I2) is proportional to the degree of imbalance. When the degree of imbalance is greater, the greater the |I1+I2|, the greater the absolute value of the compensation amount. In this embodiment, the degree of imbalance refers to the difference between the positive half-cycle loop impedance and the negative half-cycle loop impedance, which is reflected in |I1| and |I2|. When balanced, |I1| and |I2| are equal and have opposite signs. The greater the difference between |I1| and |I2|, the greater the degree of imbalance. The greater the degree of imbalance, the greater the compensation amount (that is, the greater the absolute value of the compensation amount), so that the loop compensation will be adjusted faster and stability will be reached faster.

[0069] Furthermore, the design of the K value is related to the actual current and the sampling coefficient. Generally, the compensation amount K*(I1+I2) is 0.1 times the output (the output here refers to the output of the PI regulator, that is, the output of the PI controller in the output voltage single voltage loop). In this embodiment, the actual current mentioned above refers to the load current, because the larger the load current, the larger |I1| and |I2|; the sampling coefficient mentioned above refers to the amplification factor of the current sampling circuit. The current sampling circuit and its amplification factor belong to the prior art and will not be described here.

[0070] As a preferred technical solution, introducing the compensation amount into a single voltage loop to control PWM output includes: introducing the compensation amount into a PI controller in the single voltage loop.

[0071] Further, after the compensation amount is introduced into the PI controller in the single voltage loop, it includes:

[0072] In the positive half cycle when the switch tube Q1 and the switch tube Q4 are turned on, the output of the PI controller is subtracted from the compensation amount, and then sent to the PWM generator after being limited by the limiter to generate a PWM wave with a duty cycle D1 to control the switch tube Q1 and the switch tube Q4;

[0073] In the negative half cycle when the switch tubes Q2 and Q3 are turned on, the output of the PI controller is summed with the compensation amount, and then sent to the PWM generator after being limited by the limiter to generate a PWM wave with a duty cycle D2 to control the switch tubes Q2 and Q3.

[0074] In this embodiment, the control method of the output voltage single voltage loop introducing the compensation amount is as follows: Figure 6 As shown, compared with the traditional single voltage loop control method (such as Figure 4 Compared with the above, there is one more level after the PI controller output (i.e., the compensation amount constructed above is introduced), as follows:

[0075] In the positive half cycle (the positive half cycle here refers to the time when Q1 and Q4 are turned on), the difference between the PI controller output and K*(I1+I2) is made, and then it is sent to the PWM generator after being limited by the limiter to generate a duty cycle D1 to control Q1 and Q4 (that is, a PWM wave with a duty cycle D1 is generated to control the on and off of Q1 and Q4). The PWM wave with a duty cycle D1 in this embodiment is Figure 2 , Figure 3 Vg1 in

[0076] In the negative half cycle (the negative half cycle here refers to the time when Q2 and Q3 are turned on), the output of the PI controller is summed with K*(I1+I2), and then sent to the PWM generator after being limited by the limiter to generate a duty cycle D2 to control Q2 and Q3 (that is, a PWM wave with a duty cycle D2 is generated to control the on and off of Q2 and Q3). The PWM wave with a duty cycle D2 in this embodiment is Figure 2 , Figure 3 Vg2 in.

[0077] In the output voltage single voltage loop of the present embodiment, the difference is taken for the positive half cycle and the sum is taken for the negative half cycle because the current I1 sampled in the positive half cycle is defined to be positive and the current I2 sampled in the negative half cycle is defined to be negative, which is related to the specific current sensor and circuit design. Figure 6 Vout refers to the output voltage of the DC-DC, and Vref refers to the given voltage. The difference between Vout and Vref is input into the PI controller, which belongs to the prior art and will not be described in detail here. Figure 6 Two PWM generators are drawn in the block diagram. This method is used to distinguish that I1 and I2 are not at the same time. In fact, what is implemented in the software program is only one PWM generator.

[0078] For the above-mentioned output voltage single voltage loop control with compensation introduced:

[0079] When the device parameters and the positive and negative half-cycle loop impedances are completely consistent, I1+I2=0, D1=D2;

[0080] When the positive half-cycle loop impedance increases, |I1|<|I2|, I1+12<0, after loop compensation, D1>D2, that is, the positive half-cycle conduction time becomes longer, and the negative half-cycle conduction time becomes shorter, |I1| increases, and |I2| decreases. After several switching cycles, the positive and negative currents are balanced again to ensure magnetic balance;

[0081] When the negative half-cycle loop impedance becomes larger, |I1|>|I2|, I1+I2>0. After loop compensation, D1<D2, that is, the positive half-cycle conduction time becomes shorter, and the negative half-cycle conduction time becomes longer, |I1| decreases, and |I2| increases. After several switching cycles, the positive and negative currents are balanced again to ensure magnetic balance.

[0082] When the positive and negative currents reach balance again, I1+I2=0, D1=D2.

[0083] In this embodiment, the positive half-cycle loop refers to the loop from the positive pole of the input power supply through Q1, the primary coil of the transformer, the current sensor, Q4, and then back to the negative pole of the power supply; the negative half-cycle loop refers to the loop from the positive pole of the input power supply through Q2, the current sensor, the primary coil of the transformer, Q3, and then back to the negative pole of the power supply; the change in loop impedance may be due to factors such as line aging caused by long-term operation and inconsistent conduction voltage drop of MOSFET tubes. The above loop compensation is to introduce the compensation amount K*(I1+I2) for compensation.

[0084] Generally speaking, inconsistent device parameters will also lead to inconsistent loop impedance. The essence of magnetic flux imbalance is current imbalance. Inconsistent device parameters and inconsistent loop impedance will both cause current imbalance, which can be suppressed by the embodiments of the present invention.

[0085] In summary, the embodiment of the present invention adopts a single voltage loop + primary positive and negative current difference compensation control, which is simple to control and easy to adjust parameters. It can effectively suppress the magnetic flux imbalance problem caused by inconsistent device parameters or loop impedance; while solving the magnetic flux imbalance problem, it will not increase the power supply cost and volume, and can better meet the actual application needs.

[0086] The following is a simulation comparison of the flux imbalance caused by inconsistent loop impedance.

[0087] The switching frequency set in this simulation case is 80kHz, that is, the switching period is 12.5us. This embodiment uses the case where a 1Ω resistor is inserted in series in the negative half-cycle loop (here, the 1Ω resistor is inserted in series in the negative half-cycle loop to simulate the increase in the impedance of the negative half-cycle loop) as an example for simulation:

[0088] The simulation results of traditional control methods are as follows Figure 7 As shown, the duty cycle of the positive and negative half cycles is consistent and is 0.3. At this time, due to the inconsistency caused by the loop impedance, the current in the positive half cycle is large, the current in the negative half cycle is small, and the magnetic flux is unbalanced.

[0089] After introducing the compensation amount K*(I1+I2) for compensation, the simulation results are as follows Figure 8 As shown in the figure, the duty cycle of the positive half cycle is 0.298 (the corresponding conduction time is 2.98us), and the duty cycle of the negative half cycle is 0.305 (the corresponding conduction time is 3.05us). The transformer current is positive and negative symmetrical, achieving flux balance.

[0090] The on-time of the switch is related to the switching frequency. For example, if the switching frequency is 100kHz, the switching period is 10us. For example, the duty cycle of the positive half cycle is 0.298, and the corresponding on-time is 2.98us, as shown above.

[0091] The second embodiment of the present invention further provides a magnetic balance control device for a full-bridge DC-DC converter, wherein the full-bridge DC-DC converter includes a transformer and a full-bridge inverter module arranged on the primary side of the transformer, wherein the full-bridge inverter module includes four switch tubes Q1, Q2, Q3, and Q4. Fig. 9 As shown, the device comprises:

[0092] The current sampling module is used to trigger the sampling of the primary current of the transformer to obtain the current sampling value I1 at the midpoint of the conduction time when the switch tubes Q1 and Q4 are turned on; and is also used to trigger the sampling of the primary current of the transformer to obtain the current sampling value I2 at the midpoint of the conduction time when the switch tubes Q2 and Q3 are turned on;

[0093] A compensation amount building module, used for building a compensation amount according to the current sampling value I1 and the current sampling value I2;

[0094] The magnetic balance control module is used to introduce the compensation amount into the single voltage loop to control the PWM output.

[0095] Regarding the device in the above embodiment, the specific manner in which each module performs operations has been described in detail in the embodiment of the method, and will not be elaborated here.

[0096] Based on the same inventive concept, the present invention also provides a computer-readable storage medium storing one or more programs, which, when executed, can implement the aforementioned magnetic balance control method of the full-bridge DC-DC converter.

[0097] Based on the same inventive concept, the present invention also provides an electronic device, such as Fig.10 As shown, it includes a processor, a communication interface, a computer-readable storage medium as described above, and a communication bus; wherein the processor, the communication interface, and the computer-readable storage medium communicate with each other via the communication bus; and the processor is used to execute the program stored in the computer-readable storage medium.

[0098] In the several embodiments provided in the present application, it should be understood that the disclosed devices and methods can be implemented in other ways. For example, the device embodiments described above are only schematic. For example, the division of the modules is only a logical function division. There may be other division methods in actual implementation, such as multiple modules or components can be combined or integrated into another system, or some features can be ignored or not executed. Another point is that the mutual coupling or direct coupling or communication connection shown or discussed can be through some interfaces, indirect coupling or communication connection of devices or modules, which can be electrical, mechanical or other forms.

[0099] The modules described as separate components may or may not be physically separated, and the components displayed as modules may or may not be physical modules, that is, they may be located in one place, or they may be distributed on multiple network modules. Some or all of the modules may be selected according to actual needs to achieve the purpose of the scheme of this embodiment. In addition, the functional modules in the various embodiments of the present invention may be integrated into a processing module, or each module may exist physically separately, or two or more modules may be integrated into one module. The above-mentioned integrated modules may be implemented in the form of hardware or in the form of software functional modules.

[0100] If the integrated module is implemented in the form of a software function module and sold or used as an independent product, it can be stored in a computer-readable storage medium. Based on this understanding, the technical solution of the present invention is essentially or the part that contributes to the prior art or all or part of the technical solution can be embodied in the form of a software product. The computer software product is stored in a storage medium, including several instructions for a computer device (which can be a personal computer, a server, or a network device, etc.) to perform all or part of the steps of the method described in each embodiment of the present invention. The aforementioned storage medium includes: U disk, mobile hard disk, read-only memory (ROM, Read-Only Memory), random access memory (RAM, Random Access Memory), disk or optical disk and other media that can store program codes.

[0101] It should be noted that, for the convenience of description, the aforementioned method embodiments are all described as a series of action combinations, but those skilled in the art should be aware that the present invention is not limited by the described action sequence, because according to the present invention, certain steps can be performed in other sequences or simultaneously. Secondly, those skilled in the art should also be aware that the embodiments described in the specification are all preferred embodiments, and the actions and modules involved are not necessarily required by the present invention.

[0102] In the above embodiments, the description of each embodiment has its own emphasis. For parts that are not described in detail in a certain embodiment, reference can be made to the relevant descriptions of other embodiments.

[0103] Parts not involved in the above embodiments are the same as the prior art or can be implemented by using the prior art, and will not be further described here.

[0104] Although the present invention has been described in detail with reference to the aforementioned embodiments, those skilled in the art should understand that they can still modify the technical solutions described in the aforementioned embodiments, or make equivalent substitutions for some of the technical features therein; and these modifications or substitutions do not make the essence of the corresponding technical solutions deviate from the spirit and scope of the technical solutions of the embodiments of the present invention.

Claims

1. A magnetic balance control method for a full-bridge DC-DC converter, wherein the full-bridge DC-DC converter comprises a transformer and a full-bridge inverter module arranged on the primary side of the transformer, wherein the full-bridge inverter module comprises four switch tubes Q1, Q2, Q3 and Q4, wherein: The method comprises: At the midpoint of the conduction time of the switch tube Q1 and the switch tube Q4, the primary current of the transformer is sampled to obtain a current sampling value I1; At the midpoint of the conduction time of the switch tube Q2 and the switch tube Q3, the primary current of the transformer is sampled to obtain a current sampling value I2; Constructing a compensation amount according to the current sampling value I1 and the current sampling value I2; The compensation amount is introduced into the single voltage loop to control the PWM output.

2. The magnetic balance control method of a full-bridge DC-DC converter according to claim 1, characterized in that: The midpoint time of the on-time is determined by counting the counter.

3. The magnetic balance control method of a full-bridge DC-DC converter according to claim 1, characterized in that: The method of constructing a compensation amount according to the current sampling value I1 and the current sampling value I2 includes: The current sampling value I1 and the current sampling value I2 are summed and then multiplied by the compensation coefficient K to obtain the compensation amount K*(I1+I2).

4. The magnetic balance control method of a full-bridge DC-DC converter according to claim 3, characterized in that: The absolute value of the compensation amount K*(I1+I2) is proportional to the degree of imbalance.

5. The magnetic balance control method of a full-bridge DC-DC converter according to claim 3, characterized in that: The compensation amount K*(I1+I2) is 0.1 times the output amount of the PI controller in the output voltage single voltage loop.

6. A magnetic balance control method for a full-bridge DC-DC converter according to any one of claims 1 to 5, characterized in that: The step of introducing the compensation amount into a single voltage loop to control PWM output comprises: The compensation amount is introduced after the PI controller in the single voltage loop.

7. The magnetic balance control method of a full-bridge DC-DC converter according to claim 6, characterized in that: After introducing the compensation amount into the PI controller in the single voltage loop, the method comprises: In the positive half cycle when the switch tube Q1 and the switch tube Q4 are turned on, the output of the PI controller is subtracted from the compensation amount, and then sent to the PWM generator after being limited by the limiter to generate a PWM wave with a duty cycle D1 to control the switch tube Q1 and the switch tube Q4; In the negative half cycle when the switch tubes Q2 and Q3 are turned on, the output of the PI controller is summed with the compensation amount, and then sent to the PWM generator after being limited by the limiter to generate a PWM wave with a duty cycle D2 to control the switch tubes Q2 and Q3.

8. A magnetic balance control device for a full-bridge DC-DC converter, the full-bridge DC-DC converter comprising a transformer and a full-bridge inverter module arranged on the primary side of the transformer, the full-bridge inverter module comprising four switch tubes Q1, Q2, Q3, and Q4, characterized in that: The device comprises: The current sampling module is used to trigger the sampling of the primary current of the transformer to obtain the current sampling value I1 at the midpoint of the conduction time when the switch tubes Q1 and Q4 are turned on; and is also used to trigger the sampling of the primary current of the transformer to obtain the current sampling value I2 at the midpoint of the conduction time when the switch tubes Q2 and Q3 are turned on; A compensation amount building module, used for building a compensation amount according to the current sampling value I1 and the current sampling value I2; The magnetic balance control module is used to introduce the compensation amount into the single voltage loop to control the PWM output.

9. A computer-readable storage medium storing one or more programs, characterized in that: When the one or more programs are executed, the magnetic balance control method of the full-bridge DC-DC converter according to any one of claims 1 to 7 can be implemented.

10. An electronic device comprising a processor, a communication interface, the computer-readable storage medium of claim 9, and a communication bus; wherein: The processor, the communication interface, and the computer-readable storage medium communicate with each other via a communication bus; It is characterized in that The processor is configured to execute a program stored in a computer-readable storage medium.

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