Radar device with self-interference-based compensation

Abstract

There is described a radar apparatus comprising: i) a transmitter configured to transmit an RF signal; ii) a first receiver configured to receive a first reflection of the RF signal; iii) a second receiver configured to receive a second reflection of the RF signal; and iv) control means configured to: a) determine a first target phase ([phi] 1) and a first self-interference phase ([phi] 1SI) with respect to a first reflection of the RF signal, and compensate a first change in the first target phase ([phi] 1) based on the first self-interference phase ([phi] 1SI) to obtain a compensated first target phase, b) determining a second target phase ([phi] 2) and a second self-interference phase ([phi] 2SI) with respect to the second reflection of the RF signal, and compensating a second change in the second target phase ([phi] 2) based on the second self-interference phase ([phi] 2SI) to obtain a compensated second target phase. Thus, the first change is different from the second change.

Description

Technical Field

[0001] The present disclosure relates to a radar device having a transmitter and at least two receivers, and in particular to determining a phase difference of arrival (PDoA) and / or an angle of arrival (AoA). The radar device is configured to compensate for receiver variations (e.g., time delays and / or phase variations) based on self-interference effects. The present disclosure further relates to a method of compensating for variations in a radar system, and a specific use of a self-interference peak in a channel impulse response (CIR) for compensating for such variations.

[0002] Therefore, the present disclosure may relate to the technical field of radar devices, in particular, ultra-wideband (UWB) applications. Background Art

[0003] Radar operation applies radio frequency (RF) signals / waves to determine properties of a target (object of interest or person or vehicle), such as distance, angle or speed. Thus, a transmitter transmits an RF signal to a target, and one or more receivers obtain reflections (echoes) of the RF signal that have been reflected by the target. Ultra-wideband (UWB) wireless technology, which enables safe and accurate ranging to an accuracy of less than 5 cm, can also be used as a radar device for receiving reflected signals from a target.

[0004] For example, a UWB device can be used as a pulse radar, where a short duration pulse is sent and the reflection of the short duration pulse is received. This reflection is captured as a change in the channel impulse response (CIR) measured by the receiver. In the case where the corresponding antennas of multiple receivers are separated by, for example, half a wavelength, the AoA can be estimated by calculating the PDoA between the CIR values ​​of the two receivers at the tap point index where the reflected signal is detected.

[0005] In a conventional ultra-wideband radar system 200, as Figure 4 As shown in , a transmitter 210 radiates a sequence of short pulses into the environment, which are reflected from a nearby target 240. A first receiver 220 collects the target reflections via a first antenna 221 and determines a complex-valued channel impulse response (CIR), where the target is visible as a peak (with an associated phase Φ1) at a specific CIR tap point (CIR RX1). In order to determine the direction (angle of arrival, AoA) at which the target 240 is located, a second receiver 230 receives the reflections via a physically separated second antenna 231, where the target 240 is again visible as a peak (associated with a phase Φ2) at substantially the same CIR tap point (CIR RX2). However, due to the path differences, the time it takes for the reflections to reach each antenna 221, 231 will include not only a common delay τ, but also a delay difference Δτ. This delay difference will then cause a difference in the phases Φ1 and Φ2 associated with the target peak. By calculating the arrival phase difference:

[0006] PDoA=Φ2-Φ1,

[0007] The angle of arrival can be obtained by the following formula

[0008] AoA=arcsin((λ / L)*(PDoA / 2π)),

[0009] where L is the physical antenna spacing and λ is the wavelength of the carrier signal.

[0010] However, it can be seen that a disadvantage of conventional radar systems is that each receiver 220, 230 has its own (group) delay, as indicated by Figure 4 , as indicated by τ1 and τ2 in . This delay will vary with technology (ie, different receivers will have different delays) and with receiver gain settings. In other words, each receiver 220, 230 may have device-specific variations, characteristics for that particular receiver.

[0011] If not compensated for, the variation (e.g., additional delay difference) may cause PDoA distortion, leading to erroneous AoA results. Figure 5 An example calculation is given in . For example, if a UWB pulse is transmitted at a carrier frequency f (eg, 8 GHz), where ω=2πf, then when the receivers 220, 230 are subject to different delays, the estimated PDoA is given by:

[0012] PDoA=ωΔτ+ω(τ2-τ1).

[0013] like Figure 5 As shown in , PDoA then includes a first true term PDoAtrue and a second error term PDoAerr caused by the receiver delay difference (based on receiver specific variations). This delay difference may cause PDoA distortion, resulting in erroneous AoA results. The delay difference may not only be caused by process variations, but also by the fact that different RX gain settings may result in different delays.

[0014] Figure 6 Plotting the simulated delay (y-axis) of a typical UWB RF analog front end versus gain setting (x-axis) shows that the delay, while independent of temperature, can vary with gain by up to hundreds of picoseconds, which is hundreds of degrees at 8 GHz. In practice, the two receivers can operate at different gains when, for example, a transmitter is closer to one receiver than the other (self-interference).

[0015] From the above PDoA equation, it can be seen that if τ2=τ1 (in other words, both receivers show the same variation), then the estimated PDoA will correspond to the desired PDoA. Therefore, conventional methods for compensating for receiver-specific variations attempt to make the two variations (e.g., delays) equal. A first approach would be to calibrate the receiver-specific delays. Another approach would be gain equalization, where both receivers are forced to have the same gain setting, but this may result in signal saturation or degradation of radar performance (e.g., noise figure or signal-to-noise ratio), and may not account for the general process delay spread between receivers.

[0016] exist Figure 7 The known principles of a UWB pulse radar system are shown in the example of Figure 8 The coupling between the co-located transmitter 110 and the receiver 120 is shown in FIG. The received channel impulse response is calculated by coherent integration of the transmitted pulse. The CIR can be modeled as:

[0017] h(t,τ)=h(s)(τ)+ρej2πf_dtp(τ-τtgt)

[0018] The first term h(s)(τ) is the coupling path assumed to be time-invariant (static), and the second term is the target reflection that varies depending on the Doppler frequency f_d of the target at distance d=(τtgtc) / 2. The reflected signal amplitude is ρ, and the transmitted pulse is p(τ). The coupling peak in the CIR can be called the self-interference of the receiver.

[0019] When the coupling path (self-interference) and the reflected signal (from the target) pass through, for example, an RF filter, the RF signal (pulse) can be Fig. 9 . At a given delay τ, this results in a superposition (overlap) of two signals, one from the coupling path and one from the target reflection. In certain cases, separation may not be possible (see further discussion below). The problem may be that at any given tap point τ of the CIR, the CIR is the sum of the contribution from the coupling path and the contribution from the reflected signal, as Fig.10 This figure shows the CIR at a tap point τ in the complex plane. The static part of the CIR (which is the coupling path) is a complex number and can be represented by a fixed phasor or arrow AB from the origin to this number. The "moving part" of the CIR (which is the target reflection) can be represented by a moving phasor BC. This phasor moves in the complex plane relative to B. If the target moves sinusoidally, the target phasor will move along a circular trajectory, as shown in Fig.10 As seen in.

[0020] Fig.11 The known principle of arrival angle determination is shown. For a UWB radar system with multiple receivers 120, 130, Fig.11 shows the reflected signals arriving at different receivers 120, 130, and Fig.12 It is shown how the arrival time differences between the multiple receivers 120, 130 produce phase differences between the signals received at the different receivers.However, as discussed above, accurate determination of AoA is only possible with compensation device (particularly receiver) specific variations. Summary of the invention

[0021] It may be desirable to compensate for device-specific variations in radar systems, in particular delay or phase variations, in an efficient and reliable manner.

[0022] A radar device, a method and a method of use are provided.

[0023] According to one aspect of the present disclosure, a radar device (e.g., a chip of a mobile phone with UWB function) is described, comprising:

[0024] i) a transmitter configured to transmit an RF signal (eg, a UWB pulse);

[0025] ii) a first receiver configured to receive a first reflection (echo) of the RF signal;

[0026] iii) a second receiver configured to receive a second reflection of the RF signal (particularly having a different phase than the first reflection); and

[0027] iv) a control device (e.g., a processor, an integrated circuit, etc.), the control device being configured to:

[0028] a) determining a first target phase relative to a first reflection of the RF signal (e.g., based on a first target peak in the first CIR) and a first self-interference phase (e.g., based on a self-interference peak in the first CIR), and

[0029] compensating a first (device, receiver) change (e.g., time delay and / or phase change) of a first target phase based on the first self-interference phase (e.g., by subtracting the self-interference phase / peak from the target phase / peak) to obtain a compensated first target phase; and / or

[0030] b) determining a second target phase relative to a second reflection of the RF signal (e.g., based on a second target peak in a second CIR) and a second self-interference phase (e.g., based on a second self-interference peak from the second CIR), and compensating for a second (device, receiver) change in the second target phase based on the second self-interference phase to obtain a compensated second target phase.

[0031] Thereby, the first change specifically differs from the second change (such that the compensated first target phase and the compensated second target phase have (substantially) no (receiver and / or transmitter) change and are directly comparable, specifically without calibration).

[0032] According to another aspect of the present disclosure, a method for compensating for receiver variations in a radar system is described, the method comprising:

[0033] i) sending RF signals;

[0034] ii) receiving a first reflection of the RF signal;

[0035] iii) receiving a second reflection of the RF signal;

[0036] iv) determining a first target phase and a first self-interference phase relative to a first reflection of the RF signal;

[0037] v) compensating a first change of the first target phase based on the first self-interference phase to obtain a compensated first target phase;

[0038] vi) determining a second target phase and a second self-interference phase relative to a second reflection of the RF signal; and

[0039] vii) compensating a second change of the second target phase based on the second self-interference phase to obtain a compensated second target phase.

[0040] Specifically, the first change is different from the second change.

[0041] According to another aspect of the disclosure, a method of using a self-interference peak of a channel impulse response (CIR) to compensate for receiver variations (eg, at least one of time delay and phase variation) relative to a target peak is described.

[0042] In the context of this document, the term "target" may specifically refer to an entity (e.g., an object, a person, an animal, etc.) to be investigated by radar specifically with respect to spatial characteristics such as position, velocity, etc. Thus, a "target peak" may be a detected response signal caused by reflection of a transmitted RF signal from a target. For example, a CIR may be calculated based on reflections of an RF signal, and a target peak may be one or more peaks in the CIR resulting from reflections at a target. Based on the target peak, a target phase (the phase of the received reflected RF signal) may be determined.

[0043] In the context of this document, the term "self-interference" may refer to signals that are not caused by the target but are caused by signal interference within the radar device. An example may be interference between a transmitter and a receiver of the radar device. Such self-interference may also be referred to as spillover, cross-coupling, crosstalk, etc. Figure 8 and 9 A specific example of self-interference is described. Within the CIR, one or more self-interference peaks may be visible. Based on the one or more self-interference peaks, a self-interference phase may be determined.

[0044] According to an exemplary embodiment, the present disclosure may be based on the idea that when a self-interference phase and a target phase are determined for each receiver (each receiver including its own specific variation), receiver variations (specifically delay or phase variations) in a radar system may be compensated in an efficient and reliable manner, and a compensated target phase may be obtained from the target phase by taking into account the self-interference phase (e.g., by subtraction).

[0045] In conventional methods (see above), attempts are made to compensate for receiver variations by using (complex) calibration methods. Self-interference within radar devices is generally undesirable and is attempted to be avoided. The present disclosure now teaches a completely different approach (contrary to conventional approaches): self-interference is required in order to compensate for receiver variations. The inventors have found that the self-interference phase of each receiver shares the same receiver variation as the target phase. Therefore, each target phase variation can be compensated by the corresponding self-interference phase. In this way, accurate position measurements based on the compensated target phases of two or more receivers can be achieved.

[0046] In this way, calibration-free compensation for receiver variations can be provided. The compensated target phase can then be directly compared to determine position characteristics such as PDoA and (PDoA-based) AoA. Since no calibration is required, system complexity can be reduced, thereby achieving, for example, lower memory space and no additional algorithms. The described method can be directly integrated into existing radar devices, in particular where a loopback mode and / or switching of the receiver may not be required.

[0047] The aspects as defined above and further aspects of the disclosure are apparent from the examples of embodiment described hereinafter and are explained with reference to the examples of embodiment.The disclosure will be described in more detail hereinafter with reference to examples of embodiment but to which the disclosure is not limited.

[0048] Exemplary Embodiments

[0049] According to an embodiment, the control device is additionally configured to:

[0050] The (position / spatial) characteristics of the target (e.g., position, velocity, acceleration, movement in space) are determined based on the compensated first target phase and the compensated second target phase. Since the first target phase and the second (or more) target phases are compensated, the accurate spatial position (within the radar system) can be determined.

[0051] According to an embodiment, determining the target characteristic includes determining PDoA. Figure 1 and 2 Describe the procedure. Based on the PDoA, the AoA can be determined, see e.g. Fig.11 and 12 AoA may be an established parameter used to determine the position of a target in space or another characteristic. The present disclosure may, for example, allow tracking of an object / person / vehicle target in two dimensions by estimating the angle in addition to the target's distance. Applications may additionally include several radar use cases such as breathing detection, person tracking, camera autofocus, etc.

[0052] According to an embodiment, the first change and / or the second change comprises at least one of the following: a time delay, a phase change. The change may be a device-specific change of the transmitter and / or the receiver and may involve a time delay and / or a phase change. Since the self-interference undergoes the same changes as the target, efficient compensation may be possible. According to an embodiment, the first change is specific to the first receiver, and / or the second change is specific to the second receiver, and / or the other change is specific to the transmitter.

[0053] According to an embodiment, the control device is additionally configured to compensate for the fixed self-interference delay and / or phase. Thus, the accuracy can be improved, see e.g. Figure 1 The formula in:

[0054] PDoA=(Φ2-Φ1)-(Φ2SI-Φ1SI)+ωτSI

[0055] The additional delay τ SI (or phase ω τ SI ) may be determined and compensated, for example, as part of antenna measurement. This SI delay is thus a fixed value that may not depend on device (chip) conditions (eg process scaling).

[0056] According to an embodiment, the radar device is an ultra-wideband UWB device.Thereby, established and economically important standards can be directly applied.

[0057] In the context of this document, the term "ultra-wideband communication" (or ultra-wideband) may refer to a radio technology that can use extremely low energy levels to communicate over a large portion of the radio spectrum for short-range, high-bandwidth communications. UWB may refer to a technology for transmitting information over a large bandwidth (>500MHz). UWB may be defined as an antenna transmission whose transmission signal bandwidth exceeds the smaller of 500MHz or 20% of the arithmetic center frequency. UWB transmission can transmit information by generating radio energy at specific time intervals and occupying a larger bandwidth, thereby achieving pulse position or time modulation. The range of UWB may be, for example, within a range of tens of meters. UWB (RF) ranging systems may use the time-of-flight principle to determine the distance between a mobile device and a target device (e.g., a car to be driven) and / or an anchor device (tag) on ​​the target device. Typically, the transmitter of a transceiver sends a waveform that is reflected by an object or retransmitted by a second transceiver, which is typically a chirp or pulse. Based on the amount of time it takes for the reflection or retransmission to reach the receiver of the originating transceiver, the distance between objects can be calculated. The range between the receiver and the transmitter thus determined is then used as a control point (to enable access). In the same way, the payload can be transmitted using UWB. In this context, "UWB communication" includes impulse radio ultra-wideband (IR-UWB) (see, for example, standards IEEE802.15.4a and IEEE802.15.4z).

[0058] In another embodiment, another wireless communication scheme may be applied (specifically depending on the standard), such as WiFi, NFC, RFID, etc.

[0059] According to an embodiment, the RF signal comprises at least one pulse, in particular a pulse sequence. According to an embodiment, the target phase is based on one or more target peaks. According to an embodiment, the self-interference phase is based on one or more self-interference peaks.

[0060] According to an embodiment, the control device is additionally configured to:

[0061] i) determining a first channel impulse response (CIR1) based on reflections of the RF signal obtained at the first receiver, and ii) determining a second channel impulse response (CIR2) based on reflections of the RF signal obtained at the second receiver.

[0062] According to an embodiment, the first target phase is associated with at least one first target peak in the first CIR, and the first self-interference phase is associated with at least one first self-interference peak in the first CIR. According to an embodiment, the second target phase is associated with at least one second target peak in the second CIR, and the second self-interference phase is associated with at least one second self-interference peak in the second CIR.

[0063] According to an embodiment, at least one of the first self-interference phase and the second self-interference phase is at least partially caused by coupling between the transmitter and at least one of the first receiver and the second receiver (see e.g. Figure 8 and 9 )produce.

[0064] According to an embodiment, the transmitter is associated with the first receiver or the second receiver. According to an embodiment, the transmitter is spatially closer to one of the first receiver and the second receiver than to the other of the first receiver and the second receiver. According to an embodiment, the transmitter shares a common antenna with the first receiver or the second receiver. According to an embodiment, the transmitter and the first receiver and / or the second receiver are implemented in the same unit or in different units.

[0065] According to an embodiment, a communication system is described, comprising:

[0066] i) a radar apparatus as hereinbefore described; and

[0067] ii) a target from which the RF signal is reflected, wherein the target is a moving target or a fixed target.

[0068] According to an embodiment, determining at least one of the first target phase and the second target phase comprises compensating for signal overlap (superposition, clutter) (e.g., by subtraction) based on a reference response (particularly a reference CIR) at a specific time point (τ0). According to an embodiment, the specific time point (τ0) is selected such that there is a Doppler effect (i.e., a moving target).

[0069] In an example, PDoAξ can be calculated from CIR, and PDoAξ can be mapped back to AoAθ. The CIR of two receivers can be modeled as:

[0070] hrx1(t,τ)=hrx1(s)(τ)+ρej2πf_dtp(τ-τtgt)

[0071] hrx2(t,τ)=hrx2(s)(τ)+ρej2πf_dtp(τ-τtgt)ejξ

[0072] However, in case of strong coupling between co-located transmitter and receiver, PDoA calculation may not be straightforward since different coupling paths (self-interference) for different receiver paths may bias the phase of the reflected signal to a large extent.

[0073] Estimating the line-of-sight AoA of a radar system can be accomplished by calculating the PDoA between multiple receivers. However, for co-located transmitters and receivers, there may be strong coupling from the transmitter antenna to the receiver antenna, which may make PDoA calculation difficult. However, the described method can estimate AoA using a channel impulse response that includes the effects of the coupling path in the presence of strong coupling (including overlap / superposition) from the transmitter to the receiver.

[0074] From the above equations, it can be seen that PDoA cannot be calculated from hrx1(t,τ) and hrx2(t,τ) without removing hrx1(s)(τ) and hrx2(s)(τ). The process of removing (overlapping) coupling paths is called "clutter removal". If hrx1(s)(τ) and hrx2(s)(τ) are not accurately estimated and removed, the residual from clutter removal will result in an incorrect PDoA. A widely used strategy to remove the coupling path contribution is to estimate the average CIR and subtract the average from each CIR:

[0075]

[0076] It can be seen that the residual Will still affect PDoA calculations unless

[0077] The inventors have now found that the accuracy of PDoA estimation can be highly improved in an efficient manner. Fig. 9 ) and cannot directly determine the target peak (target phase).

[0078] Therefore, the CIR at time t0 is regarded as the reference CIR:

[0079] hrx1(ref)(τ)=hrx1(t0,τ)=hrx1(s)(τ)+ρej2πf_dt_0p(τ-τtgt)hrx2(ref)(τ)=hrx2(t0,τ)=hrx2(s)(τ)+ρej2πf_dt_0p(τ-τtgt)ejξ

[0080] And the CIR can be calculated:

[0081]

[0082] It can be seen that And now the PDoA can be calculated accurately.

[0083] Thus, the selection of t0 may affect the accuracy of the PDoA calculation, because for a Doppler frequency f_d close to 0, (ej2πf_dt-ej2πf_dt_0) will also be close to 0. However, in the case of a moving target (e.g., a human target with motion (breathing or walking) with non-zero Doppler), an appropriate window may be selected based on the expected Doppler effect for a given use case. Then, t0 may be selected as the first CIR in the window.

[0084] In the example, the angle can then be found by PDoA is calculated by replacing t=n×RFRI with a discrete time index n running over all CIRs within a given window.

[0085] According to certain embodiments, the present disclosure allows PDoA estimation without having to separately estimate the coupling from the radar transmitter to the co-located radar receiver. According to an example, the present disclosure can be implemented using both phase and delay. According to an example, the self-interference delay can be estimated simply from the (known) antenna spacing. According to an example, at least two phases for each antenna are determined, and the cross coupling (self-interference) is used. According to an example, no calibration is (substantially) required. BRIEF DESCRIPTION OF THE DRAWINGS

[0086] Figure 1 A radar system having a radar device and a target according to an exemplary embodiment of the present disclosure is shown.

[0087] Figure 2 Channel impulse responses with corresponding self-interference are shown according to an exemplary embodiment of the present disclosure.

[0088] Figure 3 A comparison of a measured angle of arrival and a true angle of arrival is shown according to an exemplary embodiment of the present disclosure.

[0089] Figure 4 A conventional radar system is shown.

[0090] Figure 5 A conventional channel impulse response is shown.

[0091] Figure 6 A comparison of gain settings and delays according to a conventional radar system is shown.

[0092] Figure 7 The known principle of a UWB radar system is schematically shown.

[0093] Figure 8 The coupling between a transmitter and a receiver of a radar system according to an exemplary embodiment of the present disclosure is shown.

[0094] Fig. 9A channel impulse response with an overlap of a self-interference peak and a target peak is shown according to an exemplary embodiment of the present disclosure.

[0095] Fig.10 Two phasors in the complex plane are shown, one static phasor from the coupling path and one moving phasor from the target reflection.

[0096] Fig.11 and 12 The known principles of angle of arrival measurement are respectively shown. DETAILED DESCRIPTION

[0097] Before describing the embodiments in more detail with reference to the drawings, some basic considerations upon which the embodiments of the present disclosure are formed will be outlined.

[0098] According to exemplary embodiments, the present disclosure may include at least one of the following aspects:

[0099] i) determining a target peak value in the first channel impulse response and the second channel impulse response;

[0100] ii) determining a self-interference peak in the first channel impulse response and the second channel impulse response;

[0101] iii) Calculate the phase / PDoA associated with the target peak and the self-interference peak;

[0102] iv) Compensating for the delay / phase variation of the target peak phase / PDoA based on the self-interference phase / PDoA.

[0103] Figure 1 A radar system 150 having a radar device 100 (e.g., a UWB radar chip) and a target 140 according to an exemplary embodiment of the present disclosure is shown. The radar device 100 includes a transmitter 110 having a transmitter antenna 111, the transmitter 110 being configured to transmit an RF signal, which is here a UWB pulse. The radar device 100 further includes a first receiver 120, which is configured to receive a first reflection of the RF signal (at the target 140). In this embodiment, the transmitter 110 and the first receiver 120 are associated with each other and share a common antenna 111 (e.g., a common RF pin). Therefore, the transmitter 110 is spatially closer to the first receiver 120 (here co-assigned on the chip) than to the second receiver 130, which is configured to receive a second reflection of the RF signal, which is different (in phase) from the first reflection. The transmitter 110 and the receivers 120, 130 are coupled to a control device (not shown), which determines the PDoA based on the different phases of the first reflection and the second reflection.

[0104] Since the transmitter 110 is strongly coupled to the first receiver 120 but loosely coupled to the second receiver 130, the chains of the two receivers 120, 130 will have different gains due to the receiver variations, resulting in Figure 5 10 (see description above). The figure also shows the delay τ0 through the transmitter 110 (which also depends on the process) and the delay τSI due to the SI path. As can be seen from the equations shown in the figure, by using the self-interference SI as a reference, the delays on the transmitter 110, the first receiver 120 and the second receiver 130 are completely eliminated. The additional delay τSI (or phase ωτSI) can be determined and compensated as part of the antenna measurement. This SI delay is thus a fixed value that does not depend on device (chip) conditions (e.g., process expansion).

[0105] Figure 2 1 and 2 show a first channel impulse response (CIR) of a first receiver RX1 and a second channel impulse response of a second receiver RX2 according to an exemplary embodiment of the present disclosure. Figure 5 ), Figure 2 It is also shown that Figure 1 The present inventors have now recognized that the SI propagating through the first receiver 120 and the second receiver 130 experiences the same variable delay (same variation). Therefore, by, for example, subtracting the SI phase from the corresponding target phase, the delay variation is completely eliminated; in other words, compensating each variable delay of the target phase is done by using the SI phase as a reference.

[0106] Figure 3 A comparison of the measured angle of arrival (y-axis) and the true angle of arrival (x-axis) according to an exemplary embodiment of the present disclosure is shown. It can be seen that a very accurate angle determination is possible. In addition, the overall curve can be observed, as in the case of a calibrated system. It follows that the described (calibration-free compensation method) can produce comparable or better results than conventional methods, but without the need for a costly calibration process.

[0107] Reference numerals

[0108] 100 Radar Device

[0109] 110 Transmitter

[0110] 111 Transmitter antenna, common antenna

[0111] 120 First Receiver

[0112] 121 First receiver antenna

[0113] 130 Second Receiver

[0114] 131 Second receiver antenna

[0115] 140 Target

[0116] 150 Radar System

[0117] 200 Conventional radar system

[0118] 210 Conventional transmitter

[0119] 211 Conventional transmitter antenna

[0120] 220 Conventional first receiver

[0121] 221 Conventional first receiver antenna

[0122] 230 Conventional Second Receiver

[0123] 231 Conventional Second Receiver Antenna

[0124] 240 General objectives.

Claims

1. A radar device (100), characterized in that: include: a transmitter (110) configured to transmit an RF signal; a first receiver (120) configured to receive a first reflection of the RF signal; a second receiver (130) configured to receive a second reflection of the RF signal; as well as A control device, the control device being configured to: a) determining a first target phase (Φ1) and a first self-interference phase (Φ1) relative to the first reflection of the RF signal SI ),and Based on the first self-interference phase (Φ1 SI ) to compensate for a first receiver change of the first target phase (Φ1) to obtain a compensated first target phase; and b) determining a second target phase (Φ2) and a second self-interference phase (Φ2) relative to the second reflection of the RF signal SI ),and Based on the second self-interference phase (Φ2 SI ) and compensating a second receiver variation of the second target phase (Φ2) to obtain a compensated second target phase; Wherein the first receiver variation is different from the second receiver variation.

2. The radar device (100) according to claim 1, characterized in that The control device is further configured to: A target characteristic is determined based on the compensated first target phase and the compensated second target phase.

3. The radar device (100) according to claim 2, characterized in that Determining the target characteristic comprises determining at least one of: Phase difference of arrival PDoA, Angle of arrival AoA.

4. The radar device (100) according to any of the preceding claims, characterized in that At least one of the first receiver variation and the second receiver variation comprises at least one of: Time delay (τ1, τ2), Phase change.

5. The radar device (100) according to any of the preceding claims, characterized in that The control device is further configured to compensate for at least one of: Fixed self-interference delay, Fixed self-interference phase.

6. The radar device (100) according to any of the preceding claims, characterized in that The radar device (100) is an ultra-wideband (UWB) device.

7. The radar device (100) according to any of the preceding claims, characterized in that The control device is further configured to: determining a first channel impulse response (CIR1) based on the first reflection of the RF signal, and determining a second channel impulse response (CIR2) based on the second reflection of the RF signal, wherein the first target phase (Φ1) is associated with at least one first target peak in the first CIR, and wherein the first self-interference phase (Φ1 SI ) is associated with at least one first self-interference peak in the first CIR, and wherein the second target phase (Φ2) is associated with at least one second target peak in the second CIR, and wherein the second self-interference phase (Φ2 SI ) is associated with at least one second self-interference peak in the second CIR.

8. A communication system (150), characterized in that: include: A radar device (100) according to any preceding claim; as well as A target (140) from which the RF signal is reflected, wherein the target (140) is a moving target or a fixed target.

9. A method for compensating for receiver variations in a radar system (150), characterized in that The method comprises: Sending RF signals; receiving a first reflection of the RF signal; receiving a second reflection of the RF signal; determining a first target phase (Φ1) and a first self-interference phase (Φ1SI) relative to the first reflection of the RF signal; compensating a first change of the first target phase (Φ1) based on the first self-interference phase (Φ1SI) to obtain a compensated first target phase; determining a second target phase (Φ2) and a second self-interference phase (Φ2SI) relative to the second reflection of the RF signal; compensating a second change of the second target phase (Φ2) based on the second self-interference phase (Φ2SI) to obtain a compensated second target phase; Wherein the first variation is different from the second receiver variation.

10. A method for using a self-interference peak value of a channel impulse response CIR, characterized in that: The self-interference peak is used to compensate for at least one of a time delay and a phase change relative to a target peak.

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