Superspiral sliding mode control method for three-phase voltage source inverter

Abstract

The invention belongs to the technical field of inverters, and particularly relates to a superspiral sliding mode control method of a three-phase voltage source inverter, which comprises the following steps: detecting IL abc, Vo abc and LC filter output current Io abc; converting the frequency of the inverter into an angle through a formula ph = 2pi * omega; a PBC controller is constructed; constructing an error energy function; a super-spiral sliding mode is introduced into a voltage outer ring, and an STSM is introduced into an inner ring; iL abc, Vo abc and Io abc are transformed from an abc coordinate system to an inductive current reference value # imgabs0 # under an alpha-beta coordinate system through coordinate transformation, and a modulation signal Sinv alpha beta under the alpha-beta coordinate system is obtained; and converting the modulation signal into Sinvabc under an abc coordinate system, and outputting a PWM signal of an inverter switching device in a carrier modulation mode. According to the invention, under the condition that the nonlinear load is connected, the system has relatively strong robustness, and can generate a relatively good compensation effect on the harmonic voltage, so that the total harmonic distortion rate of the output voltage is controlled within a relatively low range, and the problem of voltage buffeting is eliminated, thereby improving the efficiency and reliability of the three-phase inverter.

Description

Technical Field

[0001] The invention belongs to the technical field of inverters, and in particular relates to a super-helical sliding mode control method for a three-phase voltage source inverter. Background Art

[0002] With the advancement of the energy transformation strategy, the proportion of new energy generation represented by photovoltaics in the power system continues to increase. Due to the dispersion of new energy generation locations and line impedance and other factors, the public power grid gradually presents the characteristics of a weak power grid. There are many nonlinear loads in the actual power system, and their access has an impact on the power grid. The nonlinear characteristics of these loads will cause current waveform distortion or three-phase current imbalance. These currents act on the impedance of the transmission line, especially in areas where the power grid is relatively weak and the line impedance is large, which will cause the voltage waveform asymmetry at the point of common coupling (PCC), and introduce additional harmonic voltage components, namely background harmonics. The increase in grid impedance can easily lead to resonance between the inverter and the grid, affecting the stability of the grid-connected inverter system, while background harmonics will cause the inverter grid-connected current to be distorted, resulting in output voltage distortion. In an environment with weak grid strength, current distortion not only weakens the stable operation of the system, but also may cause economic losses, and in severe cases even affect the normal operation of the grid. Therefore, measures must be taken to reduce the adverse effects of resonance and improve the stability of the operation of the grid-connected inverter.

[0003] Generally, when an inverter is connected to a nonlinear load, the output voltage will be distorted. The common voltage-current dual closed-loop controller and active controller have limited compensation effects on harmonic voltages, while the traditional proportional resonant controller can only compensate for specific harmonic voltages (such as the fifth, seventh, eleventh, and thirteenth) but cannot compensate for all harmonic voltages. This results in poor stability in the operation of the current inverter system. Summary of the invention

[0004] The present invention aims to provide a super-helical sliding mode control method for a three-phase voltage source inverter to enhance the controller's ability to suppress harmonics, improve the efficiency and reliability of the three-phase inverter, and reduce costs; adjust the controller parameters, effectively compensate for harmonic voltages, and improve the system's adaptability.

[0005] In order to achieve the above object, the present invention provides a super spiral sliding mode control method for a three-phase voltage source inverter, comprising the following steps:

[0006] Step 1: Detect the inductor current I L abc , detect output voltage V o abc , detect the LC filter output current Io abc ;

[0007] Step 2: Convert the inverter frequency to angle using the formula ph = 2π × ω;

[0008] Step 3: Construct a PBC controller, where the inductor current reference value is designed as:

[0009]

[0010] The debugging signal is designed as:

[0011]

[0012] Step 4: Construct the error energy function, where the error energy function is:

[0013]

[0014] Step 5: Introduce super spiral sliding mode in the outer loop of voltage and STSM in the inner loop, where the super spiral sliding mode is:

[0015]

[0016] Step 6: I L abc 、V o abc and I o abc Through coordinate transformation from the abc coordinate system to the αβ axis coordinate system, the filter inductor current reference value is obtained through the voltage outer loop

[0017] Step 7: Input the current reference value into the current inner loop controller to obtain the modulation signal Sinv in the αβ coordinate system αβ ;

[0018] Step 8: Convert the modulated signal to Sinv in the abc coordinate system abc , and finally output the PWM signal of the inverter switching device through carrier modulation.

[0019] The working principle and beneficial effect of this solution are: by detecting the inductor current I L abc , output voltage V o abc , LC filter output current I o abc , and then transform from the abc coordinate system to the αβ axis coordinate system through coordinate transformation to obtain the I required by the controller Lαβ 、V oαβ ,I oαβ, after passing through the voltage outer loop, the filter inductor current reference value is obtained The current reference value is input into the current inner loop controller to obtain the modulation signal Sinv in the αβ coordinate system. αβ , convert the modulated signal into Sinv in the abc coordinate system abc , and finally output the PWM signal of the inverter switch device through carrier modulation. Through the action of PWM drive signal, the inverter outputs stable three-phase AC power. Enhance the controller's ability to suppress harmonics, improve the efficiency and reliability of the three-phase inverter, and reduce costs; adjust the controller parameters, effectively compensate for harmonic voltage, and improve the system's adaptability.

[0020] Optionally, a filter inductor current sensor is used to detect the inductor current I L abc , use the filter capacitor voltage sensor to detect the output voltage V o abc , use the output current sensor to detect the LC filter output current I o abc .

[0021] Optionally, in step 1, the inductor current I on the inverter side is detected Labc . The number of sensors can be effectively reduced.

[0022] Optionally, in step 3, add a damping term R i , and R i > 0, inject damping r in α and β axes i1 , r i2 :

[0023]

[0024] Among them, r i1 is the current inner loop damping gain, r i2 is the voltage outer loop damping gain.

[0025] To speed up the convergence of the energy function and ensure the global asymptotic stability of the system.

[0026] Optionally, in step 5, the relationship between the damping gain and the output voltage steady-state error is established, where r i1 =30, r i2 = 0.03. Enhanced robustness.

[0027] Optionally, the output voltage in step 3 adopts feed-forward based on LC filter model to increase the dynamic performance of the system.

[0028] Optionally, in step 5, a super-helical sliding mode is added to the voltage outer loop, and the super-helical sliding mode is:

[0029]

[0030] Effectively suppresses harmonic problems and eliminates chattering. BRIEF DESCRIPTION OF THE DRAWINGS

[0031] Figure 1 Schematic diagram of a super-helical sliding mode control method for a three-phase voltage source inverter according to an embodiment of the present invention;

[0032] Figure 2 A simplified topological structure diagram of an inverter system in an embodiment of the present invention;

[0033] Figure 3 is a control block diagram of the damping gain and the state quantity in an embodiment of the present invention;

[0034] Figure 4 This is a block diagram of the current inner loop control in an embodiment of the present invention;

[0035] Figure 5 This is a block diagram of the voltage outer loop control in an embodiment of the present invention;

[0036] Figure 6 The waveform of the output current tracking reference current after the system control method is applied in the embodiment of the present invention;

[0037] Figure 7 The waveform of the output voltage tracking the reference voltage after the system control method is applied in the embodiment of the present invention;

[0038] Figure 8 The output voltage and current waveforms after the system control method is applied in the embodiment of the present invention;

[0039] Fig. 9 It is the harmonic distortion THD after the system control method is applied in the embodiment of the present invention. DETAILED DESCRIPTION

[0040] The following is further described in detail through specific implementation methods:

[0041] Example

[0042] This embodiment is a super spiral sliding mode control method for a three-phase voltage source inverter, and its principle diagram is shown in the attached figure. Figure 1 As shown, the rectangular short dot-dash frame is a harmonic compensation and jitter elimination module, and the rest is a power module. This embodiment uses the following simulation experiments to prove that the designed controller is parameterized, and at the same time, a super-helical sliding mode is added to the voltage outer loop to ensure the stability of the controller. The specific derivation and result proof are combined with the accompanying drawings of the specification as follows:

[0043] The parameters in this simulation verification experiment are as follows:

[0044] Switching frequency fsw =20KHz; filter inductor L1=2mH; r1=0.2Ω; r2=0.1Ω; ω=2π×50Hz; filter capacitor C=22μF; V dc =600V.

[0045] Step 1: Detect the inductor current I through a specific sensor L abc , detect output voltage V o abc , detect the LC filter output current I o abc ;

[0046] Step 2: Convert the inverter frequency to angle using the formula ph = 2π × ω;

[0047] Step 3: Construct a PBC (passive) controller. The specific process is as follows:

[0048] The simplified topology of the inverter system is as follows: Figure 2 As shown, the three-phase voltage of the system is:

[0049]

[0050] The three-phase current is:

[0051]

[0052] From the coordinate transformation formula We can get:

[0053]

[0054] Rewrite equation (1) into EL form: but Where M is a positive definite matrix, J is an antisymmetric matrix, and R is a semi-positive definite matrix, then in is the rate of energy change, f T g is the input energy rate, Q(X) is the dissipation rate, and thus we can get Where X T u is the energy acquisition rate, X T RX is the rate of energy dissipation.

[0055] Let the actual value of the state variable be X and the reference value of the state variable be X r , the state variable error is X e , then X=X e +X r , From this we can get In order to speed up its convergence, the damping term R is added i , then:

[0056]

[0057] R i > 0, inject the same damping r into the α-axis and β-axis i1 , r i2 :

[0058]

[0059] When X e =0, equation (2) can be written as:

[0060]

[0061] Therefore, the inductor current reference value can be designed as:

[0062]

[0063] The debugging signal is designed as:

[0064]

[0065] Step 4: Construct the error energy function. The specific process is as follows:

[0066]

[0067] Among them C e , L 1e 、r 1e is the estimated value of C, L1, r1. Then the error energy function is:

[0068]

[0069] but:

[0070]

[0071] when When , He converges and the system is globally asymptotically stable.

[0072] Substituting (3) into (1) we obtain:

[0073]

[0074] Further we can get:

[0075]

[0076] Similarly:

[0077]

[0078] Further we can get:

[0079]

[0080] Substituting (5) and (6) into (4), we obtain:

[0081]

[0082] If (7) C e , L 1e 、r 1e are equal to C, L1, and r1 respectively, then:

[0083]

[0084] Then the global state is asymptotically stable, and the damping gain term is further ignored: It can be seen that the introduction of the damping term accelerates the decay rate of the error energy function, ignoring the damping gain term. for:

[0085]

[0086] From the above derivation, it can be concluded that it is impossible to guarantee that is negative, and the damping term can still make the energy function change rate when there is an offset in the model parameters This can ensure the global asymptotic stability of the system, but excessive damping can also cause system oscillation;

[0087] Step 5: Establish the relationship between damping gain and output voltage steady-state error. The specific process is as follows:

[0088] The control block diagram of damping gain and state quantity is as follows: Figure 3 As shown, the voltage outer loop damping gain is r i2 , the current inner loop damping gain is r i1 , the two state quantities are i Lαβ , V 0αβ , the feedforward term based on the LC filter model is C e S.L. 1e S+r 1e . With the control block diagram, the transfer function is Then the steady-state error in time domain Further we can get e ss (∞)≠0, so when Amp>0, even if there is no model parameter deviation in the system, the output voltage still has a sinusoidal steady-state error, and the damping r i1 、r i2 The bigger it is, the smaller the Amp is.

[0089] Design simulation parameters:

[0090] Switching frequency f sw =20KHz; filter inductor L1=2mH; r1=0.2Ω; r2=0.1Ω; ω=2π×50Hz; filter capacitor C=22μF; V dc =600V;

[0091] The current inner loop control block diagram is as follows: Figure 4 As shown, r can be obtained by debugging signal design i1 ≈30, the voltage outer loop control block diagram is as follows Figure 5 As shown, r can be obtained by designing the inductor current reference value. i2 =0.27. i1 =30, r i2 =0.27 is the critical stability point of the system, obtained by simulation debugging, taking r i1 =30, r i2 When =0.03, the system is highly robust.

[0092] Step 6: Introduce super spiral sliding mode in the outer loop of voltage and STSM in the inner loop. The specific process is as follows:

[0093] The output voltage has harmonics and is accompanied by jitter problems. Consider introducing sliding mode into the system. There is an inverter model:

[0094]

[0095] Control Model:

[0096]

[0097] set up (V 0α and V 0β The calculation process is the same for the same coordinate axis. The calculation under the α coordinate axis is taken as an example below. Then we have:

[0098]

[0099] Simplified: so Further Depend on Available, Cλ1=10~15, the saturation function sat is used instead of the sign function sign, the simulation waveform also has jitter, and the harmonic problem cannot be solved.

[0100] To solve this problem, a PBC (passive) controller + inner loop sliding mode + outer loop sliding mode is used. make (iLα and i Lβ The calculation process is the same for the same coordinate axis. The calculation under the α coordinate axis is taken as an example below. so:

[0101]

[0102] Further:

[0103]

[0104] so

[0105] This can solve the harmonic problem, but there is still a chattering problem. Consider introducing a super-helical sliding mode in the voltage outer loop:

[0106]

[0107] because It is difficult to simplify. simplify:

[0108]

[0109] The voltage outer loop is:

[0110]

[0111] Depend on but so Further we can get:

[0112]

[0113] The super spiral sliding mode is introduced in the voltage outer loop to eliminate the chattering, and the STSM is introduced in the inner loop. The current inner loop is:

[0114]

[0115] make Depend on We can get:

[0116]

[0117] so Further we can get:

[0118]

[0119] In summary, the controller model introducing the super-helical sliding mode can be written as:

[0120]

[0121] Step 7: I L abc 、V o abc and I o abc Through coordinate transformation from the abc coordinate system to the αβ axis coordinate system, the filter inductor current reference value is obtained through the voltage outer loop

[0122] Step 8: Input the current reference value into the current inner loop controller to obtain the modulation signal Sinv in the αβ coordinate system αβ ;

[0123] Step 9: Convert the modulated signal to Sinv in the abc coordinate system abc ,Finally, the PWM signal of the inverter switching device is output through carrier modulation.

[0124] The simulation output effect of the inverter when connected to nonlinear load and balanced load is as follows Figures 6 to 9 As shown, Figure 6 For the output current to track the waveform of the reference current, Figure 7 For the output voltage to track the waveform of the reference voltage, Figure 8 is the output voltage and current waveform, Fig. 9 is the harmonic distortion rate of the output voltage. The nonlinear load is connected in 0-0.4s, and the balanced load is connected in 0.4s-0.8s. Figure 6 and Figure 7 It can be seen that the tracking error between the output current and the reference current is about 0.8A, which is in a small range, and the error between the output voltage and the reference voltage is about 0.4V; Figure 8 It can be seen that connecting different loads will have different degrees of impact on the output current. Fig. 9 It can be seen that the harmonic distortion rate of the inverter output voltage is 0.41%, indicating that the voltage output quality of the inverter system using this method is relatively high. From the above simulation experiments, it can be concluded that when different loads are connected and the load changes, the method proposed in this paper can better guarantee the quality of the output voltage, has a relatively ideal compensation effect on the harmonic voltage, and eliminates the voltage jitter problem, ensuring the good output effect of the three-phase voltage source inverter.

Claims

1. A super-helical sliding mode control method for a three-phase voltage source inverter, characterized in that: The following steps are involved: Step 1: Detect the inductor current I L abc , detect output voltage V o abc , detect the LC filter output current I o abc ; Step 2: Convert the inverter frequency to angle using the formula ph = 2π × ω; Step 3: Construct a PBC controller, where the inductor current reference value is designed as: The debugging signal is designed as: Step 4: Construct the error energy function, where the error energy function is: Step 5: Introduce super spiral sliding mode in the outer loop of voltage and introduce STSM in the inner loop, where the super spiral sliding mode is: Step 6: I L abc 、V o abc and I o abc Through coordinate transformation from the abc coordinate system to the αβ axis coordinate system, the filter inductor current reference value is obtained through the voltage outer loop Step 7: Input the current reference value into the current inner loop controller to obtain the modulation signal Sinv in the αβ coordinate system αβ ; Step 8: Convert the modulated signal to Sinv in the abc coordinate system abc , and finally output the PWM signal of the inverter switching device through carrier modulation.

2. The super spiral sliding mode control method of a three-phase voltage source inverter according to claim 1, characterized in that: Using filter inductor current sensor to detect inductor current I L abc , use the filter capacitor voltage sensor to detect the output voltage V o abc , use the output current sensor to detect the LC filter output current I o abc .

3. The super spiral sliding mode control method of a three-phase voltage source inverter according to claim 2, characterized in that: In step 1, the inductor current I on the inverter side is detected L abc .

4. The super spiral sliding mode control method of a three-phase voltage source inverter according to claim 1, characterized in that: In step 3, add the damping term R i , and R i > 0, inject damping r in α and β axes i1 , r i2 : Among them, r i1 is the current inner loop damping gain, r i2 is the voltage outer loop damping gain.

5. The super spiral sliding mode control method of a three-phase voltage source inverter according to claim 4, characterized in that: In step 5, the relationship between the damping gain and the output voltage steady-state error is established, where r i1 =30, r i2 =0.

03.

6. The super spiral sliding mode control method of a three-phase voltage source inverter according to claim 1, characterized in that: The output voltage in step 3 adopts feed-forward based on LC filter model.

7. The super spiral sliding mode control method of a three-phase voltage source inverter according to claim 1, characterized in that: In step 5, a super-helical sliding mode is added to the voltage outer loop, and the super-helical sliding mode is:

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