A single-stage bidirectional power conversion topology capable of wide-range soft switching and a control method thereof

Abstract

The application discloses a single-stage bidirectional power conversion topology structure capable of realizing wide-range soft switching and a control method thereof. The bridge-in and bridge-between phase shift control of a double half-bridge circuit and a high-frequency switch matrix provides a basis for realizing soft switching of all power tubes. The soft switching of a wide voltage variation range is realized by real-time adjustment of a switching frequency according to the size of a port voltage. The bridge arm current connected with the primary side of a second transformer is the sum of the primary side current of the second transformer and the inductance current connected with the primary side of the second transformer in parallel. By designing the inductance value of the parallel inductance connected with the primary side of the second transformer, the bridge arm current connected with the primary side of the second transformer can be effectively changed, so that the soft switching realization range of the switching tube of the bridge arm connected with the primary side of the second transformer under the wide power transmission range can be expanded.

Description

TECHNICAL FIELD

[0001] The present application relates to the field of high-frequency isolated single-stage converter technology in power electronic converters, and particularly relates to a single-stage bidirectional power conversion topology capable of realizing wide-range soft switching and a control method thereof. BACKGROUND

[0002] The dual active bridge converter has the advantages of electrical isolation, simple circuit structure, easy modularization, bidirectional power transmission, and soft switching, and has been widely researched and applied in many bidirectional DC / DC conversion and bidirectional AC / DC conversion fields such as battery energy storage, new energy electric vehicles, solid-state transformers, and household energy storage systems.

[0003] The initial application scenario of the dual active bridge converter is bidirectional DC / DC conversion. The single-stage high-frequency isolated bidirectional DC / DC converter based on the dual active bridge (DAB) can realize soft switching of all switch tubes of the converter when the load is full or the transmission power is large. However, when the transmission power of the converter becomes small or the voltages of the two DC ports deviate from the optimal ratio determined by the transformer ratio (i.e., the two port voltages are mismatched), the converter is difficult to realize soft switching of all switch tubes. The loss of soft switching of the switch tubes will increase the switching loss of the converter, reduce the efficiency, and increase the electromagnetic interference (EMI), which seriously reduces the performance and reliability of the converter, and even cannot work normally. Therefore, a large number of researchers have analyzed and researched the circuit structure and control technology of the DAB converter, including the modulation techniques of double phase shift control (DPS), extended phase shift (EPS), and triple phase shift (TPS) based on the basic single phase shift control (SPS), and the combination application method of these modulation techniques, and the use of variable structure power circuit to reduce the voltage deviation to expand the soft switching range. However, whether the modulation strategy is changed or the circuit structure is changed, there are problems such as complex design calculation, complex modulation algorithm, and complex power circuit structure, which are difficult to be applied in practice or difficult to improve the comprehensive performance.

[0004] With the development of electrochemical energy storage technology, in recent years, the technology of bidirectional AC / DC converter has been widely valued and applied. The existing bidirectional AC / DC converter is mostly composed of a non-isolated bidirectional AC / DC converter (PWM (pulse width modulation) converter) and an isolated bidirectional DC / DC converter (such as the aforementioned DAB converter) two-stage structure. Due to the wide range of battery port voltage, the two-stage converter also has the problem of wide voltage range application of the DAB DC converter. Although the output voltage of the AC / DC converter can be adjusted to match the voltage at both ends of the DAB converter, this makes the control of the converter complex. In order to overcome the shortcomings of the two-stage bidirectional AC / DC converter scheme, such as multiple power conversion stages and hard switching of AC / DC conversion, in recent years, many researchers have carried out research on high-frequency isolated single-stage bidirectional AC / DC conversion technology based on DAB converter. The bidirectional AC / DC converter based on DAB reduces the number of power conversion stages, and through the optimization design of circuit parameters and control strategy, it can realize soft switching of power switches in a certain load size and part of the AC voltage period, which has certain technical advantages over the two-stage conversion scheme. However, due to the inherent wider instantaneous value range of the AC side voltage (from 0 to peak value), the performance of the DAB converter in wide voltage range is poor.

[0005] The existing optimization methods of soft switching of single-stage bidirectional AC / DC converter based on DAB are optimized from the modulation strategy, but these methods only optimize the soft switching range of the converter at full load. For example, some scholars have proposed a modulation strategy based on variable switching frequency to realize critical current continuous mode operation, which can realize soft switching of all switching tubes at full load. However, when the transmission power decreases, this modulation strategy will result in high switching frequency of the converter, and most of the voltage period cannot realize soft switching, thus greatly increasing the switching loss and not having practicality.

[0006] Therefore, it is of great theoretical and practical significance to propose a bidirectional power converter structure and modulation technology that can realize wide-range soft switching of power tubes in a wide voltage and load range. SUMMARY

[0007] The present application provides a single-stage bidirectional power conversion topology structure and control method that can realize wide-range soft switching, which can effectively solve the problem of limited soft switching range of switching tubes of the converter based on DAB structure when the voltage is not matched or the converter is under light load.

[0008] Technical solution: The present invention discloses a single-stage bidirectional power conversion topology structure capable of realizing wide-range soft switching, comprising: a dual half-bridge circuit, a first transformer, a second transformer, an inductor, a power transmission inductor, a high-frequency switch matrix circuit, and a filter; the dual half-bridge circuit is connected to the first port of the converter, which is a DC input or output port; the dual half-bridge circuit includes two capacitors C d1 with C d2 And four power switches S1~S4, the first transformer and the second transformer primary are connected in series, the connection point of the two transformer primary sides and C d1 with C d2 The two switches in each arm of the dual half-bridge circuit operate in a complementary manner with a 50% duty cycle. The switching frequency is the same as the switching frequency of the power transistors in the high-frequency switch matrix circuit, converting DC power into a frequency-variable alternating square wave with symmetrical positive and negative amplitudes, which is applied to the primary of the corresponding transformer. The switches in the two arms are controlled by internal bridge phase shifting. The secondary output of the series transformer is a quasi-square wave, the width of which is determined by the internal bridge phase shift control angle θ. The power switches in the high-frequency switch matrix circuit also use 50% duty cycle modulation to convert the industrial frequency AC or DC voltage at the other port into a high-frequency modulated AC voltage with a 50% duty cycle. The switching frequency of the high-frequency switch is determined by the instantaneous value of the voltage at that port. The voltages across the power transfer inductor are the quasi-square wave voltage output by the series transformer and the alternating voltage modulated by the high-frequency switch matrix, respectively. The phases of the two waveforms are determined by the inter-bridge phase shift control angle θ between the dual half-bridge circuit and the high-frequency switch matrix circuit. Decision: The filter is connected between the high-frequency switch matrix and the second port of the converter, and the second port is a DC or AC output or input port.

[0009] Furthermore, the high-frequency switch matrix circuit selects a full-bridge matrix circuit, a half-bridge matrix circuit or other circuits that can convert a DC or low-frequency AC voltage into a high-frequency alternating voltage output.

[0010] Furthermore, the power transmission inductor is an independent inductor or the sum of the secondary equivalent leakage inductances of the first transformer and the second transformer, or is composed of one or more independent inductors and the leakage inductances of the first transformer and the second transformer.

[0011] Furthermore, the first transformer and the second transformer are transformers with independent magnetic cores, or are magnetic integrated transformers composed of a single magnetic circuit combined with a magnetic core.

[0012] Furthermore, the inductor connected in parallel with the primary side of the second transformer may be an independent magnetic core, or may be integrated with the magnetic core of the second transformer into a single magnetic core.

[0013] Further, the filter is an LC filter, a C filter, or other type of low-pass filter through which low-frequency current components pass and high-frequency current components are filtered out.

[0014] Further, the converter AC side is connected to the power grid to realize rectifier, grid-connected inverter, energy storage converter, or other applications; or as an off-grid inverter application to supply power to AC loads.

[0015] Correspondingly, a control method of a single-stage bidirectional power conversion topology capable of realizing wide-range soft switching includes the following steps:

[0016] Step 1: When used for AC / DC conversion, the relationship between the magnitude of the AC side current i ac and the control phase shift angle θ and is derived according to modal analysis;

[0017] Step 2: The bridge-to-bridge control phase shift angle and the bridge-to-bridge control phase shift angle θ are in proportional relationship, that is, where the proportional coefficient k is a constant determined according to the optimized design of the circuit performance;

[0018] Step 3: The AC current and the bridge-to-bridge control phase shift angle θ are in linear relationship to simplify the implementation of the control strategy.

[0019] Further, in Step 1, the relationship between the magnitude of the AC side current i ac and the control phase shift angle θ and is derived according to modal analysis:

[0020]

[0021] In the formula: C = 2 or 4 or other constant, C = 2 when the amplitude of the high-frequency modulated AC voltage output by the high-frequency switching matrix circuit is equal to the amplitude of the AC side voltage, and C = 4 when the amplitude of the high-frequency modulated AC voltage output by the high-frequency switching matrix circuit is half of the amplitude of the AC side voltage; V dc is the voltage of the DC port, n is the turns ratio of the secondary side to the primary side of the first transformer and the second transformer, T s is the switching period, and L lk is the power transmission inductance value including the equivalent leakage inductance to the secondary side of the two transformers.

[0022] Further, in Step 2, the relationship between the magnitude of the AC side current i ac and the control phase shift angle θ becomes:

[0023]

[0024] Further, in Step 3, it is determined that:

[0025]

[0026] Wherein, T smin is the minimum switching period of the switching tube of the converter determined according to the specific power level of the converter, the type of power device selected and the optimization of the comprehensive electrical performance of the system, the switching period T s is determined by θ as follows:

[0027]

[0028] The switching frequency of the high-frequency switching tube varies with the instantaneous value of the AC side voltage, and the switching frequency is higher when the voltage is lower;

[0029] When the AC side voltage varies sinusoidally, i.e.:

[0030] |v ac |=V acpeak |sinωt|

[0031] Wherein, V acpeak is the peak value of the AC side voltage v ac , and when the unit power factor control is realized, the bridge internal phase shift control angle θ is:

[0032] θ=θ peak |sinωt|

[0033] Wherein θ peak is the maximum value of the design-selected double half-bridge bridge internal phase shift control angle θ;

[0034] Then the switching period T s is related to the AC side voltage as follows:

[0035]

[0036] Advantages: Compared with the prior art, the present application has the following remarkable advantages: the bridge internal and inter-bridge phase shift control of the double half-bridge circuit and the high-frequency switching matrix in the present application provides a basis for realizing the soft switching of all power tubes; the real-time adjustment of the switching frequency according to the port voltage realizes the soft switching in a wide voltage variation range; the bridge arm current connected to the primary side of the second transformer is the sum of the primary current of the second transformer and the inductance current connected in parallel with the second transformer, and by designing the inductance value of the inductance connected in parallel with the primary side of the second transformer, the bridge arm current connected to the primary side of the second transformer can be effectively changed, so that the soft switching realization range of the switching tube of the bridge arm connected to the primary side of the second transformer under the wide power transmission range can be expanded. BRIEF DESCRIPTION OF DRAWINGS

[0037] Figure 1 is the schematic diagram of the converter circuit structure of the present application.

[0038] Figure 2 This is a circuit topology diagram of a bidirectional AC / DC converter used in the present invention.

[0039] Figure 3 The present invention is applied Figure 2 The working waveform of a switching cycle in the positive half cycle of the converter AC.

[0040] Figure 4 for Figure 3 The current i flowing into the lagging power tube corresponding to the working cycle son 's analysis diagram.

[0041] Figure 5(a) shows Figure 2 Circuit modal diagram of the topology at the [t0-t1] stage.

[0042] Figure 5(b) shows Figure 2 Circuit modal diagram of the topology at the [t1-t2] stage.

[0043] Figure 5(c) shows Figure 2 Circuit modal diagram of the topology at the [t2-t3] stage.

[0044] Figure 5(d) shows Figure 2 Circuit modal diagram of the topology at the [t3-t4] stage.

[0045] Figure 5(e) shows Figure 2 Circuit modal diagram of the topology at the [t4-t5] stage.

[0046] Figure 5(f) shows Figure 2 Circuit modal diagram of the topology at the [t5-t6] stage.

[0047] Figure 5(g) shows Figure 2 Circuit modal diagram of the topology at stage [t6-t7].

[0048] Figure 5(h) shows Figure 2 Circuit modal diagram of the topology at stage [t7-t8].

[0049] Figure 5(i) shows Figure 2 Circuit modal diagram of the topology at stage [t8-t9].

[0050] Figure 5(j) shows Figure 2 Circuit modal diagram of the topology at the [t9-t10] stage.

[0051] Figure 6(a) shows Figure 2 Experimental waveform of the AC voltage peak of switch tube S2 when the topology is lightly loaded.

[0052] Figure 6(b) shows Figure 2 Experimental waveform of switch S2 at a certain middle moment in the power frequency cycle when the topology is lightly loaded.

[0053] Fig. 6(c) is Figure 2 Fig. 6(c) is

[0054] Fig. 6(d) is Figure 2 Fig. 6(d) is

[0055] Fig. 6(e) is Figure 2 Fig. 6(e) is

[0056] Fig. 6(f) is Figure 2 Fig. 6(f) is

[0057] Fig. 6(g) is Figure 2 Fig. 6(g) is

[0058] Fig. 6(h) is Figure 2 Fig. 6(h) is

[0059] Fig. 6(i) is Figure 7 Fig. 6(i) is

[0060] Figure 1 Fig. 6(i) is DETAILED DESCRIPTION

[0061] As Figure 2 shown in the figure, a single-stage bidirectional power conversion topology capable of realizing wide-range soft switching includes: a power conversion circuit composed of a double half-bridge circuit, a first transformer, a second transformer, an inductor connected in parallel with the second transformer, a power transmission inductor, a high-frequency switching matrix circuit, a filter, etc.; two capacitors C d1 and C d2 and four power switches S1-S4 (including active switches and antiparallel freewheeling diodes) constitute a double half-bridge circuit; the double half-bridge circuit is connected to a first port of the converter, and the first port is a direct-current input or output port; the primary windings of the first transformer and the second transformer are connected in series, and the connection point of the primary windings of the two transformers is connected to C d1 and C d2The first transformer and the second transformer are connected in series, and the power transmission inductor is connected in series with the secondary side of the series transformer; the two switches of each bridge arm of the double half-bridge circuit are operated complementarily with a duty cycle of 50%, and the switching frequency is the same as the switching frequency of the power tube of the high-frequency switching matrix, so as to convert the direct current into a frequency-variable positive-negative amplitude symmetrical alternating square wave, which is applied to the primary side of the corresponding transformer; the switches of the two bridge arms are controlled by phase-shifting, and the output of the secondary side of the series transformer is a quasi-square wave, and the quasi-square wave width is determined by the phase-shifting control angle θ of the bridge; the power switch of the high-frequency switching matrix circuit is also modulated with a duty cycle of 50%, so as to convert the power-frequency alternating voltage or direct current voltage of the other port into a high-frequency modulated alternating voltage with a duty cycle of 50%, and the switching frequency of the high-frequency switch tube is determined by the instantaneous value of the voltage of the port; the voltages at the two ends of the power transmission inductor are respectively the quasi-square wave voltage output by the series transformer and the high-frequency alternating voltage modulated by the high-frequency switching matrix, and the phase of the two waveforms is determined by the phase-shifting control phase angle between the bridge of the double half-bridge circuit and the bridge of the high-frequency switching matrix circuit , and the phase angle determines the effective value of the current of the power transmission inductor, that is, the transmission power between the two ports is determined; the filter is connected between the high-frequency switching matrix and the second port of the converter, and the second port is a direct current or alternating current output or input port.

[0062] As Figure 3 shown, it is a specific implementation circuit of the application when AC / DC conversion is performed, wherein the high-frequency switching matrix circuit is composed of a capacitor C b , switch tubes S5-S8, a capacitor C1 and a capacitor C2. Taking the case that the converter works in the inverter mode as an example, Figure 4 a typical working waveform diagram thereof is shown, the time lag of the driving signal of the switch tube S4 relative to the driving signal of the switch tube S1 is (0.5-θ)T s , and the time lead of the driving signal of the switch tube S4 relative to the driving signal of the switch tube S5 is , wherein θ is twice . The size of the switching period T s is in a functional relationship with the alternating current side voltage, so as to realize the linear relationship between the size of the alternating current side output current i ac and the control phase-shifting angle θ. Under the modulation strategy, all the switch tubes on the alternating current side and the switch tubes S1 and S2 on the direct current side can realize zero-voltage turn-on in the full output power range and the full power frequency period, and when the output power decreases, the switch tubes S4 and S3 of the lagging bridge arm on the direct current side are difficult to realize zero-voltage turn-on. For the switch tubes S3 and S4 on the direct current side, the existence of the inductor L m2 parallelly connected to the transformer T2 is helpful to the zero-voltage turn-on of the switch tubes, as Figure 2 shown. When the load is light, the converter is grid-connected inverter, and the alternating current side voltage v ac is positive, and the mode analysis is as follows, wherein the switch tubes S7 and S8 are in the positive half cycle of the alternating current side voltage v acThe constant on state is achieved.

[0063] Mode 1 [t0, t1]: As shown in Fig. 5(a), at this time, the switch tubes S1, S3, S6, S7 and S8 are turned on. During this period, the voltage v ab is 0, and the voltage v cd at the right end of the inductor is -v ac / 2. The inductor current linearly increases during this phase.

[0064] Mode 2 [t1, t2]: As shown in Fig. 5(b), at this time, the switch tube S3 is turned off, the switch tube S4 has not been turned on, and the switch tubes S1, S6, S7 and S8 are turned on. The leakage inductance current i lk is greater than zero, and the primary current i p of the transformer T2 is greater than zero. However, at this time, due to the existence of the inductor L m2 , the current i son flowing into the bridge arm of the switch tubes S3 and S4 is less than zero, and the current i son discharges the junction capacitor of the switch tube S4 to 0 and charges the junction capacitor of the switch tube S3 to V dc , thereby creating conditions for soft switching of the switch tube S4.

[0065] Mode 3 [t2, t3]: As shown in Fig. 5(c), this mode is still in the dead time of the switch tubes S3 and S4, and the switch tubes S1, S6, S7 and S8 are turned on. At this time, the current i son flows through the body diode of the switch tube S4, so that the switch tube S4 is turned on at zero voltage in the next mode.

[0066] Mode 4 [t3, t4]: As shown in Fig. 5(d), at t3, the switch tube S4 is turned on at zero voltage, and the switch tubes S1, S6, S7 and S8 are turned on. During this phase, the voltage at the left end of the inductor is nV dc , and the voltage v cd at the right end of the inductor is v ac / 2. The inductor current linearly increases during this phase.

[0067] Mode 5 [t4, t5]: As shown in Fig. 5(e), at t4, the switch tube S6 is turned off, and the switch tube S5 has not been turned on during this phase. During this phase, the switch tubes S1, S4, S7 and S8 are turned on. During this phase, the leakage inductance current i lk discharges the junction capacitor of the switch tube S5 to zero, and the junction capacitor of the switch tube S6 is charged to v ac , thereby creating conditions for soft switching of the switch tube S5.

[0068] Mode 6 [t5, t6]: As shown in Fig. 5(f), this mode is still in the dead time of the switch S5, S6, at the moment t5, the junction capacitance voltage of the switch S5 drops to zero, in this stage, the switches S1, S4, S7, S8 are turned on, the inductor current flows through the body diode of the switch S5 to create conditions for the zero voltage turn-on of the switch S5.

[0069] Mode 7 [t6, t7]: As shown in Fig. 5(g), since the body diode of the switch S5 has been turned on in the previous stage, at the moment t6, the switch S5 realizes zero voltage turn-on, in this mode, the switches S1, S4, S5, S7, S8 are turned on, the left end voltage of the inductor is nV dc , the right end voltage of the inductor is v ac / 2, and the inductor current linearly rises.

[0070] Mode 8 [t7, t8]: As shown in Fig. 5(h), at the moment t7, the switch S1 is turned off, and the switch S2 has not been turned on, in this stage, the switches S4, S5, S7, S8 are turned on, in this mode, the current i p charges the junction capacitance of the switch S1 to V dc , and discharges the junction capacitance of the switch S2 to 0.

[0071] Mode 9 [t8, t9]: As shown in Fig. 5(i), this mode is still in the dead time of the switches S1, S2, at the moment t8, the junction capacitance voltage of the switch S2 is 0. In this stage, the switches S4, S5, S7, S8 are turned on, in this stage, the inductor current i p flows through the body diode of the switch S2 to create conditions for the zero voltage turn-on of the switch S2.

[0072] Mode 10 [t9, t10]: As shown in Fig. 5(j), since the body diode of the switch S2 has been turned on in the previous stage, at the moment t9, the switch S2 realizes zero voltage turn-on. At this time, the voltage v ab is 0, the right end voltage of the inductor v cd is -v ac / 2, and the inductor current linearly decreases.

[0073] The latter half of the switching period is symmetrical to the mode principle of the former half of the period, and will not be described again.

[0074] In the implementation circuit, the size of the switching period T s is a function of the AC side voltage to realize the linear relationship between the size of the AC side output current i ac and the control phase shift angle θ. A specific scheme is as follows:

[0075] Step 1: According to the mode analysis, the size of the AC side current i ac is a function of the control phase shift angle θ and The relationship between the step 1 and the step 2 is:

[0076]

[0077] The step 2 is to make the inter-bridge control phase shift angle proportional to the intra-bridge control phase shift angle θ, that is, wherein the proportional coefficient k is a constant determined according to the optimization design of the circuit performance. In this way, the magnitude of the alternating current i ac is changed with the control phase shift angle θ as follows:

[0078]

[0079] The step 3 is to make the alternating current linearly related to the intra-bridge control phase shift angle θ to simplify the implementation of the control strategy, and to determine:

[0080]

[0081] wherein T smin is the minimum switching period (the highest switching frequency) of the switching tube of the converter determined according to the specific power level, the input and output voltage, the selected power device type and the comprehensive electrical performance optimization of the system. In this way, the switching period T s is determined by θ as follows:

[0082]

[0083] In this way, the switching frequency of the high-frequency switching tube is changed with the instantaneous value of the alternating voltage, and the switching frequency is higher when the voltage is lower.

[0084] When the alternating voltage is sinusoidal, that is:

[0085] |v ac | = V acpeak |sinωt|

[0086] wherein V acpeak is the peak value of the alternating voltage v ac . When the unit power factor control is implemented, the intra-bridge control phase shift angle θ is:

[0087] θ = θ peak |sinωt|

[0088] wherein θ peak is the maximum value of the intra-bridge control phase shift angle θ of the double half-bridge selected by design.

[0089] The relationship between the switching period T s and the alternating voltage is:

[0090] ​

[0091] When the converter is actually working, the switching frequency of the high-frequency switch tube can be determined based on the pre-designed parameters and the instantaneous value of the AC side voltage detected in real time based on the above formula, and the turn-on and turn-off moments of each switch tube can be determined based on this and the size of the phase shift angle.

[0092] The aforementioned switching period T s The size of the AC side voltage is a function of the AC side output current i ac The linear relationship between the size of and the phase shift angle θ is an implementation scheme that can simplify the control algorithm, but it is not the only solution. Based on the power conversion circuit topology of the present invention and the principle of phase shift and frequency conversion hybrid control to achieve wide range soft switching, only changing θ and The relationship or / and change of switching period T s The functional relationship between θ and θ falls within the protection scope of the present invention.

[0093] The key to the single-stage bidirectional power conversion topology and control method proposed in the present invention, which can achieve wide-range soft switching, is that the intra-bridge and inter-bridge phase shift control of the dual half-bridge circuit and the high-frequency switch matrix provides the basis for achieving soft switching of all power tubes; the switching frequency is adjusted in real time according to the port voltage to achieve soft switching in a wide voltage variation range; the current of the bridge arm connected to the primary side of the second transformer is the sum of the primary current of the second transformer and the current of the inductor connected in parallel with the second transformer. By designing the inductance value of the inductor connected in parallel with the primary side of the second transformer, the current of the bridge arm connected to the primary side of the second transformer can be effectively changed, thereby expanding the soft switching implementation range of the switch tube of the bridge arm connected to the primary side of the second transformer under a wide power transmission range.

[0094] For structures such as ​ The converter was experimentally verified. Figure 6 shows the experimental waveforms of the drive voltage and drain-source voltage of the converter MOSFET switch tube under different power frequency voltage periods at light load (30% of the rated load). It can be seen from Figure 6 that all the switch tubes of the converter achieve zero voltage turn-on within the entire power frequency cycle under light load. Among them, the DC side switch tube S4 (S3), which is the most difficult to meet the soft switching conditions, also achieves zero voltage turn-on within the entire power frequency cycle under light load. The experimental results show that the power conversion topology and control method proposed in the present invention can achieve soft switching of all the switch tubes of the converter under a wide voltage range and a wide transmission power range.

Claims

1. A single-stage bidirectional power conversion topology capable of wide range soft switching, characterized by, Comprise: Double half-bridge circuit, first transformer, second transformer, inductor, power transfer inductor, high frequency switch matrix circuit and filter; The double half-bridge circuit is connected to the first port of the converter, and the first port is a direct current input or output port; the double half-bridge circuit comprises two capacitors C d1 and four power switches S1-S4 d2 The primary windings of the first transformer and the second transformer are connected in series, and the connection point of the two primary windings is connected to the connection point of the two capacitors C d1 and C d2 The primary winding of the second transformer is connected in parallel with an inductor, the secondary windings of the first transformer and the second transformer are connected in series, and the power transmission inductor is connected in series with the series-connected secondary windings; the two switches of each bridge arm of the double half-bridge circuit work complementarily with a 50% duty ratio, the switching frequency is the same as the switching frequency of the power tubes of the high-frequency switch matrix circuit, direct current is converted into a positive and negative amplitude symmetrical alternating square wave with a variable frequency, which is applied to the primary winding of the corresponding transformer, the switches of the two bridge arms are controlled by bridge internal phase shift control, the output of the series-connected secondary windings is a quasi-square wave, and the quasi-square wave width is determined by the bridge internal phase shift control angle θ; the power switches of the high-frequency switch matrix circuit are also modulated with a 50% duty ratio, and the power frequency alternating current or direct current voltage of the other port is converted into a 50% duty ratio high-frequency modulated alternating voltage, and the switching frequency of the high-frequency switch tube is determined by the instantaneous value of the voltage of the port; the voltages across the power transmission inductor are the quasi-square wave voltages output by the series-connected transformers and the alternating voltages modulated by the high-frequency switch matrix, and the phases of the two waveforms are determined by the bridge-to-bridge phase shift control angle between the double half-bridge circuit and the high-frequency switch matrix circuit; the filter is connected between the high-frequency switch matrix and the second port of the converter, and the second port is a direct current or alternating current output or input port.

2. The single-stage bidirectional power conversion topology capable of wide range soft switching according to claim 1, wherein, High frequency switch matrix circuit selects full-bridge matrix circuit, half-bridge matrix circuit or other circuit which can convert DC or low frequency AC voltage to high frequency alternating voltage output.

3. The single-stage bidirectional power conversion topology capable of wide range soft switching according to claim 1, wherein, Power transfer inductor is independent inductor or the sum of equivalent leakage inductance of secondary side of first transformer and second transformer, or composed of one or more independent inductors and leakage inductance of first transformer and second transformer.

4. The single-stage bidirectional power conversion topology capable of wide range soft switching according to claim 1, wherein, First transformer and second transformer are independent magnetic core transformers, or magnetic integrated transformers composed of single magnetic circuit combined magnetic core.

5. The single-stage bidirectional power conversion topology capable of wide range soft switching according to claim 1, wherein, Inductor in parallel with primary side of second transformer can be independent magnetic core, or single magnetic core integrated with second transformer.

6. The single-stage bidirectional power conversion topology capable of wide range soft switching according to claim 1, wherein, Filter is LC filter, C filter or other type of low-pass filter which allows low frequency current component to pass through AC or DC port while high frequency current component is filtered out.

7. The control method of the single-stage bidirectional power conversion topology capable of wide-range soft switching according to claim 1, characterized in that, Comprise following steps: Step 1. When used for AC / DC conversion, the magnitude of the AC side current i ac is derived from modal analysis in terms of the relationship of the control phase angle θ and ​ Step 2, shifting the phase angle of the inter-bridge control and the phase angle of the intra-bridge control θ is in proportional relationship, that is wherein the proportional coefficient k is a constant determined according to the optimization design of the circuit performance; Step 3, make AC current linearly related to phase shift control angle θ in bridge to simplify control strategy implementation.

8. The control method of the single-stage bidirectional power conversion topology capable of wide-range soft switching according to claim 7, wherein, In step 1, the magnitude of the AC side current i ac is derived from the modal analysis in terms of the relationship of the control phase shift angle θ and ​ In the formula: C=2 or 4 or other constant, when the amplitude of the high-frequency modulated AC voltage output by the high-frequency switch matrix circuit is equal to the amplitude of the AC side voltage, C=2, when the amplitude of the high-frequency modulated AC voltage output by the high-frequency switch matrix circuit is half of the amplitude of the AC side voltage, C=4; V dc is the voltage of the DC port, n is the turns ratio of the secondary side to the primary side of the first transformer and the second transformer, T s is the switching period, L lk is the power transmission inductance value including the equivalent leakage inductance of the secondary side of the two transformers.

9. The control method of the single-stage bidirectional power conversion topology capable of wide-range soft switching according to claim 7, wherein, In step 2, the magnitude of the AC side current i ac becomes:

10. The control method of the single-stage bidirectional power conversion topology capable of wide-range soft switching according to claim 7, wherein, In step 3, determine: Wherein, T smin is the minimum switching period of the converter switch tube determined according to the specific power level of the converter, the type of power device selected and the optimization of the comprehensive electrical performance of the system, and the switching period T s is determined by θ according to the following functional relationship: Switching frequency of high frequency switch tube changes with AC side voltage instantaneous value, when voltage is lower, switching frequency is higher; When AC side voltage changes sinusoidally, i.e. |v ac |=V acpeak |sinωt| where V acpeak is the peak value of the AC side voltage v ac , and θ is the phase-shift control angle of the bridge when unity power factor control is implemented. θ = θ peak |sinωt| where θ peak is the maximum value of the design-selected in-bridge phase-shift control angle θ for the double half-bridge. The switching period T s The relationship with the AC side voltage is:

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