Reactive power optimization control method suitable for series resonance dual-active bridge

Abstract

The invention discloses a reactive power optimization control method suitable for a series resonance dual active bridge, relates to the technical field of power electronic converter control, and solves the problem that an effective compensation scheme is not provided for an incomplete soft switching problem caused by dead time in the prior art. And the requirement of all-working-condition efficient operation in engineering application is difficult to meet. Based on triple phase shift control, the optimal phase shift angle combination is accurately solved through the power factor Lagrange function and the KKT condition, the reactive power of the primary side or the secondary side can be counteracted in a targeted manner, and the energy circulation of the resonant cavity is effectively inhibited; meanwhile, soft switching boundary analysis and critical phase compensation are combined, so that zero-voltage switching of all main switching tubes in a full-voltage gain and full-load range can be ensured, and switching loss is reduced; and a mode switching mechanism is designed in control logic to adapt to dynamic working condition response requirements, so that the energy transmission efficiency and working condition adaptability of the series resonance dual-active bridge converter are greatly improved.

Description

Technical Field

[0001] This invention relates to the field of power electronic converter control technology, and in particular to a reactive power optimization control method suitable for series resonant dual active bridges. Background Technology

[0002] With the increasing depletion of fossil fuels and the advancement of "dual carbon" goals, the large-scale grid connection of renewable energy sources such as photovoltaics and wind power, as well as energy storage devices, is driving the rapid development of DC microgrid technology. The Dual Active Bridge Series Resonant Converter (DBSRC) is widely used in DC microgrids and energy storage applications due to its advantages such as electrical isolation, bidirectional power transmission, and low current stress.

[0003] Traditional DBSRCs often employ single-phase shift (SPS) control, achieving power control solely by adjusting the external phase shift angle between the primary and secondary H-bridges. However, when the input and output voltages are mismatched (i.e., voltage gain...),... When operating under light load conditions, SPS control has significant drawbacks: high reactive power leads to energy circulation in the resonant cavity, resulting in a significant increase in reactive power on both the primary and secondary sides, increased conduction losses, and reduced transmission efficiency; the switching transistors struggle to achieve zero-voltage switching (ZVS) across all operating conditions, and hard switching is prone to occur under light load or when the voltage gain deviates from a unit value, leading to a sharp increase in switching losses; it cannot simultaneously meet the requirements of wide voltage gain and wide load, resulting in a significant decrease in efficiency and reliability in scenarios such as DC microgrid voltage fluctuations and energy storage system charging and discharging.

[0004] Among existing optimization schemes, while Extended Phase Shift (EPS) and Dual Phase Shift (DPS) controls can partially improve reactive power issues, they still have limitations such as insufficient control degrees of freedom and complex soft-switching constraints. Triple Phase Shift (TPS) control, although possessing three control degrees of freedom (primary side inner phase shift angle), still has limitations. Secondary side inward shift phase angle Outward phase angle However, a complete system of "reactive power optimization - soft switching constraint - closed-loop control" has not yet been formed. In particular, no effective compensation scheme has been proposed for the incomplete soft switching problem caused by dead time, which makes it difficult to meet the needs of efficient operation under all working conditions in engineering applications.

[0005] Therefore, a reactive power optimization control method suitable for series resonant dual active bridges is needed. Summary of the Invention

[0006] In order to solve the problem that the prior art does not propose an effective compensation scheme for the non-complete soft switching problem caused by the dead time, and it is difficult to meet the demand of efficient operation in all working conditions in engineering application, the application provides a reactive power optimization control method suitable for a series resonant dual active bridge, which can realize the minimization of reactive power and full switch ZVS in a wide voltage gain and wide load range through integrated design of steady-state modeling, soft switching constraint, power factor optimization, phase compensation and closed-loop control, and significantly improve the transmission efficiency and working condition adaptability of the converter. The specific technical scheme is as follows: A reactive power optimization control method suitable for a series resonant dual active bridge, comprising the following steps: Step one, based on the fundamental wave analysis method, a steady-state model of the converter is constructed, the fundamental wave components of the primary side and secondary side midpoint voltages are obtained through Fourier series expansion, the fundamental wave phasor of the resonant cavity current is calculated, and the power transmission equation containing the active power, the primary side reactive power, the secondary side reactive power and the total reactive power is derived; and according to the corresponding relationship between the resonant cavity current direction and the switch-on time of the switch, the phase shift angle combination constraint condition for realizing zero voltage switching of all switches is established, and the soft switching feasible region is determined; Step two, define the power factor of the converter and its power factor angle, construct the Lagrange function containing the maximum power factor objective function and the soft switching constraint condition, and solve the optimal phase shift angle combination; Step three, for the non-complete soft switching problem caused by the parasitic capacitance of the switch and the dead time, the phase compensation is carried out based on the principle that the charge transfer amount of the resonant current in the dead time should be greater than the charge and discharge charge amount of the parasitic capacitance; Step four, a closed-loop control strategy of phase shift angle and switch frequency cooperation is adopted: the voltage gain value is detected, and within the set gain dead zone range, it is switched to variable frequency single phase shift control, and the output power is controlled by adjusting the switch frequency and the outer phase shift angle; outside the gain dead zone range, based on the optimal phase shift angle combination and the phase compensation amount, the switch frequency is adjusted to make the output voltage track the reference value in combination with the output voltage closed-loop regulation, and finally the PWM signal for driving the primary and secondary H-bridge switches of the series resonant dual active bridge converter is generated according to the switch frequency and the final phase shift angle.

[0007] Preferably, in step two, the optimization strategy is determined according to the size relationship between the voltage gain and 1: when the voltage gain is less than 1, the inner phase shift angle of the secondary side is zero and the inner phase shift angle and the outer phase shift angle of the primary side are optimized to make the primary side reactive power zero; when the voltage gain is greater than 1, the inner phase shift angle of the primary side is zero and the inner phase shift angle and the outer phase shift angle of the secondary side are optimized to make the secondary side reactive power zero; when the voltage gain is equal to 1, the single phase shift control with the inner phase shift angles of the primary side and the secondary side being zero is adopted.

[0008] Preferably, in step three, the phase compensation is based on the principle that the amount of resonant current charge transferred in the dead time should be greater than the amount of charge of the parasitic capacitor charging and discharging, and specifically: when the voltage gain is less than 1, the primary side lagging bridge arm is compensated for the primary side internal phase angle when it is at the soft switching critical point; when the voltage gain is greater than 1, the secondary side leading bridge arm is compensated for the external phase angle when it is at the soft switching critical point, to obtain the final phase angle control amount.

[0009] Preferably, the zero voltage switching constraint condition in step one is obtained by the following method: The current direction criterion is determined that the resonant inductor current is negative at the primary side upper tube turn-on time, the primary side lower tube turn-on time, the secondary side upper tube turn-on time, and the secondary side lower tube turn-on time, and the soft switching inequality constraint containing the primary side internal phase angle, the secondary side internal phase angle, and the external phase angle is derived by combining the resonant cavity current time domain expression, and the overall soft switching feasible region is obtained by integrating the soft switching boundaries of the four bridge arms.

[0010] Preferably, the Lagrange function construction and solving in step two specifically includes: the power factor angle target function and the primary side H-bridge main switch tube soft switching constraint condition, the secondary side H-bridge main switch tube soft switching constraint condition are combined to form a Lagrange function, a Lagrange multiplier is introduced to convert the constraint optimization problem into an unconstrained optimization problem, and the Karush-Kuhn-Tucker condition is applied to obtain the analytical expression of the optimal phase angle combination.

[0011] Preferably, the compensation amount is obtained as follows: When the voltage gain is less than 1, the primary side bridge internal phase angle Δ α 1 can be calculated by the following formula: When the voltage gain is greater than 1, the primary side bridge internal phase angle Δ α 1 can be calculated by the following formula: Wherein, C 1 , C 2 is the parasitic parallel capacitance of the switch tube, U 1 、U 2 is the voltage of the parasitic capacitance before the switch tube is turned on, t d is the dead time interval, i sw is the resonant cavity current size.

[0012] Preferably, the resonant cavity current size i sw is expressed as follows: .

[0013] A computer-readable storage medium comprising a stored program, wherein the program, when executed, controls a device in which the computer-readable storage medium is located to perform the reactive power optimization control method for series resonant dual active bridge as described above.

[0014] A processor for running a program, wherein the program, when executed, performs the reactive power optimization control method for series resonant dual active bridge as described above.

[0015] Compared with the prior art, the present application has the following advantages: Based on triple-phase shift control, the present application can precisely solve the optimal phase shift angle combination through power factor Lagrange function and KKT condition, can counteract the reactive power of the primary side or the secondary side, can effectively suppress the resonant cavity energy circulation, can ensure that all main switch tubes realize zero voltage switching in the full voltage gain and full load range by combining soft switch boundary analysis and critical phase compensation, can reduce switching loss, and can design mode switching mechanism in the control logic to adapt to dynamic working condition response demand, and can greatly improve the energy transmission efficiency and working condition adaptability of the series resonant dual active bridge converter. BRIEF DESCRIPTION OF DRAWINGS

[0016] In order to more clearly illustrate the specific embodiments of the present application or the technical solutions in the prior art, the drawings needed in the specific embodiments or the prior art description will be briefly introduced below. In all the drawings, similar elements or parts are generally identified by similar reference numerals. In the drawings, each element or part is not necessarily drawn according to the actual scale.

[0017] Figure 1 is an embodiment example topology structure diagram of a series resonant dual active bridge converter of the present application; Figure 2 is a typical working waveform of an embodiment example of a series resonant dual active bridge converter of the present application; Figure 3 is a running boundary diagram of all switch tubes realizing soft switching of a series resonant dual active bridge converter of the present application when the voltage gain M=0.7 and the secondary side inner phase shift angle α2=0; Figure 4 is a typical waveform comparison diagram of a switch tube in a completely soft switching state in a series resonant dual active bridge converter of the present application; Figure 5 is a typical waveform comparison diagram of a switch tube in a non-completely soft switching state in a series resonant dual active bridge converter of the present application; Figure 6: is a typical waveform comparison chart of the hard switching state of the switch tube in the series resonant dual active bridge converter of the application; Figure 7 : is a principle diagram of the dead zone phase compensation of the primary side lag bridge arm in the reactive power optimization control strategy of the series resonant dual active bridge converter of the application; Figure 8 : is a principle diagram of the dead zone phase compensation of the primary side lag bridge arm in the reactive power optimization control strategy of the series resonant dual active bridge converter of the application; Figure 9 : is a principle diagram of the dead zone phase compensation of the secondary side lead bridge arm in the reactive power optimization control strategy of the series resonant dual active bridge converter of the application; Figure 10 : is a closed loop control block diagram of the reactive power optimization control strategy of the series resonant dual active bridge converter of the application; Figure 11 : is a simulation waveform chart of the resonant cavity of the traditional SPS control when the transmission power is 80W in the implementation of the reactive power optimization control strategy of the series resonant dual active bridge converter of the application; Figure 12 : is a simulation waveform chart of the resonant cavity of the reactive power optimization control when the transmission power is 80W in the implementation of the reactive power optimization control strategy of the series resonant dual active bridge converter of the application; Figure 13 : is a simulation waveform comparison chart of the resonant cavity of the traditional SPS control and the reactive power optimization control when the transmission power is 100W in the implementation of the reactive power optimization control strategy of the series resonant dual active bridge converter of the application; Figure 14 : is a resonant cavity experimental waveform chart under different voltage gains in the implementation of the reactive power optimization control strategy of the series resonant dual active bridge converter of the application; Figure 15 : is a resonant cavity experimental waveform comparison chart of the traditional SPS control and the reactive power optimization control when the transmission power is 150W in the implementation of the reactive power optimization control strategy of the series resonant dual active bridge converter of the application; Figure 16 : is a resonant cavity experimental waveform comparison chart of the traditional SPS control and the reactive power optimization control when the transmission power is 100W in the implementation of the reactive power optimization control strategy of the series resonant dual active bridge converter of the application; Figure 17 : is a relationship curve chart of the transmission efficiency and the transmission active power under the traditional SPS control and the reactive power optimization control strategy of the series resonant dual active bridge converter of the application. DETAILED DESCRIPTION

[0018] With reference to the drawings of the embodiments of the present application, the technical solutions in the embodiments of the present application will be clearly and completely described. Obviously, the described embodiments are a part rather than all of the embodiments of the present application. Based on the embodiments of the present application, all other embodiments obtained by a person of ordinary skill in the art without creative effort should fall within the protection scope of the present application.

[0019] It should be understood that the terms "comprising" and "including" as used in the specification and the appended claims indicate the presence of the described features, integers, steps, operations, elements, and / or components but do not preclude the presence or addition of one or more other features, integers, steps, operations, elements, components, and / or groups thereof.

[0020] It should also be understood that the terms used in the present application specification are only for the purpose of describing specific embodiments and are not intended to limit the present application. As used in the present application specification and the appended claims, the singular forms "a", "an" and "the" are intended to include the plural forms unless the context clearly indicates otherwise.

[0021] It should be further understood that the term "and / or" as used in the present application specification and the appended claims means any combination of one or more of the associated listed items and all possible combinations thereof.

[0022] The circuit topology of the series resonant dual active bridge converter described in the present application is shown in Figure 1 , which mainly includes a high-frequency transformer, a resonant cavity, a primary side H-bridge and a secondary side H-bridge. Among them, V 1 is a DC input voltage, V 2 is a DC output voltage, I LC is the primary side resonant cavity current. The primary H-bridge includes a first main switch , a second main switch , a third main switch , a fourth main switch , and each main switch is anti-parallel with a body diode; the secondary H-bridge includes a fifth main switch , a sixth main switch , a seventh main switch , an eighth main switch , and each main switch is anti-parallel with a body diode; the primary side H-bridge and the secondary side H-bridge respectively output the primary side midpoint voltage V AB and the secondary side midpoint voltage V CD. The high-frequency transformer not only realizes the electrical isolation of the primary and secondary sides, but also the leakage inductance can participate in resonant transmission power as part of the resonant inductance. The resonant cavity mainly includes a resonant inductance Lr and resonant capacitance C r is composed of, where L r is the sum of the transformer leakage inductance and auxiliary transmission inductance. The magnitude and direction of the transmitted power are controlled by adjusting the phase difference between the voltages at the midpoints of the two full-bridge sides. When the voltage at the midpoint of the primary H-bridge... V The phase of AB leads the voltage at the midpoint of the secondary H-bridge. V During CD phase, power is transferred in the forward direction; when the voltage at the midpoint of the primary-side H-bridge... V The phase of AB lags behind the voltage at the midpoint of the secondary H-bridge. V In CD phase, power is transmitted in reverse.

[0023] To simplify the analysis, the following reasonable assumptions are made based on the engineering application scenario: all power devices (switches, diodes), resonant elements ( L r , C r Both the transistor and the high-frequency transformer are ideal devices, and parasitic parameters are ignored. The influence of the parasitic capacitance of the switching transistor is only corrected in the subsequent phase compensation stage. The resonant current is approximately sinusoidal within the operating frequency range. The DC-side capacitor has a sufficiently large capacitance, and the input voltage... Output voltage Fluctuations of less than 5% can be approximated as constant values.

[0024] Typical DBSRC operating waveforms are as follows: Figure 2 As shown, α 1 represents the phase shift angle inside the primary H-bridge. α 2 represents the phase shift angle inside the secondary H-bridge. θ The phase shift angle between the primary H-bridge and the secondary H-bridge is... T s is the switching period. The inward phase shift duty cycle of the primary-side H-bridge is defined as... d 1, their numerical relationship is as follows α 1= d 1π; the inward phase duty cycle of the secondary H-bridge is defined as... d 2, their numerical relationship is as follows α 2= d 2π; the duty cycle of the outward phase shift between the primary and secondary H-bridges is defined as follows: d The numerical relationship is as follows: θ = d π. Phase-shift control can be achieved by combining different phase-shift angles (π). θ , α 1, α 2) To achieve multiple working modes, when α 1 ≠ α 2When ≠ 0, the converter works in triple-phase-shifted control mode. Triple-phase-shifted control mode provides flexible operation strategy and precise regulation ability for DBSRC.

[0025] In summary, for the series resonant dual active bridge converter, the core is to realize energy bidirectional transmission and electrical isolation through the coordinated work of the primary H-bridge, the secondary H-bridge and the LC series resonant tank. The typical working mode is around the commutation process in a switching cycle. In each switching cycle, the primary H-bridge and the secondary H-bridge generate bridge arm midpoint voltages through their respective switches VAB and VCD , and the difference between the two is added to the series resonant tank in the form of high-frequency alternating current, which is composed of a resonant inductor Lr and a resonant capacitor Cr, so as to excite an alternating resonant current iLC in the resonant tank. The resonant current is transmitted to the secondary side through the high-frequency transformer, realizing bidirectional energy transmission between the primary and secondary sides; the direction and size of the power are determined by the phase relationship between the bridge arms, the outer phase shift angle θ , the internal phase shift angle of the midpoint voltage α 1 , α 2 and the switching frequency fs . When a bridge arm on one side is turned off, the resonant current flows through the parasitic parallel capacitor and its anti-parallel diode during the dead time, thereby meeting the zero voltage switching (ZVS) condition to reduce switching loss.

[0026] On the other hand, the power transmission and soft switching characteristics of DBSRC depend on the coordinated control of phase shift angle and switching frequency. In single-phase-shifted control mode, only by adjusting the outer phase shift angle θ between the primary and secondary H-bridges can the transmission power be changed, θ when increased, the positive transmission power is increased, θ when decreased, the power is reduced; at the same time, the switching frequency f s needs to be close to the resonant frequency fᵣ , so that the LC resonant tank presents inductive impedance, ensuring i LC that the switching tube diode is turned on in advance to realize soft switching. Although the SPS control logic is simple, it can achieve good soft switching effect and lower reactive power in the unit operation condition with voltage gain M =1, but this ideal state depends on the matching of input and output voltages: when M deviates from 1 or the load fluctuates, the sinusoidal degree of the resonant tank current waveform decreases, the soft switching range narrows, and energy circulation is easy to occur on the primary side and the secondary side, resulting in an increase in reactive power.

[0027] The reactive power optimization control strategy described in the application is based on triple phase shift (TPS) control, and combines power factor optimization and soft switch phase compensation for closed loop mode switching. The specific implementation steps are as follows: 1. Steady-state modeling and soft switch boundary analysis Based on the first harmonic approximation (FHA) method, a DBSRC steady-state model is constructed, only considering the fundamental component of the resonant cavity current, and ignoring the influence of high-order harmonics: Step 1.1 Get the fundamental expression of the primary H bridge midpoint voltage , the secondary H bridge midpoint voltage (reduced to the primary side) by Fourier series expansion: Step 1.2 Combine the resonant cavity impedance X LC =jꞷL r +1 / jꞷC r , the resonant cavity current fundamental phase: Further get the converter transmission active power P o , the primary side reactive power Q 1 , the secondary side reactive power Q 2 and the total reactive power of the converter Q : Where M is the voltage gain, the ratio of the output voltage to the input voltage after conversion to the primary side, Z r is the characteristic impedance, , F n is the switching resonant frequency ratio, i.e. the ratio of switching frequency to resonant frequency.

[0028] Step 1.3 Based on the correspondence between the direction of the resonant cavity current i LC and the turn-on time of the switch, according to Figure 1The current positive direction shown, can be concluded that the ZVS implementation conditions: A bridge arm to achieve ZVS, the resonant inductor current needs to be negative when the upper tube S1 open time; B bridge arm to achieve ZVS, the resonant inductor current needs to be negative when the lower tube S4 open time; C bridge arm to achieve ZVS, the resonant inductor current needs to be positive when the upper tube S5 open time; D bridge arm to achieve ZVS, the resonant inductor current needs to be positive when the lower tube S8 open time. Based on the above analysis, the constraint conditions of all switch tubes of DBSRC soft switching can be derived, the specific expression is as follows: In the formula, i ( S x) represents the current of the resonant cavity when the switch tube S x is turned on, x = 1, 4, 5, 8 The phase shift angle combination of all switch tubes zero voltage turn-on can be obtained by combining the above formula with the time domain expression of the resonant cavity current ( θ 、 α 1、 α 2) The optimization constraint conditions are as follows: In order to intuitively express the relationship between the soft switching boundary and the phase shift angle, by combining the soft switching boundaries of the four bridge arms, the overall boundary of all switch tubes soft switching (ZVS) can be obtained, as Figure 3 shown. The shaded part in the figure represents that DBSRC can achieve soft switching of all switch tubes in this range.

[0029] 2. Phase shift angle optimization based on maximum power factor Step 2.1 Introduce key parameters, converter power factor PF and power factor angle β . Among them, the power factor PF is defined as the ratio of active power to apparent power, and the power factor angle β represents the phase difference between the voltage and current fundamental components. The expressions of power factor PF and power factor angle β are as follows: Step 2.2 Construct the Lagrange function, combine the power factor optimization target with the soft switching constraint.

[0030] Among them, is the power factor angle target function, is the soft switching constraint condition of the main switch tube of the primary side H bridge, The soft switching constraint condition of the secondary side H-bridge main switch tube, , is the Lagrange multiplier; Step 2.3. Solve the above function by Karush-Kuhn-Tucker (KKT) condition to obtain the optimal phase shift angle combination: When M <1, the optimal solution is α 2 =0 (no internal phase shift of the secondary side), α 1 , θ satisfies Q 1 =0; when M >1, the optimal solution is α 1 =0 (no internal phase shift of the primary side), α 2 , θ satisfies Q 2 =0; when M =1, α 1 = α 2 =0, at this time the single phase shift control strategy achieves the optimal control effect.

[0031] 3. Critical soft switching phase compensation In view of the "non-complete soft switching" problem caused by the parasitic capacitance and dead time of the switch tube in the actual circuit as shown in Figure 4-6 , based on the soft switching boundary of step 1.3, the phase shift angle of the critical state is compensated: Step 3.1. Perform compensation logic judgment, when M <1, the primary side lagging leg ( S 2 , S 3 ) is at the soft switching critical point, and the primary side internal phase shift angle ∆α 1 needs to be compensated; when M >1, the secondary side leading leg ( S 5 , S 8 ) is at the soft switching critical point, and the external phase shift angle ∆θ needs to be compensated; Step 3.2. Calculate the phase angle compensation amount, based on the relationship that the resonant current charge transfer amount within the dead time should be greater than the parasitic capacitance charge and discharge charge amount, and deduce the compensation angle correction value. For exampleFigure 7-8 As shown, when M <1, the phase-shift angle of the primary side bridge that needs to be compensated ∆ α 1 can be calculated by the following formula: As shown, when Figure 9 As shown, when M >1, the phase-shift angle of the primary side bridge that needs to be compensated ∆ α 1 can be calculated by the following formula: Where, C 1 , C 2 C is the parasitic parallel capacitance of the switch tube, U 1 、U 2 V is the voltage of the parasitic capacitance before the switch tube is turned on, t d T is the dead time interval. i sw I is the resonant tank current, and the expression is: Step 3.3 Compensation of the phase-shift angle: the final control phase-shift angle is α 1 '=α 1 +∆α 1 ( M <1), θ'=θ+∆θ ( M >1), to ensure that the parasitic capacitance is fully charged and discharged within t d , and all switch tubes achieve ZVS.

[0032] 4. Power factor optimization closed-loop control Step 4.1 Based on the above derivation, considering the maximum power factor of DBSRC and the zero-voltage turn-on of all switch devices, the transmission power of the converter can be obtained as: As can be seen from the above formula, when the switching frequency and the parameters of the converter are determined, the optimization of the phase-shift angle and the transmission power only depends on the DC voltage gain M of the converter. This relationship leads to the fact that the optimization target can only be achieved at a certain load.

[0033] To achieve the power factor optimization in the full load and full voltage range, the variable frequency phase-shifted control strategy is introduced to achieve the optimization goal. When the voltage gain is constant, the output power is controlled by adjusting the switching frequency, and the optimization in the full power range is achieved. When the output power is constant, the voltage gain is adjusted by adjusting the switching resonance frequency ratio and the phase shift angle at the same time, and the optimization in the full voltage range is achieved.

[0034] Step 4.2 According to the transmission power expression of the transformer in step 4.1, under the power factor optimization algorithm, when the voltage gain M =1, the output power of DBSRC is 0. In order to make the transformer work normally in the full voltage range, when 0.96 M <1.04, the reactive power is small, and the variable frequency single phase-shift control is switched, and α 1 =α 2 =0 The output power is controlled by adjusting the switching frequency f s and the outer phase shift angle θ ; When M ≤0.96 or M ≥1.04: the phase shift angle + switching frequency is adjusted cooperatively, based on the optimal phase shift angle α 1 ,α 2 ,θ ) in step 2 and the compensation amount in step 3, combined with the output voltage closed loop (PI regulator) to adjust f s , so that the output voltage tracks the reference value V o _ ref ; the power factor optimization algorithm closed loop control block diagram is shown in Figure 10 Step 4.3 PWM signal generation: the controller generates 8-way PWM drive signal according to the final phase shift angle α 1 ',α 2 ',θ' ) and the switching frequency f s , and outputs to the primary and secondary H-bridge switch tubes through the isolation drive circuit.

[0035] In order to verify the effectiveness of the power factor optimization algorithm of the proposed series resonant dual active bridge converter, a simulation model is built, and the traditional SPS control and the reactive power optimization control (Optimal Power Factor, OPF) are simulated and compared, Figure 11-12 and​Figure 13 The resonant cavity waveform diagrams of traditional SPS control and reactive power optimization control are given respectively when the transmission power is 80W and 100W. It can be seen that under SPS control, the secondary side switch tube appears hard switching phenomenon, and there is a lot of reactive power on the primary side, which causes energy circulation in the resonant cavity, increases the on-state loss and reduces the efficiency. Under the control of reactive power optimization control, all switch tubes of the converter achieve soft switching, the reactive power on the primary side is eliminated, and the total reactive power is significantly reduced. The power factor of the converter controlled by reactive power optimization control is significantly greater than that of SPS control.

[0036] To verify the effectiveness of the proposed power factor optimization algorithm of series resonant dual active bridge converter, a series resonant dual active bridge experimental prototype is built, and the proposed reactive power optimization control strategy is applied. In order to verify the optimization effect of the optimization algorithm in the full voltage range, the voltage gain M=0.75, M=1, M=1.25 is selected, the DBSRC reactive power and the soft switching realization of switch tube are observed, Figure 14 The resonant cavity waveform of reactive power optimization control under different voltage gains is shown in the experimental results, which shows that the reactive power optimization control can maintain the total reactive power at a low level while achieving full switch tube soft switching.

[0037] To verify the performance of the proposed optimization algorithm in the full power range, under the voltage gain M=1.25, 50% and 75% rated load conditions are tested respectively, the reactive power and soft switching characteristics of DBSRC are observed, and compared with SPS control. The results are as Figure 15 With Figure 16 As can be seen from the figure, under SPS control, the primary side switch tube appears hard switching phenomenon, and there is a lot of reactive power on the secondary side. While under the proposed reactive power optimization control, all switch tubes achieve soft switching, the reactive power on the secondary side is eliminated, and the total reactive power is significantly reduced.

[0038] Figure 17 The relationship curve between transmission efficiency and transmission active power of DBSRC under SPS control and reactive power optimization control when M=1.25. It can be seen that the reactive power optimization control can improve the operating efficiency of the converter in a wide load range, especially the optimization effect is more significant at light load.

[0039] Simulation and experiment show that the proposed reactive power optimization control strategy of series resonant dual active bridge can realize reactive power minimization and full switch tube ZVS in a wide voltage gain and wide load range, and significantly improve the transmission efficiency of the converter.

[0040] Those skilled in the art can understand that the units of the examples described in combination with the embodiments disclosed herein can be realized in electronic hardware, computer software or a combination of both. In order to clearly illustrate the interchangeability of hardware and software, the components of the examples have been described in general terms in the above description. Whether the functions are performed in hardware or software depends on the specific application and design constraints of the technical solution. The skilled person can use different methods to implement the described functions for each specific application, but such implementation should not be considered beyond the scope of the present application.

[0041] In the embodiments provided by the present application, it should be understood that the division of units is only a logical functional division, and there can be another division manner in actual implementation, for example, multiple units can be combined into one unit, one unit can be split into multiple units, or some features can be ignored, etc.

[0042] In addition, each functional unit in each embodiment of the present application can be integrated in one processing unit, or each unit can exist physically separately, or two or more units can be integrated in one unit. The integrated unit can be realized in the form of hardware or in the form of a software functional unit.

[0043] The integrated unit, if realized in the form of a software functional unit and sold or used as an independent product, can be stored in a computer readable storage medium. Based on such understanding, the technical solutions of the present application essentially or the part that contributes to the prior art or the whole or part of the technical solutions can be embodied in the form of a software product. The computer software product is stored in a storage medium and includes a plurality of instructions for causing a computer device (which can be a personal computer, a server or a network device, etc.) to execute all or part of the steps of the method described in each embodiment of the present application. The aforementioned storage medium includes: a U disk, a read-only memory (ROM, Read-Only Memory), a random access memory (RAM, Random Access Memory), a mobile hard disk, a magnetic disk or an optical disk, and various media that can store program codes.

[0044] Finally, it should be noted that: the above embodiments are only used to illustrate the technical solutions of the present application, and not to limit them; although the present application has been described in detail with reference to the foregoing embodiments, those skilled in the art should understand that: it can still modify the technical solutions recorded in the foregoing embodiments, or make equivalent replacement for part or all of the technical features; and these modifications or replacements do not make the essence of the corresponding technical solutions deviate from the scope of the technical solutions of the embodiments of the present application, and they should be covered in the scope of the claims and the description of the present application.

Claims

1. A reactive power optimization control method applicable to series resonant dual active bridge circuits, characterized in that, Includes the following steps: Step 1: Construct a steady-state model of the converter based on the fundamental analysis method. Obtain the fundamental components of the voltage at the midpoint of the primary and secondary sides through Fourier series expansion, calculate the fundamental phasor of the resonant cavity current, and derive the power transfer equation including active power, reactive power on the primary side, reactive power on the secondary side, and total reactive power. Based on the correspondence between the direction of the resonant cavity current and the turn-on time of the switching transistors, establish the phase shift angle combination constraint conditions for all switching transistors to achieve zero-voltage switching, and determine the feasible region of soft switching. Step 2: Define the converter power factor and its power factor angle, construct a Lagrangian function that includes the objective function of maximizing the power factor and the soft-switching constraint, and solve for the optimal combination of phase shift angles; Step 3: To address the incomplete soft-switching problem caused by the parasitic capacitance and dead time of the switching transistor, phase compensation is performed based on the principle that the charge transfer of the resonant current during the dead time should be greater than the charge charge of the parasitic capacitance. Step 4: Adopt a closed-loop control strategy that coordinates phase shift angle and switching frequency: detect the voltage gain value, switch to frequency conversion single phase shift control within the set gain dead zone, and control the output power by adjusting the switching frequency and the external phase shift angle. Outside the gain dead zone, based on the optimal phase shift angle combination and phase compensation, the output voltage is adjusted by the closed-loop control of the switching frequency to make the output voltage track the reference value. Finally, the PWM signal driving the H-bridge switches of the primary and secondary sides of the series resonant dual active bridge converter is generated according to the switching frequency and the final phase shift angle.

2. The reactive power optimization control method applicable to a series resonant dual active bridge according to claim 1, characterized in that, In step two, the optimization strategy is determined segment by segment based on the relationship between voltage gain and 1: when the voltage gain is less than 1, the inward phase angle of the secondary side is set to zero and the inward and outward phase angles of the primary side are optimized to make the reactive power on the primary side zero; when the voltage gain is greater than 1, the inward phase angle of the primary side is set to zero and the inward and outward phase angles of the secondary side are optimized to make the reactive power on the secondary side zero; when the voltage gain is equal to 1, single-phase shift control with both the inward and outward phase angles of the primary and secondary sides being zero is adopted.

3. The reactive power optimization control method applicable to a series resonant dual active bridge according to claim 2, characterized in that, In step three, phase compensation is performed based on the principle that the amount of charge transfer of the resonant current during the dead time should be greater than the amount of charge charge of the parasitic capacitance. Specifically, when the voltage gain is less than 1, the primary side lagging bridge arm at the soft switching critical point is compensated for the internal phase shift angle; when the voltage gain is greater than 1, the secondary side leading bridge arm at the soft switching critical point is compensated for the external phase shift angle, thus obtaining the final phase shift angle control amount.

4. The reactive power optimization control method applicable to a series resonant dual active bridge according to claim 1, characterized in that, The zero-voltage switching constraint in step one is obtained in the following way: The current direction criteria for the resonant inductor current are determined when the primary side upper transistor is negative, the primary side lower transistor is negative, the secondary side upper transistor is positive, and the secondary side lower transistor is positive. Based on the time-domain expression of the resonant cavity current, a soft-switching inequality constraint including the primary side inner phase shift angle, the secondary side inner phase shift angle, and the secondary side outer phase shift angle is derived. The overall soft-switching feasible region is obtained by integrating the soft-switching boundaries of the four bridge arms.

5. The reactive power optimization control method applicable to a series resonant dual active bridge according to claim 2, characterized in that, The construction and solution of the Lagrangian function in step two specifically includes: constructing the Lagrangian function by combining the power factor angle objective function with the soft-switching constraints of the primary H-bridge main switch and the secondary H-bridge main switch; introducing Lagrange multipliers to transform the constrained optimization problem into an unconstrained optimization problem; and applying the Karush-Kuhn-Tucker conditions to obtain the analytical expression of the optimal phase shift angle combination.

6. The reactive power optimization control method applicable to a series resonant dual active bridge according to claim 3, characterized in that, The process of obtaining the compensation amount is as follows: When the voltage gain is less than 1, the required compensation for the phase shift angle ∆ inside the primary bridge is... α 1 can be calculated using the following formula: When the voltage gain > 1, the required compensation for the phase shift angle ∆ inside the primary bridge is... α 1 can be calculated using the following formula: in, C 1 , C 2 Parasitic parallel capacitance of the switching transistor, U 1 、U 2 This represents the voltage across the parasitic capacitance before the switching transistor is turned on. t d Dead time interval i sw This represents the magnitude of the resonant cavity current during switching.

7. The reactive power optimization control method applicable to a series resonant dual active bridge according to claim 6, characterized in that, Magnitude of resonant cavity current during switching i sw The expression is as follows: 。 8. A computer-readable storage medium, characterized in that, The computer-readable storage medium includes a stored program, wherein, when the program is executed, it controls the device containing the computer-readable storage medium to perform the reactive power optimization control method applicable to a series resonant dual active bridge as described in any one of claims 1 to 7.

9. A processor, characterized in that, The processor is used to run a program, wherein the program executes the reactive power optimization control method applicable to a series resonant dual active bridge as described in any one of claims 1 to 7.

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