A Nonlinear Compensation Circuit and Method for a Signal Acquisition Probe

Abstract

The present invention discloses a non-linear compensation circuit for a signal acquisition probe. The circuit includes a sampling resistor. By using the sampling resistor and an equipotential input terminal arranged at one end of the sampling resistor, and by using a two-stage amplification method, during the first-stage amplification, a sampling resistor R2 network is used in cooperation with differential input to separate the signal / noise paths. And a dynamic potential compensation system is constructed before the amplified input to dynamically monitor and shield noise, filter the noise after the noise method, and at the same time use a filter network to realize phase compensation. Finally, the latter-stage amplifier is used to complete noise cancellation and main signal amplification. The circuit of the present invention overcomes the grounding noise interference caused by distributed multiple grounding points in the prior art. At the same time, due to the addition of dynamic potential compensation, high-sensitivity detection and noise compensation are realized. The present invention also provides a method based on the above non-linear compensation circuit.

Description

Technical Field

[0001] The present invention relates to the field of electrical signal detection, and more specifically, to a non-linear compensation circuit and method for a signal acquisition probe. Background Art

[0002] Coaxial cables are commonly used carriers for high-frequency signal transmission. Generally, they include a copper core, an inner insulating layer located outside the copper core, a metal shielding layer wrapped around the insulating layer, and an outer insulating layer located outside the metal shielding layer. Coaxial cables can transmit high-frequency signals, and at the same time, the metal shielding layer can shield most external interferences.

[0003] Currently, when using coaxial cables, the metal shielding layer is grounded. Longer coaxial cables require multiple grounding points. Grounding the coaxial cable easily generates current, causing voltage drop, ground bounce, and ground interference, ultimately leading to measurement errors in the coaxial cable. In addition, for the measurement system of coaxial cables, the measurement system often uses a DC-DC power supply to achieve low power consumption and small size.

[0004] When measuring a multi-point grounded coaxial cable, due to the large current on the coaxial cable, the cable will generate a voltage drop, and a large current will also flow through the coaxial cable. Moreover, this current is not constant and does not have an ideal linear relationship with the output signal. This interference is a distributed interference formed by the multi-point grounded common ground end.

[0005] When it is necessary to collect signals from a multi-point grounded coaxial cable, these interferences need to be removed.

[0006] Specifically, from the signal sending end to the receiving end of the coaxial cable, multiple data relaying is required, and each data relaying has a grounding end. When measuring the coaxial signal at the receiving end, distributed loop interference of multiple grounding ends is naturally introduced.

[0007] In a multi-point grounded coaxial cable system (the number of grounding points n≥2), the shielding layer forms a closed loop, and its equivalent circuit satisfies Kirchhoff's voltage law:

[0008]

[0009] Among them, , represents the distributed impedance of the shielding layer of the k-th segment.

[0010] When there is a ground potential difference between different grounding points: When, a circulating current will be generated: , among which, is the additional impedance at the position of the remote grounding path. For example: when the length of the coaxial cable exceeds , (where λ is the wavelength corresponding to the interference frequency), the circulating current in a 10-meter coaxial cable can reach 12% - 18% of the signal current at 30 MHz.

[0011] In addition, when the measurement system uses a DC-DC power supply, and the common DC-DC power supply is of the Buck type, then, , that is, the inductor current ripple; where, is the switching frequency of the switching power supply; the high-frequency oscillation waveform of the switching node contains fast edges, and this fast edge , the voltage derivative of the fast edge is: , this voltage reciprocal is coupled to the shield layer through the parasitic capacitance, and a displacement current is formed between the shield layer and the ground terminal. The displacement current is: .

[0012] Assume: , when , then the displacement current , 10 mA of

[0013] The displacement current is sufficient to form parasitic modulation in the transmission of sensitive high-frequency signals. Due to the existence of the parasitic modulation signal, if one wants to restore the original modulated high-frequency signal during measurement, the parasitic modulation signal must also be removed. Finally, for coaxial cables, the high-frequency dynamic current will form a magnetic field around the shield layer. The inductance per unit length of the shield layer is: ; when multiple coaxial cables are transmitting in close proximity, when the current change in the adjacent lines, then the induced electromotive force is: , which is the mutual inductance; if the mutual inductance coefficient k = 0.2 between the two coaxial cables and the parasitic inductance , then the coupling noise per centimeter is: , and the noise at this level is already sufficient to affect the accuracy of signal transmission and detection. Summary of the Invention

[0014] The purpose of the present invention is to provide a non-linear compensation circuit for a signal acquisition probe, aiming to solve the technical problem of distributed grounding interference caused by multi-point grounding in the prior art.

[0015] The non-linear compensation circuit for a signal acquisition probe provided by the present invention includes:

[0016] Detection terminals, the detection terminals are electrically connected to the coaxial cable. The coaxial cable includes a first copper core and a second copper core. A first shield layer is arranged outside the first copper core, and a second shield layer is arranged outside the second copper core; the first copper core has a first resistance, the second copper core has a third resistance, the first shield layer has a fifth resistance, and the second shield layer has a seventh resistance;

[0017] A sampling resistor, the first end of the sampling resistor is connected to the first copper core, the second end of the sampling resistor is respectively connected to the second copper core, the first shielding layer and the second shielding layer, and the first shielding layer is grounded;

[0018] Wherein, the first end of the sampling resistor is connected to the negative input terminal of the second amplifier through a first resistor and a second resistor in sequence;

[0019] The second end of the sampling resistor is respectively connected to the equipotential input terminal through a third resistor and a fifth resistor, and the third resistor and the fifth resistor are connected in parallel;

[0020] The equipotential input terminal is connected to the negative input terminal of the first amplifier through a sixth resistor; the equipotential input terminal is connected to the positive input terminal of the second amplifier through the fourth resistor and to the negative input terminal of the first amplifier through the sixth resistor;

[0021] The sampling resistor is connected to the positive input terminal of the first amplifier through a seventh resistor; the output terminal of the first amplifier is connected to the negative input terminal of the second amplifier through a ninth resistor; the output terminal of the first amplifier is connected to the negative input terminal of the first amplifier through an eighth resistor, and a capacitor is connected in parallel at both ends of the eighth resistor; the negative input terminal of the second amplifier is connected to the output terminal of the second amplifier through a tenth resistor; the output terminal of the second amplifier is connected to the probe output terminal through an eleventh resistor.

[0022] The circuit disclosed by the present invention achieves the goal through a three - stage architecture:

[0023] 1 - The first stage (input conditioning): The sampling resistor R2 network cooperates with differential input, that is, the fifth resistor RL3 and the seventh resistor RL4 enter the differential amplification of the first amplifier U12, so as to separate the signal / noise path.

[0024] 2 - The second stage (common - mode rejection): The first amplifier U12 constructs a dynamic ground - potential compensation system.

[0025] 3 - The third stage (signal synthesis): The second amplifier U1 completes noise cancellation and main - signal amplification.

[0026] The present invention also discloses a non - linear compensation method for a signal acquisition probe during the implementation process of the above - mentioned circuit, and the method includes:

[0027] Step S1: Sample the current between the first copper core and the second copper core and convert it into a voltage;

[0028] Step S2: Sample the noise signal between the first copper core and the first shielding layer;

[0029] Step S3: An equipotential input terminal is set between the second copper core and the first shielding layer, so that a dynamic potential balance point is formed between the signal on the second copper core and the first shielding layer, thereby reducing the influence of displacement current caused by multi-point grounding on the coaxial cable;

[0030] Step S4: The noise between the first shielding layer and the second shielding layer is differentially amplified. The common-mode interference is eliminated by differential amplification, the influence on the signal on the first copper core is isolated, and the amplified noise is filtered;

[0031] Step S5: The filtered noise signal is superimposed on the signal on the first copper core, and after the superposition, it is amplified with the signal on the second copper core and then output to the detection end;

[0032] Among them, the filtered noise signal is used for phase compensation of the signal on the first copper core.

[0033] This method separates the signal / noise path, constructs a dynamic ground potential compensation system, completes noise cancellation and main signal amplification, and can also achieve phase compensation. Description of the Drawings

[0034] Figure 1 is the circuit schematic diagram of the present invention;

[0035] Figure 2 is the flowchart of the method of the present invention. Detailed Embodiment

[0036] The present invention will be further described and explained below with reference to specific embodiments and the accompanying drawings of the specification:

[0037] Please refer to Figure 1 , the present invention discloses a non-linear compensation circuit for a signal acquisition probe. The circuit includes: a detection terminal current, the detection terminal current is electrically connected to a coaxial cable X, and the other end of the detection terminal current is connected to a detection device, such as an oscilloscope, a spectrum analyzer, etc. through a probe. Among them, the coaxial cable X includes a first copper core x1 and a second copper core x2. A first shielding layer y1 is arranged outside the first copper core x1, and a second shielding layer y2 is arranged outside the second copper core x2; the first copper core x1 has a first resistor RL1, the second copper core x2 has a third resistor RL2, the first shielding layer y1 has a fifth resistor RL3, and the second shielding layer y2 has a seventh resistor RL4.

[0038] Among them, the detection terminal current includes a sampling resistor R2. The first end of the sampling resistor R2 is connected to the first copper core x1, and the second end of the sampling resistor R2 is respectively connected to the second copper core x2, the first shielding layer y1, and the second shielding layer y2. The first shielding layer y1 is grounded to GND1.

[0039] Among them, the first end of the sampling resistor R2 is sequentially connected to the negative input terminal of the second amplifier U1 through the first resistor RL1 and the second resistor R8; the second end of the sampling resistor R2 is respectively connected to the equipotential input terminal GND through the third resistor RL2 and the fifth resistor RL3, and the third resistor RL2 and the fifth resistor RL3 are connected in parallel; the equipotential input terminal GND is connected to the negative input terminal of the first amplifier U12 through the sixth resistor R5; the equipotential input terminal GND is connected to the positive input terminal of the second amplifier U1 through the fourth resistor R65 and is connected to the negative input terminal of the first amplifier U12 through the sixth resistor R5; the sampling resistor R2 is connected to the positive input terminal of the first amplifier U12 through the seventh resistor RL4; the output terminal of the first amplifier U12 is connected to the negative input terminal of the second amplifier U1 through the ninth resistor R64; the output terminal of the first amplifier U12 is connected to the negative input terminal of the first amplifier U12 through the eighth resistor R1, and a capacitor C1 is connected in parallel at both ends of the eighth resistor R1; the negative input terminal of the second amplifier U1 is connected to the output terminal of the second amplifier U1 through the tenth resistor R10; the output terminal of the second amplifier U1 is connected to the probe output terminal through the eleventh resistor R6;

[0040] Among them, the sampling resistor R2 is less than or equal to 1Ω, the resistance value of the second resistor R8 is 249Ω, the fourth resistor R65 is 49.9Ω, the sixth resistor R5 is 1K, the eighth resistor R1 is 1K, the ninth resistor R64 is 10K, the tenth resistor R10 is 330Ω, and the eleventh resistor R6 is 49.9Ω; the capacitor C1 is 10pF.

[0041] Among them, the sampling resistor R2 is a resistor made of alloy material to reduce the error influence caused by sampling temperature drift.

[0042] Among them, the equipotential input terminal GND is used to form a redundant current path for the parallel-connected third resistor (RL2), fifth resistor (RL3), and seventh resistor (RL4) and establish a common reference ground to discharge the high-frequency noise on the second copper core x2 and the first shielding layer y1.

[0043] Among them, the negative input terminal of the first amplifier U12 is connected to the equipotential input terminal GND to break the circulating current loop on the first shielding layer y1 and / or the second shielding layer y2 of the coaxial cable X, thereby reducing the circulating current grounding interference.

[0044] Among them, the positive input terminal of the first amplifier U12 is connected to the second shielding layer y2 to detect the potential fluctuation caused by the displacement current on the second shielding layer y2 in real time through the seventh resistor RL4. The first amplifier U12 is used to amplify the potential fluctuation to the output terminal of the first amplifier U12, and a filtering and feedback branch formed by the eighth resistor R1 and the capacitor C at the output terminal of the first amplifier U12 is used to achieve noise cancellation and maintain the feedback gain, thereby reducing the influence of the noise at the front end of the first amplifier U12.

[0045] Next, the overall design principle, working process, and functions of the components of this circuit will be described.

[0046] I. Overall Circuit Architecture and Core Functional Requirements

[0047] This circuit is designed for the precise current detection of high-frequency coaxial cable X, and corely solves the following five major technical problems:

[0048] a - Distributed ground loop interference;

[0049] b - Interference coupling of shielding layer displacement current;

[0050] c - High-frequency influence of parasitic modulation effect;

[0051] d - Waveform distortion caused by phase shift;

[0052] e - Improving the acquisition accuracy of microvolt-level signals.

[0053] This application achieves the goal through a three-stage architecture:

[0054] 1 - The first stage (input conditioning): The sampling resistor R2 network cooperates with differential input, that is, the fifth resistor RL3 and the seventh resistor RL4 enter the differential amplification of the first amplifier U12, thereby separating the signal / noise path.

[0055] 2 - The second stage (common-mode rejection): The first amplifier U12 constructs a dynamic ground potential compensation system.

[0056] 3 - The third stage (signal synthesis): The second amplifier U1 completes noise cancellation and main signal amplification.

[0057] II. Layer-by-Layer Analysis of Key Modules

[0058] 1. Detection Terminal current and Shielding Network Design

[0059] Sampling Resistor R2 (≤1Ω):

[0060] The sampling resistor R2 is arranged between the first copper core x1 (the self-resistance of the first copper core is the first resistance RL1) and the second copper core x2 (the self-resistance of the second copper core is the third resistance RL2) to form a current-voltage conversion core.

[0061] Technical key points: The impedance of the sampling resistor R2 below 1Ω ensures that the voltage drop during the measurement of the cable current does not cause a temperature drift (assuming only 1V loss when the full scale is 1A).

[0062] Multi-path reflux design: The second end of the sampling resistor R2 is connected in parallel with the second copper core x2 (corresponding to the third resistance RL2), the first shielding layer y1 (corresponding to the fifth resistance RL3), and the second shielding layer y2 (corresponding to the seventh resistance RL4) to form a redundant current path.

[0063] Shielding layer processing mechanism:

[0064] The first shielding layer y1 (GND1 is directly grounded): Establish a main reference ground plane to discharge high-frequency noise to the low-impedance ground. The second shielding layer y2 (through the seventh resistance RL4 → the positive input terminal of the first amplifier U12): Dynamically monitor the displacement current and detect and compensate through the operational amplifier. Hybrid grounding strategy: The three-terminal parallel connection uses star grounding and dynamic potential tracking, that is, when the second copper core x2, the first shielding layer y1, and the second shielding layer y2 are connected to the input terminal of the first amplifier U12, a three-point star connection is formed. The star grounding can destroy the ground loop structure of the traditional multi-layer shielding.

[0065] 2. Construction of the equipotential input terminal GND

[0066] Implementation method: The parallel connection point of the second copper core x2 and the first shielding layer y1, that is, the equipotential input terminal GND, and its equivalent impedance , for example, when RL2 = 0.1Ω and RL3 = 0.05Ω, Z GND ≈0.033Ω. At this time, the role of the equipotential input terminal GND: Provide a reference benchmark point for the signal processing circuit; Bypass high-frequency noise components to the ground path; Actively correct potential offsets through the operational amplifier.

[0067] 3. The noise cancellation loop of the first amplifier U12

[0068] The positive input terminal of the first amplifier U12: Samples the voltage drop generated by the displacement current through the seventh resistance RL4 (the parasitic resistance of the second shielding layer y2). The negative input terminal of the first amplifier U12: Obtains the voltage reference of the equipotential input terminal GND through the fifth resistance R5 (1kΩ) on the first shielding layer y1.

[0069] The feedback network of the first amplifier U12 is composed of the eighth resistance R1 (1kΩ) and the capacitor C1 (10pF) connected in parallel to form a composite feedback. The composite feedback includes:

[0070] DC Channel: The eighth resistor R1 maintains a closed-loop gain of 1 to ensure a stable operating point.

[0071] High-Frequency Filtering: Forms a low-pass characteristic at 159 MHz (f = 1 / (2πRC)) to filter out UHF band noise.

[0072] Phase Compensation: Deals with the transmission delay of high-speed signals to prevent self-excited oscillation.

[0073] Core Function: Real-time detection of the potential fluctuation (ΔV RL4 ) caused by displacement current in the second shielding layer y2; amplifying the fluctuation signal and outputting it to the subsequent stage for noise cancellation; creating an active ground potential adjustment system to eliminate the phase shift of the equipotential input terminal GND itself.

[0074] 4. Main Signal Processing Channel of the Second Amplifier U1

[0075] Signal Input Layer:

[0076] a. Inverting Terminal Path, performing three-step signal processing:

[0077] The first resistor RL1 on the first copper core x1 → voltage drop correction of the cable's inherent impedance; connecting the second resistor R8 (249 Ω) on the first copper core x1 → setting the voltage attenuation ratio (cooperating with R10 = 330 Ω for amplitude stabilization); receiving the compensated voltage output by the first amplifier U12 (via R64 = 10 kΩ).

[0078] b. Non-Inverting Terminal Path:

[0079] The fourth resistor R65 (49.9 Ω) and the third resistor RL2 on the second copper core x2 form a reference input. Selecting 49.9 Ω (quasi-50 Ω) achieves high-frequency impedance matching and reduces standing wave reflection.

[0080] Calculation of the Amplification Parameter of the Second Amplifier U12:

[0081] Closed-loop gain A v = 1 + (R10) / (R8 + RL1), substituting the following parameters: R10 = 330 Ω, R8 = 249 Ω, assuming RL1 = 0.01 Ω (high-quality copper core), we get A v ≈ 1 + 330 / (249 + 0.01) ≈ 2.326 times (basic amplification ratio).

[0082] Cascaded Noise Cancellation:

[0083] The compensation signal output by the first amplifier U12 is injected into the inverting input terminal of the second amplifier U1 through the ninth resistor R64 (10 kΩ). The corresponding cancellation coefficient K = (R10 / R64) / (1 + R10 / R64) ≈ 0.032, thereby realizing the amplitude and phase adaptation correction.

[0084] 5. Output Interface Design

[0085] The eleventh resistor R6 (49.9 Ω) forms an optimal match with the 50 Ω input of the oscilloscope (SWR = 1.002) → reflection coefficient Γ = (50 - 49.9) / (50 + 49.9) ≈ 0.00099; isolate the capacitive load between the isolation op-amp output and the probe input; limit the short-circuit current to protect the front-end circuit.

[0086] Full-channel bandwidth calculation:

[0087] The dominant pole is determined by the gain-bandwidth product (GBP) of the second amplifier U1. Assuming that the second amplifier U1 selects ADA4817 (GBP = 400 MHz): the theoretical bandwidth BW of the second amplifier U1 = GBP / A v = 400 / 2.326 ≈ 172 MHz, and this bandwidth meets the requirements of most high-frequency tests (such as USB3.0, 5Gbps signal analysis).

[0088] III. Solutions to Key Technical Problems

[0089] Problem 1: Distributed ground loop interference

[0090] The adopted solution is:

[0091] a. Star-tree hybrid grounding: The first shielding layer y1 is grounded at a single point (GND1), and the second shielding layer y2 forms a dynamic potential balance through op-amp modulation.

[0092] b. Equipotential node absorption: The current components of the third resistor RL2 of the second copper core x2 and the fifth resistor RL3 of the first shielding layer are shunted by the equipotential input terminal GND. The ground potential difference is isolated through the sixth resistor R5 (1 kΩ) to reduce ground interference.

[0093] Simulation verification: Assuming that there is a 1 mV / 100 kHz interference between the ground terminal GND1 and the equipotential input terminal GND, if the CCMRR (common-mode rejection ratio) of the first amplifier U12 reaches 80 dB, a 0.01 μV residue will be generated at the output terminal. Then, after secondary suppression at the second amplifier U1 stage, the ground interference is almost undetectable.

[0094] Problem 2: Displacement current interference

[0095] Mechanism analysis: At high frequencies, the capacitive coupling effect (C Shield ) of the shield layer of the coaxial cable and the cable length generate displacement current: I dis = jω·C Shield ·V CM .

[0096] Suppression strategy:

[0097] 1. Active monitoring and compensation: The positive terminal of the first amplifier U12 monitors in real time the voltage drop V D = I dis ·RL4 across the seventh resistor RL4 on the second shield layer x2.

[0098] 2. Mirror injection: The output of the first amplifier U12 is injected into the inverting terminal of the second amplifier U1 through the ninth resistor R64, and vectorially subtracted from the main signal (the signal from the first copper core x1 and the second copper core x2).

[0099] 3. Physical isolation: The second shield layer y2 is not directly grounded, and the current is absorbed through the high-impedance input of the first amplifier U12.

[0100] Problem 3: Parasitic modulation

[0101] Use an ultra-low inductance sampling resistor (R2 ≤ 1Ω); select a 10 pF capacitor C1. The capacitor C1 introduces a controllable pole in the feedback loop formed with the eighth resistor R1 to suppress radio frequency interference; select the eleventh resistor R6 to be 49.9Ω to achieve transmission matching, eliminate signal bounce, and the transmission matching is adapted to the front-end impedance of 50Ω of the coaxial cable X.

[0102] Problem 4: Phase shift control

[0103] Mechanism and compensation: Phase shift mainly comes from the distributed parameters of the resistive and capacitive elements in the signal path, which is particularly significant at high frequencies. This circuit addresses it from three aspects:

[0104] 1. Symmetrical input path design:

[0105] Main signal path: The first resistor RL1 → the second resistor R8 → the inverting terminal of the second amplifier U1 (total impedance of 249.01Ω)

[0106] Compensation signal path: The seventh resistor RL4 → the first amplifier U12 → the ninth resistor R64 → the inverting terminal of the second amplifier U1 (total impedance = RL4 + R64)

[0107] In this application, path delay balance is achieved through precise resistor matching (R8 = 249Ω, RL4 ≈ the equivalent resistance of C1 and R1 + 10kΩ of R64).

[0108] 2. Active phase compensation network:

[0109] The capacitor C1 = 10 pF and the eighth resistor R1 = 1 kΩ are connected in parallel in the feedback loop of the first amplifier U12 to form a compensation pole.

[0110] Frequency calculation formula for the compensation pole: compensation frequency 。

[0111] Effect: Introduce phase boost at 15.9 MHz to offset the additional time delay caused by distributed capacitance (such as 2 pF / cm for PCB traces).

[0112] 3. Operational amplifier bandwidth matching (taking the operational amplifier AD8065 as an example):

[0113] The first amplifier U12 is configured as a voltage follower, and the bandwidth = gain - bandwidth product (GBW) / gain = 145 MHz / 1 = 145 MHz.

[0114] The second amplifier U1 is configured with a gain of approximately 2.3 times, and the equivalent bandwidth = 145 MHz / 2.3 ≈ 63 MHz. The phase shift superposition of the two - stage amplifier is verified by simulation to be controlled within ±0.1°, meeting the requirements of high - speed signals.

[0115] Problem 5: Realization of microvolt - level signal accuracy

[0116] The core is guaranteed by four - level technology:

[0117] 1. TCR optimization of the sampling resistor R2:

[0118] The sampling resistor R2 is selected to be made of manganin alloy (TCR ≤ 50 ppm / °C). Assuming ΔT = 10 °C, the error is only 0.05%, ensuring the minimum influence of temperature drift.

[0119] 2. Control of the input bias current of the operational amplifier:

[0120] The selection standard requires that the input bias current ≤ 1 nA. For example, the operational amplifier model ADA4528 - 1 is selected in this application;

[0121] Voltage error caused by the operational amplifier: , this error is 5 orders of magnitude lower than the 1 V voltage drop of a 1 Ω sampling resistor under 1 A current, and such a voltage error can be ignored.

[0122] Enhanced common - mode rejection:

[0123] The common - mode rejection ratio (CMRR) of the first amplifier U12 ≥ 100 dB; the CMRR of the second amplifier U1 with differential input ≥ 90 dB; the total CMRR is:

[0124]

[0125] Such an inhibition ratio can inhibit a common-mode noise of 1 V to below 0.45 μV, which has reached an extremely high level.

[0126] Thermal noise suppression design:

[0127] In this application, the total noise calculation formula: ; so the noise on the second resistor R8 is: , reaching level.

[0128] When a low-noise operational amplifier with an input noise of 1.9 nV / √Hz is selected, the low-frequency noise is: , also reaching level. Therefore, the total input noise on this circuit is approximately 8.7 μV_rms to a peak-to-peak value of about 57 μV. When the full-scale is 1 V, the signal-to-noise ratio SNR = 20log(1 V / 57 μV) ≈ 104 dB.

[0129] In this application, the circuit adopts a three-layer grounding mechanism to form a distributed noise absorption system:

[0130] a) Main ground plane GND1: Directly connected to the common ground, discharging high-frequency noise to a low-impedance path;

[0131] b) Equipotential input terminal GND: As the signal processing reference;

[0132] Equivalent circuit: A star network composed of the third resistor RL2 (second copper core), the fifth resistor RL3 (first shield layer), and the seventh resistor RL4 (second shield).

[0133] Potential stabilization mechanism: Monitored and adjusted in real time by the first amplifier U12

[0134] Second shield layer RL4: Actively regulated by the first operational amplifier, and at this time, the displacement current Idis can be guided to the virtual ground of the operational amplifier input to avoid contaminating the main signal ground.

[0135] For the first amplifier U12 in this application: Noise amplification and suppression are achieved through input differential pair detection. Among them, the positive terminal signal (reflecting the shield layer current); the negative terminal signal .

[0136] At this time, the closed-loop gain of the transfer function of the first amplifier U12 is set to 1 (due to of R1), achieving active subtraction of the shield layer noise.

[0137] In addition, for the first amplifier U12, when the capacitor C1 is introduced, the phase margin at the unity-gain crossover point of the open-loop gain curve reaches 65°, which can prevent oscillation.

[0138] For the second amplifier U1 of the present application, the summation current equation at the inverting input terminal of the second amplifier U1:

[0139] The voltage at the non-inverting input terminal of the second amplifier U1: , forms a voltage division with the third resistor RL2, and the third resistor RL2 ≈ R65. At this time, the voltage at the non-inverting input terminal of the second amplifier U1 is half of the voltage across the sampling resistor R2. Therefore, the output voltage is solved, so as to achieve vector cancellation of the noise components.

[0140] In the case of example calculation, assuming RL1 = 0.01Ω, R8 = 249Ω, R10 = 330Ω, R64 = 10kΩ, then the main signal path gain:

[0141] The compensation path gain:

[0142] The total output expression: When , partial cancellation of the shielding noise is achieved.

[0143] In the present application, the capacitor C1 has different functions in different frequency bands. Specifically:

[0144] In the low-frequency band (f < 1MHz): The impedance of C1 is much larger than that of R1 (1kΩ → 10pF, the capacitive reactance is ≈ 15.9kΩ at 1MHz), and the feedback is dominated by R1. The operational amplifier operates in the unity-gain mode to ensure DC stability.

[0145] In the mid-frequency band (1 - 15MHz): The capacitive reactance of C1 gradually decreases, and it begins to form a parallel effect with R1. At this time, phase-lead compensation is introduced to improve the transient response (reduce ringing);

[0146] In the high-frequency band (> 15MHz), C1 becomes the main feedback path, extending the bandwidth of U12 to the GBW limit, thereby suppressing the coupling of radio frequency interference (such as 900MHz mobile phone signal).

[0147] In the present application, the circuit component parameter selection is as follows in the table:

[0148]

[0149] Calculated according to the key ratio of the above table values:

[0150] Among them, the signal-to-noise ratio improvement coefficient: →After conversion, a 32 dB improvement is achieved. Among them, the ground loop isolation: →An -89 dB coupling attenuation is achieved.

[0151] The circuit in this application systematically solves the core interference problem in the detection of coaxial cable X through multi-stage feedback, common-mode compensation, and impedance matching technologies.

[0152] Its beneficial effects include:

[0153] 1. High common-mode rejection ratio (CMRR): The dual-op amp nested structure can theoretically achieve a CMRR of more than 120 dB.

[0154] 2. Wideband response: Extended to several hundred MHz through small capacitor compensation to adapt to high-speed signal sampling.

[0155] 3. Anti-ground loop design: The dynamic virtual ground technology eliminates the potential risk of multi-point grounding.

[0156] 4. Phase consistency guarantee: The symmetric layout and compensation network ensure the group delay.

[0157] This kind of circuit is particularly suitable for scenarios with strict requirements for signal integrity, such as high-frequency communication cables (such as 5G base station feeder cables) and medical ultrasound probes. By adjusting parameters (such as gain and bandwidth), it can be further adapted to different cable specifications.

[0158] Refer to Figure 1 and Figure 2 , the present invention also discloses a non-linear compensation method for a signal acquisition probe, which is used to detect high-frequency electrical signals on a coaxial cable X, and the method includes:

[0159] Step S1: Sample the current between the first copper core x1 and the second copper core x2 and convert it into a voltage;

[0160] Step S2: Sample the noise signal between the first copper core x1 and the first shielding layer y1;

[0161] Step S3: Set an equipotential input terminal between the second copper core x2 and the first shielding layer y1 to make the signals on the second copper core x2 and the first shielding layer y1 form a dynamic potential balance point, so as to reduce the influence of displacement current caused by multi-point grounding on the coaxial cable X;

[0162] Step S4: Differentially amplify the noise between the first shielding layer y1 and the second shielding layer y2, use differential amplification to eliminate common-mode interference, isolate the influence on the signal on the first copper core x1, and filter the differentially amplified noise;

[0163] Step S5: Superimpose the filtered noise signal on the signal on the first copper core x1, and after superimposition, amplify the signal on the second copper core x2 and output it to the detection end;

[0164] Among them, the filtered noise signal is used to perform phase compensation on the signal on the first copper core x1.

[0165] Among them, the second copper core x2, the first shielding layer y1, and the second shielding layer y2 form a connection loop to the noise differential amplification input end, thereby destroying the multi-point grounding structure of the coaxial cable X to avoid the formation of common-mode noise by distributed parameters.

[0166] Among them, during noise differential amplification, it also includes real-time monitoring of the potential fluctuations caused by the noise on the first shielding layer y1 and the second shielding layer y2, and adjusting the potential bias point according to the monitoring results when superimposing the filtered noise signal on the signal on the first copper core x1, maintaining the common-mode rejection ratio on the first shielding layer y1 and the second shielding layer y2 to achieve the purpose of suppressing noise.

[0167] Among them, when amplifying the signal on the second copper core x2, the superimposed signal of the filtered noise signal and the first copper core x1 is used as the reference input to eliminate the noise.

[0168] Finally, it should be noted that the above embodiments are only used to illustrate the technical solutions of the present invention, rather than limiting the protection scope of the present invention. Although the present invention has been described in detail with reference to the preferred embodiments, those of ordinary skill in the art should understand that the technical solutions of the present invention can be modified or equivalently replaced without departing from the essence and scope of the technical solutions of the present invention.

Claims

1. A non-linear compensation circuit for a signal acquisition probe, characterized in that The circuit includes: A detection terminal, which is electrically connected to a coaxial cable. The coaxial cable includes a first copper core and a second copper core. A first shielding layer is disposed around the first copper core, and a second shielding layer is disposed around the second copper core. The first copper core has a first resistance, the second copper core has a third resistance, the first shielding layer has a fifth resistance, and the second shielding layer has a seventh resistance; A sampling resistor, the first end of the sampling resistor is connected to the first copper core, and the second end of the sampling resistor is respectively connected to the second copper core, the first shielding layer, and the second shielding layer. The first shielding layer is grounded; Wherein, the first end of the sampling resistor is connected to the negative input terminal of the second amplifier through the first resistor and the second resistor in sequence; The second end of the sampling resistor is respectively connected to the equipotential input terminal through the third resistor and the fifth resistor. The third resistor and the fifth resistor are connected in parallel; The equipotential input terminal is connected to the negative input terminal of the first amplifier through the sixth resistor; the equipotential input terminal is connected to the positive input terminal of the second amplifier through the fourth resistor and connected to the negative input terminal of the first amplifier through the sixth resistor; The sampling resistor is connected to the positive input terminal of the first amplifier through the seventh resistor; the output terminal of the first amplifier is connected to the negative input terminal of the second amplifier through the ninth resistor; the output terminal of the first amplifier is connected to the negative input terminal of the first amplifier through the eighth resistor, and both ends of the eighth resistor are connected in parallel with a capacitor; the negative input terminal of the second amplifier is connected to the output terminal of the second amplifier through the tenth resistor; the output terminal of the second amplifier is connected to the probe output terminal through the eleventh resistor.

2. The non-linear compensation circuit for a signal acquisition probe according to claim 1, wherein The sampling resistor is a resistor made of alloy material to reduce the error influence caused by sampling temperature drift.

3. The non-linear compensation circuit for a signal acquisition probe as described in claim 1, characterized in that, The equipotential input terminal is used to form a redundant current path by the parallel-connected third resistor, fifth resistor, and seventh resistor and establish a common reference ground to discharge the high-frequency noise on the second copper core and the first shielding layer.

4. The non-linear compensation circuit for a signal acquisition probe according to claim 1, wherein, The negative input terminal of the first amplifier is connected to the equipotential input terminal to break the circulation loop on the first shielding layer and / or the second shielding layer of the coaxial cable, thereby reducing the circulation grounding interference.

5. The non-linear compensation circuit for a signal acquisition probe according to claim 1, wherein The positive input terminal of the first amplifier is connected to the second shielding layer to detect the potential fluctuation caused by the displacement current on the second shielding layer in real time through the seventh resistor. The first amplifier is used to amplify the potential fluctuation to the output terminal of the first amplifier, and use the filtering and feedback branch formed by the eighth resistor and the capacitor at the output terminal of the first amplifier to achieve noise cancellation and feedback gain maintenance, thereby reducing the noise influence at the front end of the first amplifier.

6. The non-linear compensation circuit for a signal acquisition probe according to claim 1, characterized in that, The sampling resistor is less than or equal to 1Ω, the resistance value of the second resistor is 249Ω, the fourth resistor is 49.9Ω, the sixth resistor is 1K, the eighth resistor is 1K, the ninth resistor is 10K, the tenth resistor is 330Ω, the eleventh resistor is 49.9Ω; the capacitor is 10pF.

7. A non - linear compensation method for a signal acquisition probe, characterized in that, The method is used to detect high-frequency electrical signals on a coaxial cable. The coaxial cable includes a first copper core and a second copper core. A first shielding layer is arranged around the first copper core, and a second shielding layer is arranged around the second copper core; The first copper core has a first resistance, the second copper core has a third resistance, the first shielding layer has a fifth resistance, and the second shielding layer has a seventh resistance; The method includes: Step S1: Sample the current between the first copper core and the second copper core and convert it into a voltage; Step S2: Sample the noise signal between the first copper core and the first shielding layer; Step S3: Set an equipotential input terminal between the second copper core and the first shielding layer to form a dynamic potential balance point between the signals on the second copper core and the first shielding layer, so as to reduce the influence of displacement current caused by multi-point grounding on the coaxial cable; Step S4: Differentially amplify the noise between the first shielding layer and the second shielding layer, use differential amplification to eliminate common-mode interference, isolate the influence on the signal on the first copper core, and filter the amplified noise; Step S5: Superimpose the filtered noise signal on the signal on the first copper core, and after superimposition, amplify the gain of the signal on the second copper core and output it to the detection end; Among them, the filtered noise signal is used to perform phase compensation on the signal on the first copper core.

8. The non-linear compensation method for a signal acquisition probe according to claim 7, characterized in that, The second copper core, the first shielding layer, and the second shielding layer form a connection loop to the noise differential amplification input terminal, thereby destroying the multi-point grounding structure of the coaxial cable to avoid the formation of common-mode noise by distributed parameters.

9. The non-linear compensation method for a signal acquisition probe according to claim 7, characterized in that During noise differential amplification, it also includes real-time monitoring of the potential fluctuations caused by the noise on the first shielding layer and the second shielding layer, and adjusting the potential bias point according to the monitoring results when superimposing the filtered noise signal on the signal on the first copper core, maintaining the common-mode rejection ratio on the first shielding layer and the second shielding layer, so as to achieve the purpose of suppressing noise.

10. The non-linear compensation method for a signal acquisition probe according to claim 7, characterized in that, When amplifying the gain of the signal on the second copper core, the superimposed signal of the filtered noise signal and the signal on the first copper core is used as the reference input to eliminate the noise.

Images