Multi-mode modulation control method suitable for low-carrier-ratio operation of permanent magnet synchronous motor

Abstract

The invention discloses a multi-mode modulation control method suitable for low-carrier-ratio operation of a permanent magnet synchronous motor, and belongs to a permanent magnet synchronous motor control technology. The method aims at solving the problems that control over the medium-high-power permanent magnet synchronous motor is limited by the switching frequency, and high-low-frequency harmonic waves appear in the output voltage and response current of an inverter. Comprising the steps of constructing a multi-mode modulation control mode; determining modulation methods of different rotating speed intervals; an optimized synchronous modulation method based on specific frequency harmonic elimination; a modulation switching method based on flux linkage path continuity is provided, stator flux linkage paths before and after switching operation are made to be continuous, an acting voltage vector is kept constant, the influence of modulation switching operation on system stability is reduced, and then full-speed-domain operation of the motor is completed. The method is used for low-carrier-ratio control of the permanent magnet synchronous motor.

Description

Technical Field

[0001] The invention relates to a multi-mode modulation control method suitable for low-carrier ratio operation of a permanent magnet synchronous motor, belonging to the permanent magnet synchronous motor control technology. Background Art

[0002] With the development of rare earth materials and control technology, high-efficiency and high-reliability permanent magnet synchronous motors have been widely used in industry, including medium and large power applications, such as traction motor control systems in high-speed trains, large wind turbine systems, and large submersible electric pump systems in flood control projects.

[0003] In medium and high power applications, switching devices need to withstand high voltage and high current. The increase in voltage and current means that the system has higher switching losses. Therefore, in order to ensure the life of the device and improve the output power efficiency of the drive inverter, the switching frequency of the inverter must be reduced in medium and high power applications to reduce the number of switching times of the switch tube state per unit time, thereby reducing switching losses.

[0004] However, although low switching frequency can effectively reduce switching losses, it also makes high-performance speed regulation of the motor very difficult. Low switching frequency leads to increased controller delay, limited control bandwidth, and the generation of low-order harmonics that are difficult to filter out. These problems make it impossible for the current loop in the control system to achieve fast and stable response performance, thus affecting the control performance of the entire system. The use of multi-mode modulation strategy during the full-speed operation of the motor can improve the inverter output voltage quality and current response performance under low carrier ratio conditions. Therefore, it is of great significance to study multi-mode modulation control methods in permanent magnet synchronous motor control. Summary of the invention

[0005] Aiming at the problem that the control of medium and large power permanent magnet synchronous motors is limited by switching frequency, and higher low-frequency harmonics appear in the inverter output voltage and response current, the present invention provides a multi-mode modulation control method suitable for low-carrier ratio operation of permanent magnet synchronous motors.

[0006] A multi-mode modulation control method suitable for low-carrier ratio operation of a permanent magnet synchronous motor of the present invention comprises:

[0007] Set the motor speed to differentiate the variables ω b , when the motor speed is less than ω b The traditional asynchronous SVPWM modulation method in the low speed area is used to control the inverter. When the motor speed is greater than or equal to ω b The inverter is controlled by using high-speed synchronous optimization modulation mode;

[0008] The traditional asynchronous SVPWM modulation method in the low-speed zone includes: according to the α-axis voltage U under the two-phase stationary shaft system αand β-axis voltage U β Determine the switching state of the switching device of the two-level inverter, use the switching state of the switching device of the two-level inverter to obtain six effective voltage vectors U1~U6 and two zero voltage vectors U0 and U7, and divide the voltage space vector into six sectors according to the distribution of the effective voltage vectors; then, based on the sector position of the given voltage vector in the voltage space vector, perform action time allocation on the two boundary effective voltage vectors of the target sector, and perform switching control on the corresponding switching devices of the two-level inverter according to the action time allocation result;

[0009] The high-speed area synchronous optimization modulation method includes: performing Fourier series expansion on the output voltage of the two-level inverter to obtain the output voltage Fourier series expansion; then according to the fact that the synchronous modulation method has 1 / 4 wave axis symmetry and half wave center symmetry, and the output voltage waveform function of the two-level inverter is an odd harmonic function, a simplified form of the output voltage Fourier series expansion is obtained, and the switching state switching angle of the two-level inverter switch device within one electrical cycle is obtained by solving, and the corresponding switch device of the two-level inverter is switched according to the switching state switching angle;

[0010] The switching control of the two modulation modes includes: distinguishing the variable ω at the motor speed based on the two modulation modes b The flux path under the two modulation modes is determined, and the overlapping path of the flux path under the two modulation modes is determined. The time point when the same effective voltage vector acts in the overlapping path is used as the switching control moment under the condition of flux path continuity, and the switch control of the corresponding modulation mode is performed.

[0011] According to the multi-mode modulation control method for low carrier ratio operation of permanent magnet synchronous motor of the present invention, the motor speed is distinguished by the variable ω b The method for determining is:

[0012] According to the system switching frequency f sw and the actual electrical frequency f when the system is running op Calculate the carrier ratio k z , the carrier ratio k z The corresponding motor speed when it is lower than the set critical value is used as the motor speed distinction variable ω b :

[0013]

[0014] Where p n is the pole pair number.

[0015] According to the multi-mode modulation control method applicable to low carrier ratio operation of a permanent magnet synchronous motor of the present invention, the method for allocating the action time of two boundary effective voltage vectors of the target sector is as follows:

[0016]

[0017] Where T b1 , T b2 are the effective voltage vector action time of the two boundaries of the target sector, m is the modulation ratio, T s is the electrical period, α is the given voltage vector phase, T z is the total action time of the two zero voltage vectors.

[0018] According to the multi-mode modulation control method for low-carrier ratio operation of a permanent magnet synchronous motor of the present invention, the output voltage of the two-level inverter is expanded by Fourier series, and the output voltage Fourier series expansion is obtained as follows:

[0019]

[0020] Where U0 is the output voltage of the two-level inverter, t is the time, n is the harmonic number, and a n is the proportion coefficient of the nth harmonic sinusoidal component, b n is the proportion coefficient of the nth harmonic cosine component, and ω is the voltage fundamental angular velocity.

[0021] According to the multi-mode modulation control method for low carrier ratio operation of permanent magnet synchronous motor of the present invention, according to the synchronous modulation mode having 1 / 4 wave axis symmetry and half wave center symmetry, b is obtained. n =0;

[0022] According to the output voltage waveform function of the two-level inverter is an odd harmonic function, we can get a n There are only odd harmonic components in ;

[0023] The simplified form of formula (3) is obtained:

[0024]

[0025] Where U dc is the bus voltage, d is the number of switching states in a quarter of an electrical cycle, α k is the switch state switching angle at the kth switching time;

[0026] According to the fact that the output voltage waveform of the two-level inverter does not contain even harmonic components, and the three-phase symmetrical load does not contain triple frequency harmonics, it is determined to eliminate the 5th, 7th, 11th, 13th... harmonic components, so that the corresponding a5, a7, a 11 、a 13 …is zero, and the switch state switching angle α corresponding to the kth switching time is obtained. k .

[0027] According to the multi-mode modulation control method for low carrier ratio operation of a permanent magnet synchronous motor of the present invention, the electromagnetic torque is calculated as:

[0028]

[0029] Where T e is the electromagnetic torque, ψ r is the rotor flux; ψ s is the stator flux, L s is the stator inductance.

[0030] According to formula (5), the overlapping path of the flux paths under the two modulation modes is selected, and then the switching control time is finally determined according to the pulse switching sequence of the actual switch control under the two modulation modes.

[0031] According to the multi-mode modulation control method for low carrier ratio operation of permanent magnet synchronous motor of the present invention, the α-axis voltage U α and β-axis voltage U β The methods of obtaining include:

[0032] According to the permanent magnet synchronous motor rotor position angle θ e Get the motor speed ω e , set the motor speed to a given value ω e_ref and motor speed ω e The comparison result is obtained by the speed controller to obtain the given current i of the q axis of the two-phase synchronous rotating shaft system q_ref , set the q-axis of the two-phase synchronous rotating shaft system to a given current i q_ref The q-axis current i of the two-phase synchronous rotating shaft system q The comparison result is obtained by PI operation to obtain the α-axis voltage U under the two-phase stationary shaft system α ; Set the d-axis of the two-phase synchronous rotating shaft system to a given current i d_ref The d-axis current i of the two-phase synchronous rotating shaft system d The comparison result is calculated by PI to obtain the β-axis voltage U under the two-phase stationary shaft system. β .

[0033] According to the multi-mode modulation control method for low-carrier ratio operation of a permanent magnet synchronous motor of the present invention, in the high-speed area synchronization optimization modulation mode, the output voltage calculation method of the two-level inverter is:

[0034] According to the permanent magnet synchronous motor rotor position angle θ e Get the motor speed ω e , based on the motor speed ω e 、α-axis voltage U under two-phase stationary shaft system α And the β-axis voltage U under the two-phase stationary shaft system β The output voltage of the two-level inverter is calculated.

[0035] According to the multi-mode modulation control method for low carrier ratio operation of a permanent magnet synchronous motor of the present invention, the q-axis current i of the two-phase synchronous rotating shaft system is q and the d-axis current i of the two-phase synchronous rotating shaft system d The calculation method is as follows: collect the two-phase output current of the two-level inverter, reconstruct the complete three-phase output current, and then combine it with the permanent magnet synchronous motor rotor position angle θ e The q-axis current i of the two-phase synchronous rotating shaft system is calculated q and the d-axis current i of the two-phase synchronous rotating shaft system d .

[0036] According to the multi-mode modulation control method suitable for low carrier ratio operation of permanent magnet synchronous motor of the present invention, the carrier ratio k z The threshold value is set to 10.

[0037] Beneficial effects of the present invention: The present invention constructs a multi-mode modulation method, adopts asynchronous modulation in the low-speed area, and adopts optimized synchronous modulation in the high-speed area, so as to improve the inverter output voltage quality and reduce the harmonic content of the system response current. At the same time, a modulation mode switching method with continuous flux path is adopted to reduce the impact of switching operation on the system, which can improve the operating performance of the permanent magnet synchronous motor under low carrier ratio conditions.

[0038] The method of the present invention takes into account the low carrier ratio problem under the switching frequency limitation, constructs a multi-mode modulation method, and adopts a synchronous modulation method in the high speed area to ensure the symmetrical characteristics of the system response current and improve the system operation performance.

[0039] The method of the present invention takes into account the problem of voltage and current harmonic distortion under low carrier ratio conditions, and proposes a synchronous modulation method for eliminating specific frequency harmonics. By performing Fourier series expansion on the inverter output voltage, a multivariate equation group is written for the target frequency harmonics, and the state switching angle of the switching device is solved to suppress the specific target frequency harmonics.

[0040] The method of the present invention designs the switching points of different modulation modes based on the principle of flux path continuity, ensures the continuity of the stator flux and the applied voltage vector before and after the switching operation, reduces the torque fluctuation caused by the modulation switching, and improves the system operation stability. BRIEF DESCRIPTION OF THE DRAWINGS

[0041] Figure 1 is a control block diagram of the multi-mode modulation control method for low carrier ratio operation of a permanent magnet synchronous motor according to the present invention; dq is the dq current vector of the two-phase synchronous rotating shaft system, the pulse mode a is the switch device control pulse output by the traditional asynchronous SVPWM modulation mode in the low-speed area, and the pulse mode b is the switch device control pulse output by the synchronous optimization modulation mode in the high-speed area;

[0042] Figure 2 This is a simulation diagram of asynchronous SVPWM modulation operation in the traditional asynchronous SVPWM modulation mode in the low-speed area. a_low is the A phase running current under the condition of carrier ratio of 7, h5 and h7 are the current I a_low The 5th and 7th harmonic content of the Fourier analysis graph;

[0043] Figure 3 This is a simulation diagram of optimized synchronous modulation operation in the high-speed area synchronous optimization modulation mode. a_high is the A phase running current under the condition of carrier ratio of 7, h 11 、h 13 is the current I a_high The 11th and 13th harmonic content of the Fourier analysis graph;

[0044] Figure 4 The flux trajectory diagram of the conventional asynchronous SVPWM modulation mode in the low speed zone and the synchronous optimization modulation mode in the high speed zone at a speed of 1000 rpm, respectively, wherein the abscissa is the α-axis component of the stator flux and the ordinate is the β-axis component of the stator flux;

[0045] Figure 5 The control pulse sequence diagram of the conventional asynchronous SVPWM modulation method in the low speed zone of the present invention at a speed of 1000 rpm, in which T 1 / 6 , T 1 / 36 They are 1 / 6 and 1 / 36 electrical cycle sequences respectively. Roman numerals Ⅰ, Ⅱ, Ⅲ are sector numbers, indicating the space voltage vector Ⅰ, Ⅱ, Ⅲ sectors; S a_low is the a-phase pulse of the asynchronous SVPWM modulation mode, S b_low is the b-phase pulse of the asynchronous SVPWM modulation mode, S c_low It is the c-phase pulse of asynchronous SVPWM modulation mode;

[0046] Figure 6 The pulse sequence diagram of the high-speed synchronous optimization modulation mode at a speed of 1000 rpm in the method of the present invention, in which S a_high To optimize the a-phase pulse of the synchronous modulation method, S b_high To optimize the b-phase pulse of the synchronous modulation method, S c_high To optimize the c-phase pulse of the synchronous modulation method;

[0047] Figure 7 The simulation waveform diagram of the modulation mode switching using the discontinuous flux path method; S a_low is the a-phase pulse of the asynchronous SVPWM modulation mode, S a_high In order to optimize the a-phase pulse of the synchronous modulation mode, two switching operations are performed at the modulation switching point, T e is the electromagnetic torque, Ia is the A phase current;

[0048] Figure 8 The figure is a simulation waveform diagram of modulation mode switching control using the method of the present invention. DETAILED DESCRIPTION

[0049] The following will be combined with the drawings in the embodiments of the present invention to clearly and completely describe the technical solutions in the embodiments of the present invention. Obviously, the described embodiments are only part of the embodiments of the present invention, not all of the embodiments. Based on the embodiments of the present invention, all other embodiments obtained by ordinary technicians in this field without creative work are within the scope of protection of the present invention.

[0050] It should be noted that, in the absence of conflict, the embodiments of the present invention and the features in the embodiments may be combined with each other.

[0051] The present invention will be further described below in conjunction with the accompanying drawings, but is not intended to be a limitation of the present invention.

[0052] Specific implementation method 1. Combination Figure 1 As shown, the present invention provides a multi-mode modulation control method suitable for low carrier ratio operation of a permanent magnet synchronous motor, comprising:

[0053] Set the motor speed to differentiate the variables ω b , when the motor speed is less than ω b The traditional asynchronous SVPWM modulation method in the low speed area is used to control the inverter. When the motor speed is greater than or equal to ω b The inverter is controlled by using high-speed synchronous optimization modulation mode;

[0054] The traditional asynchronous SVPWM modulation method in the low-speed zone includes: according to the α-axis voltage U under the two-phase stationary shaft system α and β-axis voltage U β Determine the switching state of the switching device of the two-level inverter, use the switching state of the switching device of the two-level inverter to obtain six effective voltage vectors U1~U6 and two zero voltage vectors U0 and U7, and divide the voltage space vector into six sectors according to the distribution of the effective voltage vectors; then, based on the sector position of the given voltage vector in the voltage space vector, perform action time allocation on the two boundary effective voltage vectors of the target sector, and perform switching control on the corresponding switching devices of the two-level inverter according to the action time allocation result;

[0055] The high-speed area synchronous optimization modulation method includes: performing Fourier series expansion on the output voltage of the two-level inverter to obtain the output voltage Fourier series expansion; then according to the fact that the synchronous modulation method has 1 / 4 wave axis symmetry and half wave center symmetry, and the output voltage waveform function of the two-level inverter is an odd harmonic function, a simplified form of the output voltage Fourier series expansion is obtained, and the switching state switching angle of the two-level inverter switch device within one electrical cycle is obtained by solving, and the corresponding switch device of the two-level inverter is switched according to the switching state switching angle;

[0056] The switching control of the two modulation modes includes: distinguishing the variable ω at the motor speed based on the two modulation modes b The flux path under the two modulation modes is determined, and the overlapping path of the flux path under the two modulation modes is determined. The time point when the same effective voltage vector acts in the overlapping path is used as the switching control moment under the condition of flux path continuity, and the switch control of the corresponding modulation mode is performed.

[0057] This implementation method aims at the low carrier ratio problem caused by switching frequency limitation, and constructs a multi-mode modulation control method. With the motor speed as the distinguishing variable, the full-speed domain process of the permanent magnet synchronous motor is divided into different modulation modes. The traditional asynchronous SVPWM modulation method is used at low speed, and the optimized synchronous modulation method is used at high speed, so as to improve the inverter output voltage and system response current quality and enhance the system operation stability. The traditional asynchronous SVPWM modulation method in the low-speed area finally obtains a rotating space voltage vector; the optimized synchronous modulation method in the high-speed area constructs a multivariate equation group based on the elimination of specific frequency harmonics and the Fourier series expansion formula, and directly calculates the state switching angle of the inverter switch device by solving the equation group, and then obtains all the switching angles within one electrical cycle according to the 1 / 4 wave axis symmetry and half-wave center symmetry principles, and finally outputs a pulse signal according to the switching angle.

[0058] During the switching process between different modulation methods, a modulation switching method based on continuous flux path is adopted to reduce the impact of switching operation on system operation.

[0059] Combination Figure 1 As shown in the figure, there are two modulation modes in the system operation process. The asynchronous SVPWM modulation mode uses the output of the current loop controller as the input variable to generate a pulse sequence of pulse mode a; the optimized synchronous modulation mode uses the actual motor speed ω e and the two-phase synchronous rotating shaft system dq current vector i dq As the input variable, a pulse sequence of pulse mode b is generated. According to the actual speed of the system, a multi-mode modulation switching strategy with continuous flux path is designed to select and switch between the two pulse modes, and finally achieve efficient operation of the system.

[0060] Furthermore, the motor speed is differentiated by the variable ωb The method to determine is:

[0061] According to the system switching frequency f sw and the actual electrical frequency f when the system is running op Calculate the carrier ratio k z , the carrier ratio k z The corresponding motor speed when it is lower than the set critical value is used as the motor speed distinction variable ω b :

[0062]

[0063] Where p n is the pole pair number.

[0064] In this embodiment, the multi-mode control method is a control method for coping with the change of carrier ratio at different speeds. The carrier ratio value is calculated by formula (1). Usually, the system switching frequency f sw Fixed, the speed increases during system operation, causing the carrier ratio to decrease. By setting the critical value of the carrier ratio, the motor speed differentiation variable ω can be calculated. b .

[0065] The motor speed is less than ω b The PWM waveform of the asynchronous modulation output is fine, which can ensure the symmetry of the positive and negative half-cycle pulses, and adopt the traditional SVPWM modulation method. ref Located in the sector position of the voltage space vector, it controls the conduction state and conduction time of the switch tube, for example, using the effective voltage vectors U1 and U2 to synthesize the required voltage vector U ref Repeat the above operation to realize the voltage space vector moving along the polygonal trajectory in the steady state. In the implementation process, first determine the sector position to which the voltage command vector belongs, and then determine the action time of the effective voltage vector and the zero voltage vector. According to the volt-second balance principle, the action time of different voltage vectors can be obtained.

[0066] In this implementation manner, the method for allocating the action time of two boundary effective voltage vectors of the target sector is as follows:

[0067]

[0068] Where T b1 、T b2 are the effective voltage vector action time of the two boundaries of the target sector, m is the modulation ratio, T s is the electrical period, α is the given voltage vector phase, T z is the total action time of the two zero voltage vectors.

[0069] Combination Figure 2 and Figure 3As shown in the figure, in the low speed area, the system operating frequency is low and the carrier ratio can be maintained at a high level. However, as the system operating frequency continues to increase, the carrier ratio continues to decrease due to the switching frequency limitation, resulting in serious distortion of the system response current. Figure 2 , I a_low is the A-phase current under the condition of carrier ratio of 7. Fourier analysis of the A-phase current shows that its harmonic content is complex, and there are a large number of stray harmonics and low-frequency harmonics, such as the 5th and 7th harmonics corresponding to h5 and h7, which cause low-frequency torque fluctuations in the system and affect the stability of system operation.

[0070] Furthermore, the motor speed is higher than ω b In view of the complex content of harmonic components, an optimized synchronous modulation method is used in the high-speed area to improve the output voltage quality and enhance the motor control performance. The synchronous modulation method has the characteristics of 1 / 4 wave axis symmetry and half wave center symmetry. The output voltage of the two-level inverter is expanded by Fourier series, and the output voltage Fourier series expansion is obtained as follows:

[0071]

[0072] Where U0 is the output voltage of the two-level inverter, which is an arbitrary periodic function; t is time, n is the harmonic number, and a n is the proportion coefficient of the nth harmonic sinusoidal component, b n is the proportion coefficient of the nth harmonic cosine component, and ω is the voltage fundamental angular velocity.

[0073] According to the synchronous modulation mode, the 1 / 4 wave axis symmetry and the half wave center symmetry should not exist in the Fourier series expansion expression, so b n =0;

[0074] According to the fact that the output voltage waveform function of the two-level inverter is an odd harmonic function, it is proved that its Fourier series expansion only contains odd harmonic components of sine and cosine terms, and we get a n There are only odd harmonic components in ;

[0075] Based on the above analysis of the inverter output voltage waveform, the simplified form of formula (3) is obtained:

[0076]

[0077] Where U dc is the bus voltage, d is the number of switching states in a quarter of an electrical cycle, α k is the switch state switching angle at the kth switching time;

[0078] The proportion of different harmonic components under the switching angle of the switch state at different times is obtained by formula (4). According to the characteristics of the output voltage waveform of the two-level inverter, the waveform does not contain even harmonic components, and the load is a three-phase symmetrical load, and the waveform does not contain triple frequency harmonics. Therefore, the harmonic elimination should be mainly to eliminate the 5th, 7th, 11th, 13th... harmonic components. The method to eliminate the 5th, 7th, 11th, 13th... harmonic components is to make the corresponding a5, a7, a 11 、a 13 …is zero, and the switch state switching angle α corresponding to the kth switching time is obtained. k .

[0079] Figure 2 As shown in , the 5th and 7th harmonics should be eliminated to make the corresponding a5 and a7 zero. At this time, the target switching angle can be obtained by solving the obtained multivariate equations. Figure 3 ,contrast Figure 2 It can be seen that the current distortion problem is improved under the same carrier ratio, and the 5th and 7th harmonic components in the current are effectively suppressed.

[0080] Going further, calculate the electromagnetic torque:

[0081]

[0082] Where T e is the electromagnetic torque, ψ r is the rotor flux; ψ s is the stator flux, L s is the stator inductance.

[0083] According to formula (5), the overlapping path of the flux paths under the two modulation modes is selected, and then the switching control time is finally determined according to the pulse switching sequence of the actual switch control under the two modulation modes.

[0084] Switching is performed at the point where the flux paths of the two modulation methods are continuous. According to formula (5), since the flux remains unchanged before and after switching, the impact of the switching operation on the system operation can be reduced. When selecting the switching point, the similar sectors of the flux trajectory are first selected to find the overlapping sections in the two flux paths; then the switching point is finally determined according to the actual pulse switching sequence of the two modulation methods.

[0085] See also Figure 4 It can be seen that the stator flux paths of the asynchronous modulation mode and the optimized synchronous modulation mode overlap in some areas, such as point A in the figure; on the contrary, the flux paths of the two modulation modes are inconsistent at point B. Therefore, in order to minimize the torque pulsation in the switching operation, the switching should be performed in the point A area. It can be seen from the figure that point A is located in sector II of the space voltage vector, so the switching operation should be performed in sector II.

[0086] When switching the modulation mode, it is necessary to ensure that the voltage vector applied at the switching moment remains consistent as much as possible while considering the continuity of the magnetic flux path. Figure 5 This is the pulse sequence diagram of asynchronous modulation in the low-speed area. From the above analysis, it can be seen that the switching operation is carried out in sector II. The switching sequence of the voltage vector in this sector is: 111→011→010→000→010→011→111→011→001→000→001→011; Figure 6 The pulse sequence diagram of the optimized synchronous modulation method in the high-speed zone is shown in the figure. It can be seen from the figure that the switching order of the voltage vector in sector II under the optimized synchronous modulation is: 010→011→111→011→111→011→001. In the range of 20° to 30° in sector II, both modulation methods use the voltage vector 011 for a long period of time. This shows that under this speed condition, the stator flux path and the applied voltage vector are completely consistent in this stage. Therefore, setting the switching point at the midpoint of the range of 20° to 30° in zone II can achieve the continuity of the voltage vector phase and the stator flux path before and after the modulation algorithm is switched.

[0087] Figure 7 This is the simulation waveform diagram of switching at point B. It can be seen from the figure that the electromagnetic torque pulsates after the switching operation, affecting the stability of the system. Figure 8 This is a simulation waveform diagram of switching the modulation mode using the method of the present invention. It can be seen from the figure that after the switching operation, the electromagnetic torque and phase current transition smoothly, verifying the effectiveness of the continuous switching scheme of the flux path.

[0088] Combination Figure 1 As shown, the α-axis voltage U under the two-phase stationary shaft system α and β-axis voltage U β The methods of obtaining include:

[0089] According to the permanent magnet synchronous motor rotor position angle θ e Get the motor speed ω e , set the motor speed to a given value ω e_ref and motor speed ω e The comparison result is obtained by the speed controller to obtain the given current i of the q axis of the two-phase synchronous rotating shaft system q_ref , set the q-axis of the two-phase synchronous rotating shaft system to a given current i q_ref The q-axis current i of the two-phase synchronous rotating shaft system q The comparison result is obtained by PI operation to obtain the α-axis voltage U under the two-phase stationary shaft system α ; Set the d-axis of the two-phase synchronous rotating shaft system to a given current i d_ref The d-axis current i of the two-phase synchronous rotating shaft system d The comparison result is calculated by PI to obtain the β-axis voltage U under the two-phase stationary shaft system. β .

[0090] Combination Figure 1 As shown in the figure, in the high-speed synchronous optimization modulation mode, the output voltage calculation method of the two-level inverter is:

[0091] According to the permanent magnet synchronous motor rotor position angle θ e Get the motor speed ω e , based on the motor speed ω e 、α-axis voltage U under two-phase stationary shaft system α And the β-axis voltage U under the two-phase stationary shaft system β The output voltage of the two-level inverter is calculated.

[0092] Two-phase synchronous rotating shaft system q-axis current i q and the d-axis current i of the two-phase synchronous rotating shaft system d The calculation method is as follows: collect the two-phase output current of the two-level inverter, reconstruct the complete three-phase output current, and then combine it with the permanent magnet synchronous motor rotor position angle θ e The q-axis current i of the two-phase synchronous rotating shaft system is calculated q and the d-axis current i of the two-phase synchronous rotating shaft system d .

[0093] As an example, the carrier ratio k z The threshold value is set to 10.

[0094] In summary, this embodiment first determines the boundary speed according to the carrier ratio, constructs a multi-mode modulation method, and then reduces the impact of the switching operation on the system by continuously switching the flux path, and finally realizes the low carrier ratio operation of the motor system.

[0095] Although the present invention is described herein with reference to specific embodiments, it should be understood that these embodiments are merely examples of the principles and applications of the present invention. It should therefore be understood that many modifications may be made to the exemplary embodiments and that other arrangements may be devised without departing from the spirit and scope of the present invention as defined by the appended claims. It should be understood that the various dependent claims and features described herein may be combined in a manner different from that described in the original claims. It should also be understood that the features described in conjunction with a single embodiment may be used in other described embodiments.

Claims

1. A multi-mode modulation control method suitable for low carrier ratio operation of a permanent magnet synchronous motor, characterized in that: include: Set the motor speed to differentiate the variables ω b , when the motor speed is less than ω b The traditional asynchronous SVPWM modulation method in the low speed area is used to control the inverter. When the motor speed is greater than or equal to ω b The inverter is controlled by using high-speed synchronous optimization modulation mode; The traditional asynchronous SVPWM modulation method in the low-speed zone includes: according to the α-axis voltage U under the two-phase stationary shaft system α and β-axis voltage U β Determine the switching state of the switching device of the two-level inverter, use the switching state of the switching device of the two-level inverter to obtain six effective voltage vectors U1~U6 and two zero voltage vectors U0 and U7, and divide the voltage space vector into six sectors according to the distribution of the effective voltage vectors; then, based on the sector position of the given voltage vector in the voltage space vector, perform action time allocation on the two boundary effective voltage vectors of the target sector, and perform switching control on the corresponding switching devices of the two-level inverter according to the action time allocation result; The high-speed area synchronous optimization modulation method includes: performing Fourier series expansion on the output voltage of the two-level inverter to obtain the output voltage Fourier series expansion; then according to the fact that the synchronous modulation method has 1 / 4 wave axis symmetry and half wave center symmetry, and the output voltage waveform function of the two-level inverter is an odd harmonic function, a simplified form of the output voltage Fourier series expansion is obtained, and the switching state switching angle of the two-level inverter switch device within one electrical cycle is obtained by solving, and the corresponding switch device of the two-level inverter is switched according to the switching state switching angle; The switching control of the two modulation modes includes: distinguishing the variable ω at the motor speed based on the two modulation modes b The flux path under the two modulation modes is determined, and the overlapping path of the flux path under the two modulation modes is determined. The time point when the same effective voltage vector acts in the overlapping path is used as the switching control moment under the condition of flux path continuity, and the switch control of the corresponding modulation mode is performed.

2. The multi-mode modulation control method suitable for low carrier ratio operation of a permanent magnet synchronous motor according to claim 1, characterized in that: Motor speed differentiation variable ω b The method for determining is: According to the system switching frequency f sw and the actual electrical frequency f when the system is running op Calculate the carrier ratio k z , the carrier ratio k z The corresponding motor speed when it is lower than the set critical value is used as the motor speed distinction variable ω b : Where p n is the pole pair number.

3. The multi-mode modulation control method suitable for low carrier ratio operation of a permanent magnet synchronous motor according to claim 2, characterized in that: The method for allocating the action time of the two boundary effective voltage vectors of the target sector is: Where T b1 , T b2 are the effective voltage vector action time of the two boundaries of the target sector, m is the modulation ratio, T s is the electrical period, α is the given voltage vector phase, T z is the total action time of the two zero voltage vectors.

4. The multi-mode modulation control method suitable for low carrier ratio operation of a permanent magnet synchronous motor according to claim 3, characterized in that: The output voltage of the two-level inverter is expanded by Fourier series, and the output voltage Fourier series expansion is obtained as follows: Where U0 is the output voltage of the two-level inverter, t is the time, n is the harmonic number, and a n is the proportion coefficient of the nth harmonic sinusoidal component, b n is the proportion coefficient of the nth harmonic cosine component, and ω is the voltage fundamental angular velocity.

5. The multi-mode modulation control method suitable for low-carrier ratio operation of a permanent magnet synchronous motor according to claim 4, characterized in that: According to the synchronous modulation method, which has 1 / 4 wave axis symmetry and half wave center symmetry, we can get b n =0; According to the output voltage waveform function of the two-level inverter is an odd harmonic function, we can get a n There are only odd harmonic components in ; The simplified form of formula (3) is obtained: Where U dc is the bus voltage, d is the number of switching states in a quarter of an electrical cycle, α k is the switch state switching angle at the kth switching time; According to the fact that the output voltage waveform of the two-level inverter does not contain even harmonic components, and the three-phase symmetrical load does not contain triple frequency harmonics, it is determined to eliminate the 5th, 7th, 11th, 13th... harmonic components, so that the corresponding a5, a7, a 11 、a 13 …is zero, and the switch state switching angle α corresponding to the kth switching time is obtained. k .

6. The multi-mode modulation control method suitable for low-carrier ratio operation of a permanent magnet synchronous motor according to claim 5, characterized in that: Calculate the electromagnetic torque: Where T e is the electromagnetic torque, ψ r is the rotor flux; ψ s is the stator flux, L s is the stator inductance. According to formula (5), the overlapping path of the flux paths under the two modulation modes is selected, and then the switching control time is finally determined according to the pulse switching sequence of the actual switch control under the two modulation modes.

7. The multi-mode modulation control method suitable for low carrier ratio operation of a permanent magnet synchronous motor according to claim 1, characterized in that: α-axis voltage U under two-phase stationary shaft system α and β-axis voltage U β The methods of obtaining include: According to the permanent magnet synchronous motor rotor position angle θ e Get the motor speed ω e , set the motor speed to a given value ω e_ref and motor speed ω e The comparison result is obtained by the speed controller to obtain the given current i of the q axis of the two-phase synchronous rotating shaft system q_ref , set the q-axis of the two-phase synchronous rotating shaft system to a given current i q_ref The q-axis current i of the two-phase synchronous rotating shaft system q The comparison result is obtained by PI operation to obtain the α-axis voltage U under the two-phase stationary shaft system α ; Set the d-axis of the two-phase synchronous rotating shaft system to a given current i d_ref The d-axis current i of the two-phase synchronous rotating shaft system d The comparison result is calculated by PI to obtain the β-axis voltage U under the two-phase stationary shaft system. β .

8. The multi-mode modulation control method suitable for low-carrier ratio operation of a permanent magnet synchronous motor according to claim 7, characterized in that: In the high-speed synchronous optimization modulation mode, the output voltage calculation method of the two-level inverter is: According to the permanent magnet synchronous motor rotor position angle θ e Get the motor speed ω e , based on the motor speed ω e 、α-axis voltage U under two-phase stationary shaft system α And the β-axis voltage U under the two-phase stationary shaft system β The output voltage of the two-level inverter is calculated.

9. The multi-mode modulation control method suitable for low carrier ratio operation of a permanent magnet synchronous motor according to claim 7, characterized in that: Two-phase synchronous rotating shaft system q-axis current i q and the d-axis current i of the two-phase synchronous rotating shaft system d The calculation method is as follows: collect the two-phase output current of the two-level inverter, reconstruct the complete three-phase output current, and then combine it with the permanent magnet synchronous motor rotor position angle θ e The q-axis current i of the two-phase synchronous rotating shaft system is calculated q and the d-axis current i of the two-phase synchronous rotating shaft system d .

10. The multi-mode modulation control method suitable for low-carrier ratio operation of a permanent magnet synchronous motor according to claim 2, characterized in that: Carrier ratio k z The threshold value is set to 10.

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