A control and testing method for a three-phase current-source rectifier

Abstract

This invention relates to a control and testing method for a three-phase current-source rectifier, comprising: obtaining the zero-crossing point of the three-phase input AC voltage using a software phase-locked loop method; initiating outer-loop control calculation at the zero-crossing point to stabilize the output voltage at the target value, with the output value of the outer-loop control serving as the fine-tuning value for the reference in the inner-loop PI closed-loop control; initiating the inner-loop PI closed-loop control, employing a direct current control method to achieve unity power factor; obtaining the αβ-axis space vector control variable through coordinate transformation of the output value of the inner-loop PI closed-loop control, and then obtaining a seven-segment PWM pulse through sector division and switching time calculation, which drives the MOSFETs in the main circuit; and performing detection by a host computer to obtain the change trajectory of key parameters and visualize the data. This invention not only achieves controllable output voltage of the current-source rectifier and simplifies the main circuit, but also improves the correctness of the space vector control algorithm and the reliability of rectifier operation through anti-saturation processing of the αβ-axis parameters in the stationary two-phase coordinate system.

Description

Technical Field

[0001] This invention relates to the field of high-power switching power supply technology, specifically to a control and testing method for a three-phase current-source rectifier. Background Art

[0002] Compared to three-phase voltage-source PWM rectifiers, three-phase current-source rectifiers do not have output short-circuit fault protection and have simpler input current control. However, due to the addition of a DC-side inductor, they are heavier and larger. For buck-type current-source rectifiers, using a boost-type voltage-source rectifier requires adding a DC / DC step-down stage. Clearly, current-source rectifiers are more suitable and efficient in low-voltage power supply applications. Furthermore, current-source rectifiers offer advantages such as controllable DC current, higher safety during short circuits, and ease of parallel connection to increase capacity.

[0003] Compared to three-phase voltage-source PWM rectifiers, three-phase current-source rectifiers involve additional binary / ternary logic conversion, zero-state discrimination, and more complex space vector modulation implementation. Furthermore, outer-loop and inner-loop control present design challenges. For digital power supplies, visualization of the processing of key control parameters at any given time significantly improves product design efficiency, which is also a design challenge.

[0004] Currently, the outer loop control of a three-phase current-source rectifier is used to stabilize the output current, while the output voltage remains uncontrollable. Therefore, a DC / DC converter stage needs to be added at the rectifier output to stabilize the output voltage. Furthermore, because the three-phase current-source rectifier uses Io when calculating the switching vector action time... d * c The target output current value is calculated as the command vector magnitude, without any control over the target output voltage. This increases design complexity in most applications requiring regulated output. Currently, host computer systems for digital power supply testing can display data and historical curves. However, due to inherent delays in serial communication, CAN communication, or other communication methods, the data displayed by the host computer is not real-time. Furthermore, it cannot extract multiple key real-time data points from any given time period for comparative analysis, which further complicates the development of digital power supplies. Summary of the Invention

[0005] To address the shortcomings of existing technologies, the present invention aims to provide a control and testing method for a three-phase current-source rectifier.

[0006] To achieve the above objectives, the present invention adopts the following technical solution:

[0007] A control and testing method for a three-phase current-source rectifier, comprising the following steps:

[0008] S1, Software Phase Locking

[0009] Obtain the input voltage and use a software phase-locked loop method to obtain the zero-crossing point of the three-phase input AC voltage;

[0010] S2, outer loop control

[0011] The output voltage is obtained, and at the zero-crossing point obtained by software phase-locking, the outer loop control calculation is started to stabilize the output voltage at the target value. The output value of the outer loop control is used as the reference fine-tuning amount in the inner loop PI closed-loop control.

[0012] S3, Inner Loop PI Closed-Loop Control

[0013] Obtain the input current, start the inner loop PI closed-loop control, and use the direct current control method to achieve unity power factor;

[0014] S4, seven-segment space vector modulation

[0015] The output value of the inner loop PI closed-loop control is transformed into a spatial vector control variable of the stationary two-phase coordinate system αβ axis. After sector division and switching time calculation, the spatial vector control variable is used to obtain a seven-segment PWM pulse, which drives the main circuit MOS transistor.

[0016] S5, host computer detection and visualization processing

[0017] The host computer is used to monitor the software phase-locked loop, outer loop control, inner loop PI closed-loop control and space vector control processes, obtain the change trajectory of key parameters during the control process, and perform data visualization processing.

[0018] As a further improvement to the above technical solution, in step S2, the acquisition of the output voltage, at the zero-crossing point obtained by software phase-locking, initiates outer-loop control calculation to stabilize the output voltage at the target value, and uses the output value of the outer-loop control as the reference fine-tuning amount in the inner-loop PI closed-loop control, including:

[0019] S21. Obtain the output voltage and start the outer loop control calculation at the zero-crossing point obtained by software phase-locking.

[0020] S22. Use the output value of the outer loop control as the fine-tuning value of the reference in the inner loop PI regulation and set an upper limit. Under no-load conditions, if the actual output voltage is too high, the output value of the outer loop control will exceed the upper limit, and the PWM drive pulse will be turned off to prevent over-modulation. Under normal load conditions, the outer loop control compares the actual output voltage with the reference voltage. If the actual output voltage is too high, i.e., the actual output voltage is greater than the reference voltage, the reference value of the inner loop control is decreased. Conversely, if the actual output voltage is too low, i.e., the actual output voltage is less than the reference voltage, the reference value of the inner loop control is increased. If the actual output voltage is equal to the reference voltage, the reference value of the inner loop control remains unchanged.

[0021] As a further improvement to the above technical solution, in step S3, the step of acquiring the input-side current and starting the inner-loop PI closed-loop control, using a direct current control method to achieve unity power factor, includes:

[0022] S31. Use equation (1) to determine the relationship between the inner loop control reference value and the output current target value:

[0023]

[0024] In equation (1), I drefQ Here, η is the reference value for inner-loop control, and η is the efficiency. For the target value of the output current, U out For the output voltage, U m This refers to the input voltage amplitude.

[0025] Depend on It can be seen that the condition for formula (1) to hold is the inner loop control reference value I. drefQ Equal to the input current amplitude I m .

[0026] S32. After coordinate transformation, the three-phase input AC current is used to design the control inner loop based on the dq axis rotating coordinate system; the input current is obtained, the d axis is selected as the active parameter reference axis and the q axis as the reactive parameter reference axis, and the coordinate transformation of the three-phase input AC current is performed using equations (2) and (3) to convert it to the dq axis:

[0027]

[0028]

[0029] In equation (2), I m ω0 is the amplitude of the AC current, i.e., the amplitude of the input current; ω0 is the fundamental angular frequency of the AC current; from formula (2), it can be seen that the d-axis control variable I dQ It equals the amplitude of the input current.

[0030] S33, Set the inner loop control reference I drefQ with IdQ As the input of the error amplifier, the PI regulator, after adjustment, drives the MOSFET via space vector modulation, controlling the input current amplitude I. m equals I drefQ Similarly, the q-axis variable I is adjusted via PI control. qQ When I is zero qQ When the value is zero, it indicates that the input voltage and input current are in phase, thus achieving unity power factor.

[0031] As a further improvement to the above technical solution, in step S4, the output value of the inner loop PI closed-loop control is transformed to obtain a spatial vector control variable of the stationary two-phase coordinate system αβ axis. After sector division and switching time calculation, the spatial vector control variable is used to obtain a seven-segment PWM pulse. The seven-segment PWM pulse is used to drive the main circuit MOS transistor, including:

[0032] S41. Using equation (4), the output value of the inner loop PI closed-loop control is transformed to obtain the spatial vector control variable of the αβ axis of the stationary two-phase coordinate system.

[0033]

[0034] Among them, I α and I β I is the control variable for the αβ axis. dr and I qr ω0 is the fundamental angular frequency of the AC current, and ω0 is the control variable for the dq axis.

[0035] S42. After sector division and switching time calculation, the space vector control variables are used to obtain a seven-segment PWM pulse, which is then used to drive the main circuit MOS transistor.

[0036] S43. Use equation (5) to handle the saturation phenomenon in space vector control:

[0037]

[0038] Among them, I α and I β For the control variables of the αβ axis, This is the target value for the output current.

[0039] As a further improvement to the above technical solution, in step S5, the process of using a host computer to detect the software phase-locked loop, outer loop control, inner loop PI closed-loop control, and space vector control processes, to obtain the change trajectory of key parameters during the control process, and to perform data visualization processing includes:

[0040] The S51 host computer programming software is VB6, which uses serial communication to communicate with the DSP. The communication format is: baud rate 19200bps, 1 stop bit, odd parity, 8 char bits, async mode, and data is sent or received in the following order of bytes in the data frame: [frame header][address][function code][number of bytes][data 1 high byte][data 1 low byte][data 2 high byte][data 2 low byte]……[data N high byte][data N low byte][Xor check][frame tail];

[0041] S52. Extract data of up to 3 key parameters in any time period of the software algorithm and transmit them to the external EEPROM storage area via IIC communication;

[0042] S53. The data of the key parameters is transmitted to the host computer via serial port. The host computer then performs graphical processing on the data of the key parameters and displays it.

[0043] Compared with the prior art, the advantages of the present invention are:

[0044] (1) This invention proposes a direct and stable output voltage control method that eliminates the need for an additional DC / DC converter step-down circuit at the rectifier output. This method transforms the output current control of the current-type rectifier into output voltage control. This invention not only achieves controllable output voltage for the current-type rectifier and reduces the number of DC / DC step-down circuit stages, simplifying the main circuit, but also improves the accuracy of the space vector control algorithm and the reliability of rectifier operation through anti-saturation processing of the αβ axis parameters of the stationary two-phase coordinate system.

[0045] (2) This invention also proposes a visualization method for graphically processing key real-time control parameter data at any time period. This method uses an external EEPROM chip to store the data and performs visualization processing on a host computer, which accelerates the debugging process of digital power supplies. Moreover, this method allows data to be read and written at any time without affecting the normal operation of the rectifier due to the need to transmit large amounts of data for serial communication. This invention realizes the visualization processing of key parameter data at any time period, verifies the correctness of the software algorithm, accelerates the debugging process, and allows the external EEPROM to read and write data packets of key parameters at any time without affecting the normal operation of the program. Attached Figure Description

[0046] Figure 1 This is a flowchart of the control and testing method for a three-phase current-source rectifier in this invention;

[0047] Figure 2 This is a block diagram illustrating the principle of the control and testing method for a three-phase current-source rectifier in this invention.

[0048] Figure 3 This is a structural diagram of the three-phase current-source rectifier in this invention;

[0049] Figure 4 This is a schematic diagram of the spatial vector distribution in this invention;

[0050] Figure 5 This is a schematic diagram of the six switch states of sector 1 in this invention;

[0051] Figure 6 This is a diagram showing the connection between the DSP and the external EEPROM in this invention;

[0052] Figure 7 This is a schematic diagram of the EEPROM storage area of ​​the present invention;

[0053] Figure 8 This is a control block diagram of the outer and inner loops in this invention. Figure 8 In the middle, T d For sampling and control system delay, R L L0 is the load, C0 is the output DC inductor, and C0 is the output capacitor.

[0054] Figure 9 This is a visualization waveform diagram of the three key parameter data in this invention;

[0055] Figures 10-13 The figure shows the MATLAB simulation results of the present invention, wherein, Figure 10 Simulation waveforms of the αβ axis control variables in a stationary two-phase coordinate system; Figure 11 The yellow line represents the input voltage waveform, and the red line represents the input current waveform. Figure 12 The yellow line represents the three-phase input current waveform; Figure 13 The yellow line in the middle represents the output voltage waveform. Detailed Implementation

[0056] The present invention will be further described below with reference to the accompanying drawings:

[0057] The control section of the three-phase current-source rectifier consists of software functional modules, namely space vector control, software phase-locked loop, inner loop control, and outer loop control; the detection section consists of a host computer and an EEPROM circuit. For example... Figure 2 As shown, the output voltage, after being conditioned by the sampling circuit, serves as the input variable for the outer loop control; the software phase-locked loop determines the zero-crossing point of the three-phase input AC voltage, thus providing the current grid phase angle ω0t for coordinate transformation; the input current, after being conditioned by the sampling circuit, serves as the feedback quantity for the inner loop control, thereby realizing direct current control; the host computer and EEPROM circuit realize the graphical processing of key real-time control parameter data at any time period, thereby enabling the debugging process.

[0058] like Figure 3As shown, the main circuit of the three-phase current-source rectifier includes a six-switch rectifier bridge. Each arm of the rectifier bridge uses a switch and a diode connected in series to block the reverse current on the arm and reduce the reverse voltage of the switch. The DC side uses an inductor for energy storage, which mainly filters out the ripple of the DC current.

[0059] like Figure 1 The method shown is a control and testing method for a three-phase current-source rectifier, which includes the following steps:

[0060] S1, Software Phase Locking

[0061] The input voltage is obtained, and the zero-crossing point of the three-phase input AC voltage is obtained using a software phase-locked loop method.

[0062] After software initialization, the zero-crossing point of the three-phase input AC voltage is obtained through software phase-locked loop (PLL) technology. This zero-crossing point serves as the start time for the outer loop control. Outer loop control calculations begin at the zero-crossing point to stabilize the output voltage at the target value. This operation provides an accurate start time for subsequent control steps. During unity power factor control, since the input voltage and current are in phase, starting the outer loop control at the zero-crossing point significantly reduces current spikes on the input side, making the control process more stable.

[0063] S2, outer loop control

[0064] The output voltage is acquired, and at the zero-crossing point obtained through software phase-locked loop (PLL), the outer loop control calculation is initiated to stabilize the output voltage at the target value. The output value of the outer loop control is then used as the reference fine-tuning value in the inner loop PI closed-loop control. The outer loop control calculation is simple, and the outer loop control output is only used as the reference fine-tuning value in the inner loop PI regulation.

[0065] S3, Inner Loop PI Closed-Loop Control

[0066] The input current is obtained, and the inner loop PI closed-loop control is started. The direct current control method is used to achieve unity power factor.

[0067] S4, seven-segment space vector modulation

[0068] The output value of the inner loop PI closed-loop control is transformed into a spatial vector control variable of the stationary two-phase coordinate system αβ axis. After sector division and switching time calculation, the spatial vector control variable is used to obtain a seven-segment PWM pulse, which drives the MOS transistor of the main circuit.

[0069] S5, host computer detection and visualization processing

[0070] The host computer is used to monitor the software phase-locked loop, outer loop control, inner loop PI closed-loop control, and space vector control processes, acquiring the change trajectories of key parameters during the control process and visualizing the data. The host computer is used to depict and visualize the change trajectories of key parameters during DSP software control.

[0071] As a further improvement to the above technical solution, in step S2, the acquisition of the output voltage, at the zero-crossing point obtained by software phase-locking, initiates outer-loop control calculation to stabilize the output voltage at the target value, and uses the output value of the outer-loop control as the reference fine-tuning amount in the inner-loop PI closed-loop control, including:

[0072] S21. Obtain the output voltage and start the outer loop control calculation at the zero-crossing point obtained by software phase-locking.

[0073] S22. Use the output value of the outer loop control as the fine-tuning value of the reference in the inner loop PI regulation and set an upper limit. Under no-load conditions, if the actual output voltage is too high, the output value of the outer loop control will exceed the upper limit, and the PWM drive pulse will be turned off to prevent over-modulation. Under normal load conditions, the outer loop control compares the actual output voltage with the reference voltage. If the actual output voltage is too high, i.e., the actual output voltage is greater than the reference voltage, the reference value of the inner loop control is decreased. Conversely, if the actual output voltage is too low, i.e., the actual output voltage is less than the reference voltage, the reference value of the inner loop control is increased. If the actual output voltage is equal to the reference voltage, the reference value of the inner loop control remains unchanged.

[0074] The main task of the outer loop control is to stabilize the output voltage at the target value. The above process simplifies the computational workload of the DSP and improves control efficiency. To further simplify the output voltage control of the current-type rectifier, the outer loop control does not use the traditional PI regulation method. Instead, it compares the actual output voltage with the reference voltage in the outer loop control. If the actual voltage is too high, the reference value of the inner loop control is decreased; otherwise, the reference value of the inner loop control is increased to stabilize the output voltage, thus simplifying the computational workload of the DSP.

[0075] As a further improvement to the above technical solution, in step S3, the step of acquiring the input-side current and starting the inner-loop PI closed-loop control, using a direct current control method to achieve unity power factor, includes:

[0076] S31. Use equation (1) to determine the relationship between the inner loop control reference value and the output current target value:

[0077]

[0078] In equation (1), I drefQ Here, η is the reference value for inner-loop control, and η is the efficiency. For the target value of the output current, U out For the output voltage, Um The input voltage amplitude; by It can be seen that the condition for formula (1) to hold is the inner loop control reference value I. drefQ Equal to the input current amplitude I m .

[0079] S32. After coordinate transformation, the three-phase input AC current is used to design the control inner loop based on the dq axis rotating coordinate system; the input current is obtained, the d axis is selected as the active parameter reference axis and the q axis as the reactive parameter reference axis, and the coordinate transformation of the three-phase input AC current is performed using equations (2) and (3) to convert it to the dq axis:

[0080]

[0081] In equation (2), I m ω0 is the amplitude of the AC current, i.e., the amplitude of the input current; i is the fundamental angular frequency of the AC current. A i B i C i is the instantaneous value of the grid current. dQ i qQ The grid current is converted to a dq-axis variable; from formula (2), it can be seen that the d-axis control variable I dQ It equals the amplitude of the input current.

[0082] S33, Set the inner loop control reference I drefQ with I dQ As the input of the error amplifier, the PI regulator, after adjustment, drives the MOSFET via space vector modulation, controlling the input current amplitude I. m equals I drefQ Similarly, the q-axis variable I is adjusted via PI control. qQ When I is zero qQ When the value is zero, it indicates that the input voltage and input current are in phase, thus achieving unity power factor.

[0083] As a further improvement to the above technical solution, in step S4, the output value of the inner loop PI closed-loop control is transformed to obtain a spatial vector control variable of the stationary two-phase coordinate system αβ axis. After sector division and switching time calculation, the spatial vector control variable is used to obtain a seven-segment PWM pulse. The seven-segment PWM pulse is used to drive the main circuit MOS transistor, including:

[0084] S41. Using equation (4), the output value of the inner loop PI closed-loop control is transformed to obtain the spatial vector control variable of the αβ axis of the stationary two-phase coordinate system.

[0085]

[0086] Among them, I α and Iβ I is the control variable for the αβ axis. dr and I qr ω0 is the control variable for the dq axis, and ω0 is the fundamental angular frequency of the AC current.

[0087] S42. To further reduce the harmonic content of the input current and reduce switching losses, a seven-segment space vector modulation method is proposed. After sector division and switching time calculation, the space vector control variable is used to obtain a seven-segment PWM pulse, which drives the MOSFET of the main circuit.

[0088] In step S4, space vector modulation is performed. The calculation steps for space vector modulation are as follows:

[0089] The αβ axis control quantity of the stationary coordinate system is subjected to anti-saturation processing, and the sector where the command vector is located is determined; the three nearest vectors within the sector are found to determine the switching state; the duration of each switching state is calculated from the αβ axis control quantity of the stationary coordinate system and the switching period; the calculated switching states are assigned to the corresponding switching transistors.

[0090] S43. In order to further improve the reliability of the space vector algorithm, a method for preventing saturation of the αβ axis control parameters of the stationary two-phase coordinate system is given.

[0091] The αβ axis control quantity of the stationary coordinate system is subjected to anti-saturation processing, and the sector where the command vector is located is determined. Based on the anti-saturation relationship of the αβ axis parameters of the stationary two-phase coordinate system in the space vector algorithm, anti-saturation processing is performed, that is, using equation (5) to handle the saturation phenomenon in space vector control:

[0092]

[0093] Among them, I α and I β I is the control variable for the αβ axis. d * c This is the target value for the output current.

[0094] It is necessary to ensure that formula (5) holds true at all times, otherwise saturation may occur in space vector control.

[0095] The challenge of SVPWM current control based on a fixed switching frequency lies in determining the sector into which the space vector falls, thereby obtaining the three switching state components of the synthesized command vector. For example... Figure 4As shown, this invention divides a power grid frequency cycle into six sectors. Based on the principle of minimizing the number of switching transistors, the space vector is allocated according to the sector operation sequence. This ensures that the seven-segment space vector modulation method achieves lower input current harmonics while also reducing switching losses. The sector operation sequence and the corresponding space vector allocation for each sector in the seven-segment space vector modulation method minimize switching losses to the greatest extent. The sector operation sequence and the selection of the corresponding space vector for each sector are shown in Table 1.

[0096] Table 1 Sector Action Sequence and Spatial Vector Selection Table

[0097] sector <![CDATA[I n ]]> <![CDATA[I n ]]> <![CDATA[I0]]> 1 <![CDATA[I6]]> <![CDATA[I1]]> <![CDATA[I7]]> 2 <![CDATA[I1]]> <![CDATA[I2]]> <![CDATA[I9]]> 3 <![CDATA[I2]]> <![CDATA[I3]]> <![CDATA[I8]]> 4 <![CDATA[I3]]> <![CDATA[I4]]> <![CDATA[I7]]> 5 <![CDATA[I4]]> <![CDATA[I5]]> <![CDATA[I9]]> 6 <![CDATA[I5]]> <![CDATA[I6]]> <![CDATA[I8]]>

[0098] In Table 1, sector 1 is composed of the target vector synthesized from I6, I1, and I7. When these three switch states switch, as follows: Figures 3-5 As shown, the upper bridge arm switches Q1 to Q3 remain normally open or normally closed, while only the lower bridge arm switches Q4 to Q6 are turned on or off, resulting in a low number of switching cycles. Similar to sector 1, sectors 2 to 6 each have a set of upper or lower bridge arm switches that remain normally on or normally off within one switching cycle, thereby reducing switching losses.

[0099] The space vector magnitude of a three-phase current-source PFC is Generally, the inner circle of the space vector represents the upper limit of the command vector magnitude; exceeding this limit results in overmodulation. The command vector magnitude of the inner circle is equal to the radius of the outer circle. The magnitude of the instruction vector is then... Thus, the magnitude of the command vector synthesized from the αβ axis control quantities in the stationary two-phase coordinate system does not exceed the target value of the output current. The conclusion is that the calculation of the magnitude of the synthesized command vector includes the target output voltage variable.

[0100] The above theoretical analysis shows that it is necessary to prevent saturation phenomena that may occur in space vector operations. If saturation occurs, it will lead to errors in the calculation of the pulse width of the switching transistor, and in severe cases, the switching transistor in the main circuit will burn out.

[0101] As a further improvement to the above technical solution, in step S5, the process of using a host computer to detect the software phase-locked loop, outer loop control, inner loop PI closed-loop control, and space vector control processes, to obtain the change trajectory of key parameters during the control process, and to perform data visualization processing includes:

[0102] The S51 host computer programming software is VB6, which uses serial communication to communicate with the DSP. The communication format is: baud rate 19200bps, 1 stop bit, odd parity, 8 char bits, async mode, and data is sent or received in the following order of bytes in the data frame: [frame header][address][function code][number of bytes][data 1 high byte][data 1 low byte][data 2 high byte][data 2 low byte]……[data N high byte][data N low byte][Xor check][frame tail].

[0103] S52. Data from up to three key parameters at any given time period in the software algorithm is extracted and transmitted to an external EEPROM storage area via IIC communication. Three key parameters are selected according to testing needs. The data for each key parameter consists of 210 data points, which are stored in the external EEPROM in real time via IIC communication. To further simplify the debugging process of the control algorithm, the testing method uses the IIC communication mode of the EEPROM, allowing for reading and writing of key parameter data packets at any time, and enabling data visualization on the host computer.

[0104] S53. The data of the key parameters is transmitted to the host computer via serial port. The host computer then performs graphical processing on the data of the key parameters and displays it.

[0105] These data can be transmitted to the host computer via serial port at any time for graphical representation without affecting the normal program operation of the rectifier's main control DSP. Each key parameter data consists of 210 data points, meaning that 210 data points constitute the key parameter data packet. The data point interval is the cycle of the DSP interrupt control program. For example, if the interrupt frequency is 100kHz, then three key parameter data points with an arbitrary time period of 210 / 100kHz = 2.1ms can be sampled and graphically represented simultaneously. The communication method between the EEPROM and the DSP is as follows: Figure 6 As shown, the data range allocation of the EEPROM is as follows: Figure 7 As shown.

[0106] The relationship between the target value of the output current and the output voltage in this embodiment is shown in equation (6):

[0107]

[0108] In equation (6), I m Where I is the input current amplitude, η is the efficiency, and I is the input current amplitude. d * c For the target value of the output current, U o * ut For the target output voltage, Um This represents the input voltage amplitude.

[0109] Table 2 shows the distribution of the action time of adjacent spatial vectors in the seven-segment modulation method, used to calculate the switching action time in sectors 1 to 6. For example... Figure 5 As shown, taking the seven-segment space vector control of sector 1 as an example, in Table 2, T1 represents the switching action time of stage I6 as Y, T2 represents the switching action time of stage I1 as X, and T... S -T1-T2 is the switching action time of stage I7, T s The switching cycle.

[0110] Table 2. Time Distribution of Switch Status Effects (X, Y, Z)

[0111] sector 1 2 3 4 5 6 <![CDATA[T1]]> Y -Y X -X -Z Z <![CDATA[T2]]> X Z -Z -Y Y -X

[0112] The calculation expressions for X, Y, and Z in Table 2 are as follows:

[0113]

[0114] In equation (7), T represents the DC output current. s For the switching period, I' allfa and I' beta Let αβ be the control variable in the stationary coordinate system. The magnitude of the synthesized command vector is... Substituting formula (6) into formula (7), the target output voltage value can be obtained. The switching state duration is calculated. The synthesized command vector calculation includes the target output voltage variable. To further simplify the output voltage control of the current-type rectifier, the target output voltage variable is included in the magnitude calculation of the synthesized command vector in the space vector algorithm.

[0115] like Figure 8 As shown, after determining the target output voltage value, the output voltage can be stabilized by fine-tuning the inner loop control reference value through the outer loop, and then adjusting it through the inner loop PI control. The rectifier is simulated using MATLAB, and the simulation results are as follows. Figures 10-13 As shown.

[0116] like Figure 5 As shown, taking the seven-segment space vector control of sector 1 as an example, according to the sector operation sequence and space vector selection method in Table 1, the number of switching transistors is small. The upper bridge arm switch remains normally on or normally off within one switching cycle, and the lower bridge arm switch generates PWM drive pulses based on the calculated switching state and switching time. The DSP software programming is also simple; the upper bridge arm switch does not need to calculate the pulse width and can be directly given a high / low level, while only the lower bridge arm switch needs to calculate the drive pulse width.

[0117] Taking the drive pulse programming of the upper / lower bridge arm switch of phase A as an example, the DSP programming statement for the upper bridge arm switch is as follows:

[0118] Epwm1.AQCTLA.ZRO=2;

[0119] Epwm1.AQCTLB.ZRO=2;

[0120] The DSP programming statements for the lower bridge arm switch are as follows:

[0121] Epwm1.AQCTLB.bit.CAU = 2;

[0122] Epwm1.AQCTLB.bit.CAD = 1;

[0123] EPwm1Regs.CMPA.half.CMPA=cmp1;

[0124] Epwm1.AQCTLB.bit.CBU=1; Epwm1.AQCTLB.bit.CBD=2;

[0125] EPwm1Regs.CMPb = cmp3.

[0126] As shown in formula (5), the magnitude of the synthesized instruction vector does not exceed I. d * c This can prevent saturation phenomena that may occur in space vector operations. The calculation steps for saturation prevention are as follows:

[0127] If formula (5) holds, then the control variable I of the αβ axis of the stationary two-phase coordinate system α and I β Output directly without modification; if formula (5) is not valid, then control variable I α and I β The calculation relationship is shown in equation (8):

[0128]

[0129] According to formula (8), control variable I α and I β The correction effectively prevents saturation and increases the accuracy of space vector modulation and the reliability of rectifier operation.

[0130] like Figure 6As shown, the DSP's IIC interface is directly connected to the corresponding pins of the EEPROM. EEPROM pins A0-A2 are shorted to ground, indicating that the EEPROM's physical address on the IIC bus is 0x50 and it is a slave device. The clock signal SCL generated by the master device DSP is sent to the bus. According to the IIC communication protocol, the DSP and EEPROM convert the data into serial signals and send them to the SDA line, achieving bidirectional data transmission.

[0131] like Figure 7 As shown, the EEPROM uses bytes as the storage unit and can store a maximum of 3 key parameters. Each key parameter contains 210 data points, and the data type is bytes. To obtain visualization effects as quickly as possible, the 210 data points of each key parameter are divided into 3 arrays. After the 70 data points of each array are transmitted via serial communication, the graph is drawn on the host computer. Therefore, the 3 key parameter data packets need to be drawn into graphs 9 times on the host computer, and the results of each drawing are displayed on the same interface of the host computer.

[0132] When the rectifier software control algorithm needs to capture the change trajectory of key parameters for a specific time period, the DSP extracts the relevant data for that period in the program and stores it in the DSP's memory. After the rectifier stops, the DSP begins transferring a total of 3*2^10 data packets from memory to the EEPROM. Therefore, data transmission on the IIC bus does not affect the normal operation of the rectifier. Whenever it is necessary to observe these data, the DSP reads the data from the EEPROM and stores it back in memory, then transmits the data packets to the host computer via serial communication for visualization processing.

[0133] In this embodiment, the host computer programming software is VB6. The 3*210 data points are divided into 9 data packets for serial communication. After each data packet is transmitted, the host computer plots the trajectory curve of the data change until the trajectory curves of all three key parameters are plotted. The three plotted trajectory curves are displayed on one interface, allowing for comparative analysis of the change trajectories of the three key parameters within a certain time period in the software algorithm. This enables accurate judgment of the correctness of the software algorithm and the rationality of the control parameter settings. The host computer's visual graphical plotting result is as follows: Figure 9 As shown, the simulation results for the same key parameters are as follows: Figure 10 As shown.

[0134] Figure 9The image shows the trajectory of the α-β axis control variables in the DSP software within a stationary two-phase coordinate system. The green line represents the A-phase input voltage waveform, the red line represents the α-axis control variable waveform, and the blue line represents the β-axis control variable waveform. The α-β axis variables in the stationary two-phase coordinate system are vector control variables, and the phase difference between these two variables is 90°. Furthermore, when the input voltage and input current are in phase, i.e., unity power factor is achieved, the phase difference between the α-axis control variable and the A-phase input voltage waveform is 0. Figure 9 Use a visual waveform to indicate whether unity power factor has been achieved.

[0135] Figure 10 The phase difference between the αβ axis variables in the stationary two-phase coordinate system was verified to be 90° using simulation methods. Figure 11 Simulations verified that the input voltage and input current being in phase indicates unity power factor. Figure 12 Simulations verified that the phase difference of the three-phase AC current is 120° and the three-phase AC current exhibits a sinusoidal variation, achieving the design goal of minimizing the total harmonic distortion rate of the AC input current in power factor correction. Figure 13 Simulations verified that the output voltage of the three-phase rectifier was stable, achieving the design goal of stable output voltage in power factor correction.

[0136] In summary, this invention provides a control and testing method for a three-phase current-source rectifier. The core control consists of an outer loop and an inner loop. The outer loop does not employ PI regulation, simplifying the DSP calculation process. Seven-segment space vector modulation is used, achieving lower input current harmonics while reducing switching losses. Anti-saturation processing improves the correctness of the space vector control algorithm and the reliability of rectifier operation. Visualization of key parameters accelerates the design and debugging process and reduces design complexity. The control method in this invention effectively achieves unity power factor, and the detection method graphically depicts the trajectory changes of key parameters in the software algorithm, further accelerating the design and debugging process and reducing design complexity.

[0137] The above-described embodiments are merely preferred embodiments of the present invention and are not intended to limit the scope of the present invention. Various modifications and improvements made by those skilled in the art to the technical solutions of the present invention without departing from the spirit of the present invention should fall within the protection scope defined by the claims of the present invention.

Claims

1. A control and testing method for a three-phase current-source rectifier, characterized in that, The method includes the following steps: S1, Software Phase Locking Obtain the input voltage and use a software phase-locked loop method to obtain the zero-crossing point of the three-phase input AC voltage; S2, outer loop control The output voltage is obtained, and at the zero-crossing point obtained by software phase-locking, the outer loop control calculation is started to stabilize the output voltage at the target value. The output value of the outer loop control is used as the reference fine-tuning amount in the inner loop PI closed-loop control. S3, Inner Loop PI Closed-Loop Control Obtain the input current, start the inner loop PI closed-loop control, and use the direct current control method to achieve unity power factor; S4, seven-segment space vector modulation The output value of the inner-loop PI closed-loop control is transformed to obtain a stationary two-phase coordinate system. αβ The space vector control variable of the axis, after sector division and switching time calculation, is used to obtain a seven-segment PWM pulse, which drives the MOS transistor of the main circuit. S5, host computer detection and visualization processing The host computer is used to monitor the software phase-locked loop, outer loop control, inner loop PI closed-loop control and space vector control processes, obtain the change trajectory of key parameters during the control process, and perform data visualization processing. In step S2, acquiring the output voltage, and starting the outer loop control calculation at the zero-crossing point obtained by software phase-locked loop to stabilize the output voltage at the target value, and using the output value of the outer loop control as the reference fine-tuning amount in the inner loop PI closed-loop control, includes: S21. Obtain the output voltage and start the outer loop control calculation at the zero-crossing point obtained by software phase-locking. S22. Use the output value of the outer loop control as the fine-tuning value of the inner loop PI regulation and set an upper limit. Under no-load conditions, if the actual output voltage is too high, the output value of the outer loop control will exceed the upper limit, and the PWM drive pulse will be turned off to prevent over-modulation. Under normal load conditions, the outer loop control compares the actual output voltage with the reference voltage. If the actual output voltage is too high, i.e., the actual output voltage is greater than the reference voltage, the reference value of the inner loop control is decreased. Conversely, if the actual output voltage is too low, i.e., the actual output voltage is less than the reference voltage, the reference value of the inner loop control is increased. If the actual output voltage is equal to the reference voltage, the reference value of the inner loop control remains unchanged. In step S3, obtaining the input-side current and initiating the inner-loop PI closed-loop control, using a direct current control method to achieve unity power factor, includes: S31. Use equation (1) to determine the relationship between the inner loop control reference value and the output current target value: （1） In equation (1), I drefQ This is the reference value for inner loop control. η For efficiency, The target value for output current, U out For output voltage, U m This refers to the input voltage amplitude. S32. Obtain the input side current and select... d The shaft is the active parameter reference shaft. q The axis is the reactive power parameter reference axis. Using equations (2) and (3), the coordinate transformation of the three-phase input AC current is performed to convert it to the reference axis. dq axis: （2） （3） In equation (2), I m ω0 is the amplitude of the alternating current, i.e., the input current amplitude; ω0 is the fundamental angular frequency of the alternating current. i A , i B , i C This represents the instantaneous value of the grid current. i dQ , i qQ For grid current conversion to dq Axis variable; as can be seen from formula (2), d Axis control variables I dQ Equal to the input current amplitude; S33, Set the inner loop control reference I drefQ and I dQ As the input of the error amplifier, the PI regulator, after adjustment, drives the MOSFET via space vector modulation to control the input current amplitude. I m equal I drefQ Similarly, by using PI regulation... q Axis variables I qQ When it is zero, I qQ When the value is zero, it indicates that the input voltage and input current are in phase, thus achieving unity power factor.

2. The control and testing method for a three-phase current-source rectifier according to claim 1, characterized in that, In step S4, the output value of the inner loop PI closed-loop control is transformed to obtain a stationary two-phase coordinate system. αβ axis The space vector control variable, after sector division and switching time calculation, yields a seven-segment PWM pulse. This seven-segment PWM pulse drives the main circuit MOSFET, including: S41. Using equation (4), perform coordinate transformation on the output value of the inner loop PI closed-loop control to obtain a stationary two-phase coordinate system. αβ The space vector control variable of the axis; （4） in, I α and I β for αβ The control variables of the axis, I dr and I qr for dq The control variable for the axis, ω0 is the fundamental angular frequency of the alternating current; S42. After sector division and switching time calculation, the space vector control variables are used to obtain a seven-segment PWM pulse, which is then used to drive the main circuit MOS transistor. S43. Use equation (5) to handle the saturation phenomenon in space vector control: （5） in, I α and I β for αβ The control variables of the axis, This is the target value for the output current.

3. The control and testing method for a three-phase current-source rectifier according to claim 2, characterized in that, In step S5, the process of using a host computer to detect the software phase-locked loop, outer loop control, inner loop PI closed-loop control, and space vector control processes, to obtain the change trajectory of key parameters during the control process, and to perform data visualization processing includes: The S51 host computer programming software is VB6, which uses serial communication to communicate with the DSP. The communication format is: baud rate 19200bps, 1 stop bit, odd parity, 8 char bits, async mode, and data is sent or received in the following order of bytes in the data frame: [Frame Header] [Address] [Function Code] [Number of Bytes] [Data 1 High Byte] [Data 1 Low Byte] [Data 2 High Byte] [Data 2 Low Byte] ... [Data N High Byte] [Data N Low Byte] [Xor Check] [Frame Tail]; S52. Extract data of up to 3 key parameters from any time period in the software algorithm and transmit them to the external EEPROM storage area via IIC communication; S53. The data of the key parameters is transmitted to the host computer via serial port. The host computer then performs graphical processing on the data of the key parameters and displays it.

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