Three-phase-shift frequency modulation control method and system suitable for SS compensation wireless power transmission system

Abstract

The invention discloses a three-phase shift frequency modulation control method and system suitable for an SS compensation wireless power transmission system, and belongs to the field of wireless power transmission, and the method comprises the steps: achieving the minimum reactive ZVS of all switching tubes of a primary inverter and a secondary rectifier in a system steady state through the real-time adjustment of an inter-bridge external phase shift angle delta and a switching frequency f; closed-loop adjustment of wide output voltage gain is realized through PI control of an inner phase shift angle gamma 2 between bridge arms of the rectifier; through system voltage gain derivation, parameter scanning and least square fitting, an optimal efficiency working curve gamma1 = f (gamma2) is obtained so as to realize real-time tracking of the highest efficiency working point under each working condition of the system. The four control freedom degrees of phase shift angles gamma 1, gamma 2 and delta and the switching frequency f are fully and reasonably utilized, the minimum reactive ZVS of all switching tubes of the system, system wide output voltage gain adjustment and efficiency optimization are achieved, and introduction of an extra power hardware link is avoided.

Description

Technical Field

[0001] The present invention belongs to the field of wireless power transmission, and more specifically, relates to a three-phase shift frequency modulation control method and a control system suitable for an SS compensated wireless power transmission system (Wireless Power Transfer, WPT). Background Art

[0002] At present, the control strategy of WPT system mainly focuses on the realization of zero voltage switching (ZVS) of converter, wide range power regulation and efficiency optimization. Specific control methods include frequency modulation, phase shift and pulse density modulation.

[0003] In order to achieve ZVS of the primary inverter and the secondary rectifier, some scholars have proposed a dual-active fixed-frequency phase-shift control strategy that requires real-time and accurate acquisition of the primary and secondary high-frequency current phases. However, the accuracy of current phase detection is easily affected by noise, which makes it difficult to apply to high-frequency resonant networks. To solve the above problems, a dynamic ZVS angle control method based on uniform time delay compensation (UTDCM) can effectively improve the accuracy of phase detection. Since this method involves multiple closed loops, cross-coupling between the loops may lead to system-level instability. On the other hand, some scholars have achieved output power regulation and ZVS using variable frequency control. However, the wide range of switching frequency adjustment causes serious system detuning, which will greatly reduce system efficiency. Based on this, subsequent studies further optimized the combination of frequency and phase shift angle to minimize frequency deviation. However, iterative calculation based on the model will greatly affect the dynamic performance of the system. In view of this, it is expected to develop a simple control method to achieve ZVS and adjustable output voltage without any additional DC-DC converters, switching elements or complex resonant networks.

[0004] At present, the methods for optimizing the efficiency of WPT systems can be mainly divided into three categories: constructing optimal load impedance, feedback control based on wireless communication, and feedback control without wireless communication. A direct method based on constructing the optimal load impedance is to use the duty cycle control of the DC-DC converter to power the actual load, and according to the type of power converter, a duty cycle control algorithm can be found to adjust the input impedance of the DC-DC converter so that the load is equal to the optimal load impedance value. For feedback control methods without wireless communication, a common method is to adjust the duty cycle of the transmitting inverter for any output power through the perturbation and observation (P&O) method to obtain the minimum input power, thereby automatically maximizing the energy efficiency of the entire WPT system. However, the above two methods often require the addition of additional converters or auxiliary circuits, and the power circuit structure is cumbersome. In contrast, the feedback control method based on wireless communication can directly use the active rectifier bridge to achieve accurate and stable tracking of the optimal efficiency point without adding additional power hardware units.

[0005] In summary, existing research on wireless power transmission system control methods has described various methods to ensure inverter ZVS operation, power or gain controllability, and efficiency optimization. However, how to fully utilize the control freedom without adding additional power hardware units, achieve ZVS, and maintain optimal efficiency over a wide load range still remains a design challenge. Summary of the invention

[0006] In view of the defects of the prior art and the need for improvement, the present invention discloses a three-phase shift frequency modulation control method and system suitable for an SS compensated wireless power transmission system, the purpose of which is to fully utilize the control freedom of the SS compensated dual active bridge topology without adding any additional power hardware units, so that all power MOS tubes of the system operate at the ZVS operating point with the minimum reactive power in steady state, while achieving wide range controllability of the output voltage and optimization of dcdc efficiency.

[0007] To achieve the above object, the present invention provides a three-phase shift frequency modulation control method suitable for SS compensation wireless power transmission system, which is applied to SS resonance compensation dual active bridge circuit, wherein the SS resonance compensation dual active bridge circuit includes a DC voltage source, a phase shift full bridge inverter, an SS resonance compensation network, a magnetic coupling mechanism, a phase shift full bridge rectifier, a filter capacitor, and a load resistor; the phase shift full bridge inverter includes a power MOS tube S 1 ~S 4 The phase-shifted full-bridge rectifier includes a power MOS tube S 5 ~S 8; The magnetic coupling mechanism includes a primary transmitting coil and a secondary receiving coil; The SS resonant compensation network includes a primary capacitor C1 and a secondary capacitor C2; The three-phase shift frequency modulation control method includes four independent control degrees of freedom: switching frequency f, S 5 Advanced S 1 The phase angle δ, S 1 Advanced S 4 Phase angle of opening γ 1 , S 5 Advanced S 8 Phase angle of opening γ 2 , 3 control objectives: tracking the minimum reactive ZVS operating point of all power MOS tubes by adjusting f and δ, and controlling the minimum reactive ZVS operating point of all power MOS tubes by constraining γ 1 and γ 2 The numerical relationship between them achieves the optimal dcdc efficiency and adjusts γ 1 and γ 2 Achieve a wide range of controllable output voltage gain.

[0008] Furthermore, the tracking of the minimum reactive ZVS operating point of all power MOS tubes by adjusting f and δ includes:

[0009] Analyze the phase conditions that the voltage and current of all power MOS tubes should meet when they are at the minimum reactive ZVS operating point;

[0010] Under the phase condition, a steady-state phasor model of the SS resonant compensation dual active bridge circuit is established;

[0011] In combination with Kirchhoff's law, the circuit equation is solved by the steady-state phasor model to derive the phasor expressions of key state quantities of the circuit;

[0012] The phasor diagram is drawn according to the phasor expression and phase condition. The constraints that f and δ should satisfy at the minimum reactive ZVS operating point are obtained according to the geometric relationship in the phasor diagram. Thus, the tracking of the minimum reactive ZVS operating point under different working conditions can be achieved by adjusting f and δ in real time.

[0013] Furthermore, the constraint γ 1 and γ 2 The numerical relationship between achieves the optimal dcdc efficiency, including:

[0014] γ 1 and γ 2 Sweep parameters in the range of 0 to π to obtain different γ 1 and γ 2 The system output voltage V dc and efficiency;

[0015] Select a continuous monotonic interval and require δV dc / δγ1 <0, δV dc / δγ 2 <0;

[0016] Select the upper and lower limits of the output voltage and the incremental step size to obtain the γ 1 and γ 2 Working curve between

[0017] Select the γ with the best efficiency on the working curve under each output voltage 1 and γ 2 Combine and draw a line, perform least squares linear fitting, and finally get the working curve γ 1 =f(γ 2 ) to achieve optimal dcdc efficiency.

[0018] Furthermore, by adjusting γ 1 and γ 2 Achieve a wide range of controllable output voltage gain, including: output voltage V dc Perform closed-loop control and input the deviation signal into the PI controller output γ 2 , and then by γ 1 =f(γ 2 ) to obtain γ 1 , thereby guiding the generation of PWM drive signals and driving the switching action to achieve controllable output voltage gain.

[0019] The present invention also provides a three-phase frequency modulation control system suitable for SS compensation wireless power transmission system, including: SS resonance compensation dual active bridge circuit, DSP chip, control circuit; wherein the DSP chip is used to execute the three-phase frequency modulation control method, and then receive the sampling signal and issue the control instruction through the isolation, signal amplification, filtering, driving and other control circuits to drive the switch devices in the SS resonance compensation dual active bridge circuit to complete the power control. The three-phase frequency modulation control method includes 4 independent control degrees of freedom: the unified switching frequency f of all power switch devices of the primary inverter and the secondary rectifier, S 5 Advanced S 1 The phase angle δ, S 1 Advanced S 4 Phase angle of opening γ 1 , S 5 Advanced S 8 Phase angle of opening γ 2 By adjusting the above four control variables, three core control objectives can be achieved: by adjusting f and δ, the minimum reactive ZVS working point of all power MOS tubes can be solved and output in real time; by constraining γ 1 and γ 2 The numerical relationship between them achieves the optimal dcdc efficiency and adjusts γ 1 and γ2 Achieve a wide range of controllable output voltage gain.

[0020] Furthermore, the SS resonant compensation dual active bridge circuit includes a DC voltage source, a phase-shifted full-bridge inverter, an SS resonant compensation network, a magnetic coupling mechanism, a phase-shifted full-bridge rectifier, a filter capacitor, and a load resistor; wherein the phase-shifted full-bridge inverter includes a power MOS tube S 1 ~S 4 , power MOS tube S 1 ~S 4 Each of them has an anti-parallel body diode; the phase-shifted full-bridge rectifier includes a power MOS tube S 5 ~S 8 , power MOS tube S 5 ~S 8 Each has an anti-parallel body diode; the SS resonant compensation network includes a primary capacitor C 1 and the secondary capacitor C 2 The magnetic coupling mechanism includes a primary transmitting coil and a secondary receiving coil; the circuit topology model includes a primary self-inductance L 1 , the original side line loss resistance R 1 , Secondary side self-inductance L 2 , secondary line loss resistance R 2 , mutual inductance M; let the center frequency be f c , then:

[0021]

[0022] To achieve ZVS with minimum reactive power, the phase difference between the AC voltage and current of the inverter and rectifier should be exactly equal to γ 1 / 2 and -γ 2 / 2. Perform the Thevenin equivalent on the system circuit topology and write the KVL equation on this basis to obtain the following formula:

[0023]

[0024] in, are the fundamental phasors of the inverter output voltage and the rectifier input voltage (correspondingly, U 1 , U 2 are the fundamental amplitudes of the inverter output voltage and the rectifier input voltage respectively), are the fundamental phasors of the inverter output current and the rectifier input current, respectively. and The reference direction is opposite, and The reference direction is the same (correspondingly, I 1 ,I 2are the fundamental amplitudes of the inverter output current and the rectifier input current respectively), ω=2πf is the switching angular frequency of the system, ω 0 =2πf c is the central angular frequency of the system; the meaning of k and its expression are shown in the following formula:

[0025]

[0026] Then we have:

[0027]

[0028] Ignoring the line loss resistance, when the system works at the optimal ZVS working point, the following formula is obtained:

[0029]

[0030] Where i = 1, 2, are the power factor angles at the AB and ab terminals, V in is the inverter DC bus voltage, V dc is the rectifier output voltage. According to basic equation 1, basic equation 2, and the optimal operating point constraint, the system phasor diagram can be drawn. Using the sine theorem and cosine theorem, it is easy to obtain:

[0031]

[0032] Among them, γ 1 , γ 2 It can be obtained by receiving the previous stage calculation result from the DSP chip, U 1 , U 2 V can be obtained by sampling in 、V dc After that, we get that the inductance is a known quantity. Then the unknown quantities are only k and the phase shift angle δ related to the frequency f, which can be actively adjusted to meet the constraint equation, so that the system can track the optimal ZVS operating point.

[0033] Furthermore, considering the line loss, the present invention derives the rectifier input (ab terminal) voltage amplitude and dcdc efficiency expression as follows:

[0034]

[0035] Combine constraint 1, constraint 2, fundamental equivalent 1, fundamental equivalent 2, output voltage, and the definition of variable k. 1 ,γ 2 |0<γ 1 <π,0<γ 2 <π} within the range of all operating points (γ 2 ,γ 1 ) Solve the above equations to obtain different γ1 and γ 2 The system output voltage and efficiency matrix under the combination; excluding the unsolvable intervals and selecting the continuous monotonic intervals (requiring δV dc / δγ 1 < 0, δV dc / δγ 2 < 0), finally obtaining the method feasible region; selecting the upper and lower limits of the output voltage and the increment step length, and respectively obtaining the working curves between γ 1 and γ 2 at each output voltage; selecting the working points with the optimal efficiency on the working curves of each output voltage (γ 2 , γ 1 ) and plotting them in the two-dimensional coordinate system, performing linear fitting by the least squares method, and finally obtaining the optimal efficiency working curve γ 1 = f(γ 2 ).

[0036] Furthermore, perform closed-loop control on the output voltage V dc , sample to obtain V in and V dc ; process the deviation signal and input it into the PI control method in the DSP chip to output the γ 2 instruction, and then obtain γ 1 from γ 2 = f(γ 1 ), and then input it into the ZVS constraint link to solve the constraint equations 1 and 2 to obtain δ and f; input the above 4 control variables into the PWM generation link to drive the switch to act, completing the entire control process.

[0037] Generally speaking, through the above technical solutions conceived by the present invention, the following beneficial effects can be achieved:

[0038] (1) Compared with the traditional voltage regulation method and the optimal load impedance of the wireless power transmission system, and the feedback control efficiency optimization method without wireless communication, the present invention does not need to add additional power hardware links, fully utilizes the control freedom of the inverter and the active rectifier, effectively reduces the system redundancy, and reduces the dynamic interference between the front and rear stages of the system;

[0039] (2) By simultaneously adjusting the inverter phase shift angle γ 1 and the rectifier phase shift angle γ 2 to achieve the regulation of the output voltage, compared with the single phase shift control method, the voltage regulation range can be broadened and the efficiency can be optimized;

[0040] (3) The three-phase shift frequency modulation control method for SS compensated wireless power transmission system proposed in the present invention successfully achieves the three core goals of ZVS, voltage regulation and efficiency optimization in the same system without adding any additional hardware links, which well copes with the common problems faced by wireless power transmission systems and has strong comprehensive performance and application value. BRIEF DESCRIPTION OF THE DRAWINGS

[0041] Figure 1 A block diagram and a system schematic diagram of a three-phase shift frequency modulation control method applicable to an SS compensation wireless power transmission system provided by the present invention;

[0042] Figure 2 The present invention provides (a) SS compensation wireless power transmission system circuit topology and (b) fundamental wave equivalent simplified topology;

[0043] Figure 3 When the topology of the present invention realizes full-tube ZVS 1 、i 2 The flow conditions that should be met and the corresponding u AB 、i 1 、u ab 、i 2 The phase conditions that should be met;

[0044] Figure 4 The phasor diagrams corresponding to (a) basic equation 1 and (b) basic equation 2 when the system of the present invention realizes ZVS under minimum reactive power of all power MOS tubes (taking 0<δ<90° as an example);

[0045] Figure 5 The system described in the embodiment of the present invention is in {γ 1 ,γ 2 |0<γ 1 <π,0<γ 2 (a) Gain and (b) efficiency distribution diagrams for all feasible operating points within the range <π};

[0046] Figure 6 are all the optimal efficiency operating points (γ) of the system (a) at various gains described in the embodiment of the present invention. 1 ,γ 2 ) and the optimal efficiency working curve γ obtained by least squares fitting 1 =f(γ 2 ), and (b) the η-P curve for all operating points on the curve;

[0047] Figure 7 A simulation schematic diagram of the closed-loop regulation of the system load voltage according to an embodiment of the present invention;

[0048] Figure 8When the DC output voltage V dc Voltage and current waveforms on the AC side of the inverter and rectifier when controlled to (a) 45 V and (b) 10 V. DETAILED DESCRIPTION

[0049] In order to make the purpose, technical solutions and advantages of the present invention more clearly understood, the present invention is further described in detail below in conjunction with the accompanying drawings and embodiments. It should be understood that the specific embodiments described herein are only used to explain the present invention and are not intended to limit the present invention. In addition, the technical features involved in the various embodiments of the present invention described below can be combined with each other as long as they do not conflict with each other.

[0050] In the present invention, the terms "first", "second", etc. (if any) in the present invention and the accompanying drawings are used to distinguish similar objects but not necessarily to describe a specific order or sequence.

[0051] Example:

[0052] See also Figure 1 This embodiment provides a three-phase frequency modulation control system suitable for SS compensation wireless power transmission system, which is composed of SS resonant compensation dual active bridge circuit topology, three-phase frequency modulation control method, and control circuit. Among them, the SS resonant compensation dual active bridge circuit topology is responsible for realizing power conversion, the control method is responsible for waveform and power control, and the control circuit is responsible for the hardware implementation of the control method. The basic parameters used in this embodiment are shown in Table 1.

[0053] Table 1

[0054] parameter Numeric parameter Numeric <![CDATA[Resonant frequency f c (kHz)]]> 100 <![CDATA[Primary self-inductance L 1 (μH)]]> 40 <![CDATA[Secondary self-inductance L 2 (μH)]]> 40 Mutual inductance M(μH) 10 <![CDATA[Primary side capacitor C 1 (nF)]]> 63.3 <![CDATA[Secondary side capacitor C 2 (nF)]]> 63.3 <![CDATA[Original edge line loss resistance R 1 (Ω)]]> 0.135 <![CDATA[Auxiliary side line loss resistance R 2 (Ω)]]> 0.135 Load resistance R(Ω) 5 <![CDATA[Source voltage V in (V)]]> 78.54V

[0055] First, the system needs to be modeled in a steady state. Specifically, due to the existence of the resonant compensation unit, the fundamental wave equivalent and Thevenin equivalent can be performed on the left side of port AB and the right side of port ab, respectively, and finally a simplified topology is obtained as follows: Figure 2 As shown in (b) in the figure, the phasor method is used to replace the time domain quantities in the system with phasors.

[0056] Further, in Figure 2 The zero voltage turn-on condition of each MOS tube is analyzed based on the topology shown in (a). The basic principle is that the current i 1 or 2 It can make the anti-parallel diode conduct, thereby realizing the voltage clamping at both ends of the switch tube and turning it on with a terminal voltage close to 0. Based on this idea, it is easy to obtain the zero voltage turn-on time of S1~S8 tubes. 1 with i 2 The flow conditions to be met (in Figure 2The current reference direction in (a) is positive) as shown in Table 2.

[0057] Table 2

[0058] Switch tube serial number <![CDATA[i 1 Flow Direction]]> Switch tube serial number <![CDATA[i 2 Flow Direction]]> S1 - S5 + S2 + S6 - S3 + S7 - S4 - S8 +

[0059] Furthermore, in order to meet the above current flow conditions and reduce the reactive circulating current as much as possible, the power factor angle at the inverter output or the rectifier input should be as small as possible. Therefore, if there is an operating point where the inverter and rectifier power factor angles are simultaneously minimized, it is called the minimum reactive ZVS operating point. The waveform at this time is as follows: Figure 3 shown.

[0060] Furthermore, combined with Figure 3 The constraints of the working conditions shown in the figure can respectively obtain the phasor diagrams of basic equations 1 and 2 as follows Figure 4 As shown in (a) and (b) (taking 0<δ<90° as an example). By using the sine theorem and cosine theorem and classifying them, we can get a universal ZVS constraint equation set. This equation set strictly limits the system to any feasible γ 1 ,γ 2 To achieve the minimum reactive power ZVS, the given δ and f are required. For example, if the received command from the superior is γ 1 =γ 2 =0, then after calculation we get δ=0.5π, f=100kHz.

[0061] Furthermore, for {γ 1 ,γ 2 |0<γ 1 <π,0<γ 2 <π} within the range of all operating points (γ 2 ,γ 1 ) Solve the equations to get different γ 1 and γ 2 The system output voltage and efficiency matrix under the combination; eliminate the interval without solution and select the continuous monotonic interval (requires δV dc / δγ 1 <0, δV dc / δγ 2 <0), and finally the feasible domain of the method is obtained, such as Figure 5 shown.

[0062] Get γ at each output voltage 1 and γ 2 Select the working point with the best efficiency on each output voltage working curve (γ 2 ,γ 1 ) and plotted in a two-dimensional coordinate system, and the least squares linear fitting was performed to finally obtain the optimal efficiency working curve γ 1 =f(γ2 ),like Figure 6 At this time, the η-P curves corresponding to all working points on the working curve are as follows: Figure 6 As shown in (b), when the output efficiency reaches 15% of the maximum output efficiency, the efficiency can reach more than 90%.

[0063] When the system is working, the voltage sampling + low-pass filtering module can obtain the DC side voltage of the inverter and rectifier in real time. The deviation signal obtained by subtracting the collected output voltage from the voltage reference value is processed and input into the PI control method in the DSP chip to obtain γ 2 Instructions. 2 Input to the optimal efficiency curve γ 1 =f(γ 2 ) is calculated to obtain the corresponding γ 1 . γ 1 , γ 2 And the collected input voltage V in The above four control variables are input into the ZVS constraint equation to solve for δ and f; the above four control variables are input into the PWM generation link to drive the switch action, and finally the whole control process is completed. Among them, the voltage closed loop effect is as follows Figure 7 As shown, it can be seen that the system can complete the voltage closed-loop regulation of 1V to 49V within an adjustment time of about 50ms; at the same time, the minimum reactive power ZVS of all switches can be achieved under steady-state conditions of different output voltages. The voltage and current waveforms of the inverter and rectifier AC side are as follows: Figure 8 shown.

[0064] It will be easily understood by those skilled in the art that the above description is only a preferred embodiment of the present invention and is not intended to limit the present invention. Any modifications, equivalent substitutions and improvements made within the spirit and principles of the present invention should be included in the protection scope of the present invention.

Claims

1. A three-phase shift frequency modulation control method suitable for an SS compensated wireless power transmission system, applied to an SS resonant compensation dual active bridge circuit, wherein the SS resonant compensation dual active bridge circuit comprises a DC voltage source, a phase-shifted full-bridge inverter, an SS resonant compensation network, a magnetic coupling mechanism, a phase-shifted full-bridge rectifier, a filter capacitor, and a load resistor; the phase-shifted full-bridge inverter comprises power MOS tubes S1 to S4, and the phase-shifted full-bridge rectifier comprises power MOS tubes S5 to S8; the magnetic coupling mechanism comprises a primary transmitting coil and a secondary receiving coil; the SS resonant compensation network comprises a primary capacitor C1 and a secondary capacitor C2; characterized in that: The three-phase shift frequency modulation control method comprises: By adjusting f and δ, the minimum reactive ZVS operating point of all power MOS tubes can be tracked; The optimal dcdc efficiency is achieved by constraining the numerical relationship between γ1 and γ2; Controllable output voltage gain can be achieved by adjusting γ1 and γ2; Among them, f is the switching frequency, δ is the phase angle when S5 is turned on ahead of S1, γ1 is the phase angle when S1 is turned on ahead of S4, and γ2 is the phase angle when S5 is turned on ahead of S8.

2. The three-phase shift frequency modulation control method according to claim 1, characterized in that: The method of tracking the minimum reactive ZVS operating point of all power MOS tubes by adjusting f and δ includes: Analyze the phase conditions that the voltage and current of all power MOS tubes should meet when they are at the minimum reactive ZVS operating point; Under the phase condition, a steady-state phasor model of the SS resonant compensation dual active bridge circuit is established; In combination with Kirchhoff's law, the circuit equation is solved by the steady-state phasor model to derive the phasor expressions of key state quantities of the circuit; According to the phasor expression and phase condition, the phasor diagram is drawn, and the constraints that f and δ should satisfy at the minimum reactive ZVS operating point are obtained according to the geometric relationship in the phasor diagram, so as to achieve the tracking of the minimum reactive ZVS operating point by adjusting f and δ in real time. Among them, f is the switching frequency, δ is the phase angle when S5 leads S1, γ1 is the phase angle when S1 leads S4, and γ2 is the phase angle when S5 leads S8.

3. The three-phase shift frequency modulation control method according to claim 1, characterized in that: The method of achieving the optimal dcdc efficiency by constraining the numerical relationship between γ1 and γ2 includes: Sweep γ1 and γ2 in the range of 0 to π to obtain the system output voltage V under different combinations of γ1 and γ2 dc and efficiency; Select a continuous monotonic interval and require δV dc / δγ1<0,δV dc / δγ2<0; Select the upper and lower limits of the output voltage and the incremental step size, and obtain the working curves between γ1 and γ2 at each output voltage; The combination of γ1 and γ2 with the best efficiency on the working curve under each output voltage is selected and plotted into a line, and the least squares linear fitting is performed to finally obtain the working curve γ1=f(γ2) to achieve the optimal dcdc efficiency.

4. The three-phase shift frequency modulation control method as claimed in claim 3, characterized in that: The controllable output voltage gain is achieved by adjusting γ1 and γ2, including: adjusting the output voltage V dc Perform closed-loop control, input the deviation signal into the PI controller output γ2, and then obtain γ1 by γ1=f(γ2), which in turn guides the generation of PWM drive signal and drives the switching action to achieve controllable output voltage gain.

5. A three-phase shift frequency modulation control system suitable for SS compensation wireless power transmission system, characterized in that: include: SS resonant compensation dual active bridge circuit, DSP chip, control circuit; The SS resonant compensation dual active bridge circuit includes a DC voltage source, a phase-shifted full-bridge inverter, an SS resonant compensation network, a magnetic coupling mechanism, a phase-shifted full-bridge rectifier, a filter capacitor, and a load resistor; The phase-shifted full-bridge inverter includes power MOS tubes S1-S4, and the phase-shifted full-bridge rectifier includes power MOS tubes S5-S8; the magnetic coupling mechanism includes a primary transmitting coil and a secondary receiving coil; the SS resonant compensation network includes a primary capacitor C1 and a secondary capacitor C2; The DSP chip is used to execute the three-phase shift frequency modulation control method, and receives the sampling signal and issues the control instruction through the control circuit to drive the switch device in the SS resonance compensation dual active bridge circuit to complete the power control; the three-phase shift frequency modulation control method includes: tracking the minimum reactive ZVS working point of all power MOS tubes by adjusting f and δ, achieving the optimal dcdc efficiency by constraining the numerical relationship between γ1 and γ2, and achieving the controllable output voltage gain by adjusting γ1 and γ2; wherein f is the switching frequency, δ is the phase angle of S5 leading S1, γ1 is the phase angle of S1 leading S4, and γ2 is the phase angle of S5 leading S8.

6. The three-phase shift frequency modulation control system according to claim 5, characterized in that: The power MOS tubes S1 to S4 are respectively provided with a body diode, and the cathode and anode of each body diode are respectively connected to the drain and source of the corresponding power MOS tube; the source of S1 is connected to the drain of S2, and the common terminal A of the two is the first AC output terminal, the drain of S1 is connected to the positive electrode of the DC voltage source, the source of S2 is connected to the negative electrode of the DC voltage source, and S1 and S2 form a bridge arm; the source of S3 is connected to the drain of S4, and the common terminal B of the two is the second AC output terminal, the drain of S3 is connected to the positive electrode of the DC power supply, the source of S4 is connected to the negative electrode of the DC power supply, and S3 and S4 form a bridge arm; The power MOS tubes S5 to S8 each have a body diode, and the cathode and anode of each body diode are connected to the drain and source of the corresponding MOS tube respectively; the source of S5 is connected to the drain of S6, and the common terminal a of the two is the first AC input terminal, and the drain of S5 is connected to the filter capacitor C R The positive terminal and one end of the load resistor are connected, and the source of S6 is connected to the filter capacitor C R The negative terminal and the other end of the load resistor are connected, S5 and S6 form a bridge arm; the source of S7 is connected to the drain of S8, and the common end b of the two is the second AC input terminal. The drain of S7 is connected to the filter capacitor C R The positive terminal and one end of the load resistor are connected, and the source of S8 is connected to the filter capacitor C R The negative terminal and the other end of the load resistor are connected, and S7 and S8 form a bridge arm; The first end of the primary capacitor C1 is connected to the first AC output terminal A, and the second end is connected to one end of the primary transmitting coil; the first end of the series capacitor C2 is connected to one end of the secondary transmitting coil, and the second end is connected to the first AC input terminal a.

7. The three-phase shift frequency modulation control system according to claim 5, characterized in that: The method of tracking the minimum reactive ZVS operating point of all power MOS tubes by adjusting f and δ includes: Analyze the phase conditions that the voltage and current of all power MOS tubes should meet when they are at the minimum reactive ZVS operating point; Under the phase condition, a steady-state phasor model of the SS resonant compensation dual active bridge circuit is established; In combination with Kirchhoff's law, the circuit equation is solved by the steady-state phasor model to derive the phasor expressions of key state quantities of the circuit; The phasor diagram is drawn according to the phasor expression and phase condition. The constraints that f and δ should satisfy at the minimum reactive ZVS operating point are obtained according to the geometric relationship in the phasor diagram. Thus, the tracking of the minimum reactive ZVS operating point under different working conditions can be achieved by adjusting f and δ in real time.

8. The three-phase shift frequency modulation control system according to claim 5, characterized in that: The method of achieving the optimal dcdc efficiency by constraining the numerical relationship between γ1 and γ2 includes: Sweep γ1 and γ2 in the range of 0 to π to obtain the system output voltage V under different combinations of γ1 and γ2 dc and efficiency; Select a continuous monotonic interval and require δV dc / δγ1<0,δV dc / δγ2<0; Select the upper and lower limits of the output voltage and the incremental step size, and obtain the working curves between γ1 and γ2 at each output voltage; The combination of γ1 and γ2 with the best efficiency on the working curve under each output voltage is selected and plotted into a line, and the least squares linear fitting is performed to finally obtain the working curve γ1=f(γ2) to achieve the optimal dcdc efficiency.

9. The wireless power transmission system based on the three-phase shift frequency modulation control method as claimed in claim 8, characterized in that: The controllable output voltage gain is achieved by adjusting γ1 and γ2, including: adjusting the output voltage V dc Perform closed-loop control, input the deviation signal into the PI controller output γ2, and then obtain γ1 by γ1=f(γ2), which in turn guides the generation of PWM drive signal and drives the switching action to achieve controllable output voltage gain.

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