A control system and method for a magnetic levitation motor and a magnetic levitation motor

Abstract

The present invention belongs to the field of motor control technology and discloses a control system, method, and magnetic levitation motor for a magnetic levitation motor. The control system includes: a linear extended state observer, a second-order generalized integrator, a phase-locked loop, a function calculator, a first subtractor, a first PI controller, a second subtractor, a third subtractor, a second PI controller, a third PI controller, an inverse Park converter, a space vector pulse width modulator, a Clarke converter, and a Park converter. The present invention effectively solves the problems of low speed and position estimation accuracy, limited system stability, and limited dynamic performance, achieving efficient and stable motor operation without sensors.

Description

Technical Field

[0001] The present invention belongs to the technical field of motor control, and in particular relates to a control system and method for a magnetic levitation motor and the magnetic levitation motor. Background Art

[0002] Among sensorless control methods, observer-based approaches have attracted considerable attention. The linear extended state observer (LESO) has shown promise in handling system disturbances and state estimation. The LESO can observe the total disturbance of the system as an extended state, thereby improving system robustness to a certain extent. Furthermore, the second-order generalized integrator (SOGI) has shown outstanding performance in signal processing, particularly in extracting specific frequency signals and suppressing disturbances.

[0003] However, existing linear extended state observers and second-order generalized integrators both have limitations when used alone. For example, the linear extended state observer may experience phase lag when observing back EMF, which can affect the accuracy of speed and position estimation. While the second-order generalized integrator can filter the signal, its ability to suppress DC components is limited under complex motor operating conditions, and it may not fully realize its advantages when used in conjunction with other control strategies. These issues hinder further improvements in motor control system performance.

[0004] The Chinese patent application, CN112713834A, is titled "A Sensorless Position Control Method and System for a Permanent Magnet Synchronous Motor." The method includes: establishing an extended state observer (ESO) for the permanent magnet synchronous motor, replacing the integrator in the ESO with a complex coefficient filter; obtaining speed observations based on the output of the ESO based on the complex coefficient filter; processing the speed observations using an integral prediction model to obtain a predicted speed estimate; calculating the center frequency of the complex coefficient filter based on the relationship between the predicted speed estimate and the frequency of the AC motor; feeding the center frequency back to the ESO for closed-loop control; and adaptively adjusting the frequency to ensure that the center frequency is consistent with the actual back-electromotive force frequency, thereby achieving low phase delay in rotor position estimation and obtaining accurate rotor position information, providing a better foundation for angle and speed estimation using the sensorless algorithm. However, the patent application fails to achieve precise control under various complex operating conditions. Summary of the Invention

[0005] In order to overcome the problems existing in the above-mentioned prior art, the purpose of the present invention is to provide a control system, method and magnetic levitation motor for a magnetic levitation motor. By combining an improved linear extended state observer with a second-order generalized integrator, the problems of low speed and position estimation accuracy, limited system stability and dynamic performance can be effectively solved, and efficient and stable operation of the motor can be achieved in a sensorless state.

[0006] To achieve the above object, the technical solution adopted by the present invention is:

[0007] In a first aspect, the present invention provides a control system for a magnetic levitation motor, comprising:

[0008] The linear extended state observer is used to input two stationary frame currents , and voltage 、 , output estimated back EMF 、 ; The linear extended state observer has a second-order generalized integrator;

[0009] Phase-locked loop, used to input the output of the linear extended state observer and output the speed estimate and the rotor position estimate ;

[0010] Function calculator for calculating speed estimates and The product of

[0011] The first subtractor is used to calculate the output result of the function calculator and the reference speed difference;

[0012] The first PI controller is used to input the result of the first subtractor and output a q-axis current reference value. ;

[0013] The second subtractor is used to calculate the q-axis current reference value The difference between the d-axis current id;

[0014] a third subtractor, configured to calculate a difference between the q-axis current iq and the d-axis reference current idref;

[0015] a second PI controller, configured to input a result of the second subtractor and output a q-axis voltage Uq;

[0016] a third PI controller, configured to input a result of the third subtractor and output a d-axis voltage Ud;

[0017] Inverse Park transformer, which takes as input the q-axis voltage Uq, the d-axis voltage Ud, and the estimated rotor position , output voltage of two-phase stationary coordinate system 、 ;

[0018] Space vector pulse width modulator, used to input voltage in two-phase stationary coordinate system 、 , output three-phase voltage 、 、 To magnetic levitation motor;

[0019] Clarke converter is used to convert the three-phase current of the magnetic levitation motor 、 、 Converted to the current in the two-phase stationary coordinate system , ;

[0020] Park converter, used to input current in a two-phase stationary coordinate system , and the rotor position estimate , output d-axis current id and q-axis current iq.

[0021] Optionally, output angle With input angle The transfer function between them is:

[0022]

[0023] in, and is the parameter of the PI link in the phase-locked loop PLL; s is the complex frequency variable.

[0024] Optionally, the second-order generalized integrator includes:

[0025] The internal first subtractor is used to calculate the input signal With feedback filter output signal The difference between

[0026] A gain device, used for gaining the result of the internal first subtractor;

[0027] An internal second subtractor for calculating the difference between the gain result and the gain result after a specific delay or phase shift process;

[0028] The first integrator is used to perform integration operation on the processed signal and output the filtered output signal ;

[0029] Quadrature signal generator for input, filtered and output signals , output quadrature signal .

[0030] Optionally, the second-order generalized integrator further includes: an internal frequency adjuster, configured to dynamically adapt the frequency of a signal input to the integrator according to a real-time frequency condition of the motor operation.

[0031] Optionally, the transfer function of the order generalized integrator is:

[0032]

[0033]

[0034] In the formula is the gain coefficient of the second-order generalized integrator SOGI, is the center frequency of the second-order generalized integrator SOGI, which is adjusted according to the operating frequency range of the motor and the interference frequency that needs to be suppressed and The value of .

[0035] Optionally, the phase-locked loop includes: a first phase-locked loop multiplier for calculating Shaft electromotive force component The product with the cosine function;

[0036] The second multiplier of the phase-locked loop is used to calculate the β-axis electromotive force component Multiplication with the sine function;

[0037] The first subtractor of the phase-locked loop is used to calculate the difference between the results of the first multiplier of the phase-locked loop and the second multiplier of the phase-locked loop to obtain the phase difference ;

[0038] Phase-locked loop function calculator for inputting phase difference , output the normalized phase difference result;

[0039] Phase-locked loop PI controller, used to input the result of the phase-locked loop function calculator and output the control signal and phase estimate .

[0040] Optionally, the phase-locked loop further includes an adder for adding the phase estimation value With the compensated phase estimate Add together to get the phase estimate .

[0041] In a second aspect, the present invention provides a method for using a control system of a magnetic levitation motor, comprising the following steps:

[0042] Input two stationary coordinate system currents , and voltage 、 , output estimated back EMF 、 ; To the linear extended state observer, output estimated back electromotive force 、 ;

[0043] Input the output of the linear extended state observer to the phase-locked loop and output the speed estimate and the rotor position estimate ;

[0044] Calculate the estimated speed using the function calculator and The product of

[0045] The output result of the function calculator and the reference speed are calculated by the first subtractor difference;

[0046] Input the result of the first subtractor to the first PI controller and output the q-axis current reference value ;

[0047] The q-axis current reference value is calculated by the second subtractor The difference between the d-axis current id;

[0048] Calculating the difference between the q-axis current iq and the d-axis reference current idref by a third subtractor;

[0049] The result of the second subtractor is input to the second PI controller, and the q-axis voltage Uq is output;

[0050] The result of the third subtractor is input to the third PI controller, and the d-axis voltage Ud is output;

[0051] Input q-axis voltage Uq, d-axis voltage Ud and rotor position estimation value To the inverse Park converter, output voltage in the two-phase stationary coordinate system 、 ;

[0052] Input voltage in two-phase stationary coordinate system 、 To the space vector pulse width modulator, output three-phase voltage 、 、 Control magnetic levitation motor;

[0053] The three-phase current of the magnetic levitation motor is converted by the Clarke converter 、 、 Converted to the current in the two-phase stationary coordinate system , ;

[0054] Input current in the two-phase stationary coordinate system , and the rotor position estimate To the Park converter, output the d-axis current id to the second subtractor, and output the q-axis current iq to the third subtractor, completing the cycle of the closed-loop system.

[0055] In a third aspect, the present invention provides a magnetic levitation motor, including the control system of the magnetic levitation motor.

[0056] Optionally, the magnetic levitation motor further includes a drive circuit, and the drive circuit is electrically connected to the magnetic levitation motor.

[0057] Compared with the prior art, the present invention has the following beneficial effects:

[0058] This invention combines an improved linear extended state observer (LESO) with a second-order generalized integrator (GII). By rationally introducing extended states and optimizing parameter configuration, the LESSO effectively reduces errors such as phase lag and improves the accuracy of back-EMF observation. The GII further optimizes signal quality with its excellent frequency selectivity. The synergistic effect of these two methods, combined with the precise extraction function of the normalized phase-locked loop (NPL), significantly improves the estimation accuracy of motor speed and position.

[0059] The present invention can avoid the increase in volume, mass and cost of the motor control system caused by installing mechanical physical sensors, and can avoid installation errors existing in mechanical physical sensors.

[0060] This invention allows for flexible adjustment of control strategies based on the motor's varying speed ranges and load conditions. By rationally adjusting the parameters of the second-order generalized integrator (damping coefficient and resonant frequency) and improving the gain matrix parameters of the linear extended state observer, the system can adapt to a variety of complex operating conditions. BRIEF DESCRIPTION OF THE DRAWINGS

[0061] The drawings described herein are for illustrative purposes only and are not intended to limit the scope of the present invention in any way. In addition, the shapes and proportional dimensions of the components in the drawings are only schematic and are used to help understand the present invention, and are not intended to specifically limit the shapes and proportional dimensions of the components of the present invention. In the drawings:

[0062] Figure 1 It is a principle block diagram of the method of the present invention.

[0063] Figure 2 It is the block diagram of the second-order generalized integrator SOGI structure.

[0064] Figure 3 It is a normalized phase-locked loop structure diagram.

[0065] Figure 4 This is the angle signal waveform of the linear extended state observer LESO and phase-locked loop PLL algorithm.

[0066] Figure 5 This is the waveform of the angle observation error of the linear extended state observer LESO and phase-locked loop PLL algorithm.

[0067] Figure 6 It is the speed signal waveform of the linear extended state observer LESO and phase-locked loop PLL algorithm.

[0068] Figure 7 This is the speed observation error waveform of the linear extended state observer LESO and phase-locked loop PLL algorithm.

[0069] Figure 8 This is the angle signal waveform diagram of the linear extended state observer LESO, phase-locked loop PLL and second-order generalized integrator SOGI algorithm.

[0070] Figure 9 It is the angle observation error of the linear extended state observer LESO, phase-locked loop PLL and second-order generalized integrator SOGI algorithm.

[0071] Figure 10 This is the speed signal waveform diagram of the linear extended state observer LESO, phase-locked loop PLL and second-order generalized integrator SOGI algorithm.

[0072] Figure 11 This is the speed observation error waveform of the linear extended state observer LESO, phase-locked loop PLL and second-order generalized integrator SOGI algorithm. DETAILED DESCRIPTION

[0073] In order to enable those skilled in the art to better understand the technical solutions of the present invention, the technical solutions in the embodiments of the present invention will be clearly and completely described below in conjunction with the accompanying drawings of the embodiments of the present invention. Obviously, the embodiments described are only part of the embodiments of the present invention, not all of the embodiments. Based on the embodiments of the present invention, all other embodiments obtained by ordinary technicians in this field without making creative efforts should fall within the scope of protection of the present invention.

[0074] Therefore, the following detailed description of the embodiments of the present invention provided in the accompanying drawings is not intended to limit the scope of the invention as claimed, but rather merely represents selected embodiments of the present invention. All other embodiments derived by persons of ordinary skill in the art based on the embodiments of the present invention without creative effort shall fall within the scope of protection of the present invention.

[0075] In the description of the embodiments of the present invention, it should be noted that if the terms "upper", "lower", "horizontal", "inner", etc. appear, the orientation or position relationship indicated is based on the orientation or position relationship shown in the accompanying drawings, or is the orientation or position relationship in which the product of the invention is usually placed when in use. It is only for the convenience of describing the present invention and simplifying the description, and does not indicate or imply that the device or element referred to must have a specific orientation, be constructed and operated in a specific orientation. Therefore, it cannot be understood as a limitation on the present invention.

[0076] When an element is referred to as being "disposed on" another element, it may be directly on the other element or there may also be an intermediate element. When an element is considered to be "connected to" another element, it may be directly connected to the other element or there may also be an intermediate element. The terms "vertical", "horizontal", "left", "right" and similar expressions used herein are for illustrative purposes only and are not intended to be the only embodiments. If the term "horizontal" appears, it does not mean that the component is required to be absolutely horizontal, but it can be slightly tilted. For example, "horizontal" only means that its direction is more horizontal than "vertical", and it does not mean that the structure must be completely horizontal, but it can be slightly tilted.

[0077] It should be noted that similar reference numerals and letters denote similar items in the following figures. Therefore, once an item is defined in one figure, it does not need to be further defined or explained in the subsequent figures. In the description of the present invention, it should be understood that the terms "comprises" and "comprising" indicate the presence of the described features, integers, steps, operations, elements and / or components, but do not exclude the presence or addition of one or more other features, integers, steps, operations, elements, components and / or their collections.

[0078] Unless otherwise defined, all technical and scientific terms used herein have the same meaning as commonly understood by those skilled in the art to which the present invention pertains. The terms used herein in the specification of the present invention are for the purpose of describing specific embodiments only and are not intended to limit the present invention. As used in the specification of the present invention and the appended claims, the singular forms "a," "an," and "the" are intended to include the plural forms unless the context clearly indicates otherwise.

[0079] The present invention will be described in detail below with reference to the accompanying drawings.

[0080] A control system for a magnetic levitation motor includes: a drive circuit, the drive circuit being used to provide power to the magnetic levitation motor;

[0081] Park converter is used for Park transformation, which is a linear transformation that transforms the physical quantity in the two-phase stationary coordinate system (αβ coordinate system) into the two-phase rotating coordinate system (dq coordinate system). , becomes , .

[0082] Figure 1 The system architecture of sensorless control of permanent magnet synchronous motor (PMSM) is presented. and the converted speed estimate (Depend on The output is used as the reference value of the q-axis current. , and there is a d-axis current reference value . 、 and the actual q-axis current , d-axis current Compare and the difference is output by each PI controller 、 , the voltage in the two-phase stationary coordinate system is obtained by inverse Park transformation 、 , and then sent to the space vector pulse width modulation module to generate three-phase voltage 、 、 To drive the motor. Motor three-phase current 、 、 After Clarke transformation, 、 , and then get it through Park transformation 、 Used for current loop feedback. The Linear Extended State Observer (LESO) receives the voltage and current in the two-phase stationary coordinate system, estimates the motor state and outputs 、 To the Phase-Locked Loop (PLL), the PLL outputs the speed estimate and the rotor position estimate , Feedback is provided to the Park transform and inverse Park transform to implement coordinate transformation and control algorithm. This system realizes sensorless control of permanent magnet synchronous motor (PMSM) through closed-loop structure and coordinated operation of various modules.

[0083] Figure 2The second-order generalized integrator (SOGI) is connected in parallel to the observation parameters of the linear extended state observer. It has excellent filtering characteristics near the resonant frequency, effectively filtering out harmonics and noise in the input signal and extracting the fundamental signal or signal components of specific frequencies. This improves the observation accuracy of the linear extended state observer.

[0084] The second-order generalized integrator comprises:

[0085] The internal first subtractor is used to calculate the input signal With feedback filter output signal The difference between the input signal The filtered output signal fed back Subtract the error signal .

[0086] Gainer, used to gain the result of the internal first subtractor. Error signal Gain After zooming in, enter the integration phase.

[0087] The internal second subtractor is used to calculate the difference between the gain filter result and the gain filter result after a specific delay or phase shift.

[0088] The internal frequency regulator is used to dynamically adapt the frequency of the signal input to the integrator according to the real-time frequency condition of the motor.

[0089] The first integrator is used to perform integration operation on the processed signal and output the filtered output signal .

[0090] Quadrature signal generator for input, filtered and output signals , output quadrature signal The orthogonal signal generator is implemented using a trigonometric function-based orthogonal signal generator. This generator uses the phase information and amplitude information of the input signal after specific processing to generate a signal that is orthogonal to the filtered output signal based on the orthogonality principle of trigonometric functions.

[0091] One of the integral links outputs a filtered output signal , another integral link and the resonant frequency After multiplication, Subtract the orthogonal signal of the filtered output signal , and It is then fed back to the previous operation link to form a closed loop.

[0092] The final output is The estimated voltage signal obtained after integration and other operations can track the input voltage signal , and has good filtering and orthogonal signal generation characteristics at specific frequencies.

[0093] Among them, the transfer function of the second-order generalized integrator SOGI is:

[0094]

[0095]

[0096] In the formula is the gain coefficient of the second-order generalized integrator SOGI, is the center frequency of the second-order generalized integrator SOGI, which is adjusted according to the operating frequency range of the motor and the interference frequency that needs to be suppressed and The two functions work together to process the input signal. One transfer function calculates the filtered output signal, while the other calculates its quadrature signal. In actual operation, these two transfer functions work together to enable the second-order generalized integrator SOGI to effectively extract specific frequency components near the resonant frequency and suppress interfering signals.

[0097] Figure 3 is the normalized phase-locked loop, which is observed by the linear extended state observer LESO. and , estimate the motor speed and rotor position .

[0098] The phase-locked loop includes: a first phase-locked loop multiplier for calculating Shaft electromotive force component The product with the cosine function;

[0099] The second multiplier of the phase-locked loop is used to calculate the β-axis electromotive force component Multiplication with the sine function;

[0100] The first subtractor of the phase-locked loop is used to calculate the difference between the results of the first multiplier of the phase-locked loop and the second multiplier of the phase-locked loop to obtain the phase difference ;

[0101] Phase-locked loop function calculator for inputting phase difference , bring in ;

[0102] is the synthetic amplitude of the back electromotive force in the α-β stationary coordinate system, and the phase difference It can reflect the difference between the actual and estimated motor rotor position. Combining the two can be used to calculate motor speed and position estimation. This is a common operation in some model-based or algorithm-based position sensorless control strategies to optimize control performance.

[0103] Phase-locked loop PI controller, used to input the result of the phase-locked loop function calculator and output the control signal ;

[0104] Adder for adding the phase estimate With the compensated phase estimate Add together to get the phase estimate .

[0105] First, the input signal and Multiplying the cosine (cos) and sine (sin) functions respectively is to obtain the component information of the input signal in the orthogonal coordinate system. Subtracting the two product results will give the phase difference .

[0106] The cosine function (cos) refers to the cosine value based on the motor rotor position angle. During the operation of the motor, the rotor position will change with time. Let the electrical angle corresponding to the rotor position be , then the cosine function is In phase-locked loop calculations, Shaft electromotive force component and Multiply, Shaft electromotive force component and (sine function, corresponding to cosine function, based on the same electrical angle) multiplication, through subsequent operations, can achieve the control function of estimating motor speed and rotor position. It is related to the motor's speed and operating status. It is a dynamically changing value that is continuously updated and calculated through control algorithms such as phase-locked loops.

[0107] After a denominator containing The calculation link. Using the amplitude normalization method, let the component of the signal to be processed in the α-β coordinate system be and , the calculation formula is , ,in and is the normalized signal component. Through this operation, the signal amplitude is associated with the composite amplitude of the back EMF in the coordinate system α-β to achieve normalization. This step plays a normalization role, making subsequent processing more stable and accurate.

[0108] The normalized signal is fed into the proportional integral (PI) controller, which is one of the core parts of the entire system. It can output a control signal based on the input error signal (here is the processed phase difference information). .

[0109] It has a dual role. On the one hand, it outputs the frequency estimate , used to reflect the frequency information of the input signal; on the other hand, The phase estimation value is obtained after the integration link , this phase estimate is a real-time estimate of the input signal phase. First, the back electromotive force observation value in the stationary α-β axis coordinate system is obtained by the linear extended state observer. and And input into the phase-locked loop; the phase-locked loop multiplies it with the internal orthogonal signal through a multiplier, and a signal containing phase difference information is obtained through trigonometric operation, and then the control signal is output through the PI controller proportional-integral operation to obtain the angular velocity estimate. ; In discrete systems, such as Euler integration method ( ) and other integral operations, thereby continuously updating the phase estimation value , realizing real-time estimation of the input signal phase.

[0110] In order to further improve the accuracy of phase estimation, It will also be compared with the compensated phase estimate after phase compensation. Add together to get the final phase estimate .

[0111] Example 1

[0112] Set the initial motor speed and rotor position estimates: Construct a mathematical model of the permanent magnet synchronous motor in a stationary αβ coordinate system, taking into account the symmetry of the motor's three-phase stator windings, the standard sinusoidal waveform of the stator-side induced back EMF during rotation, and ignoring core saturation.

[0113] Extract specific frequency components from the current and suppress interference: Design a second-order generalized integrator SOGI, whose input is the current signal in the αβ coordinate system;

[0114] The state variables and total disturbance of the motor are estimated, and its state equation is:

[0115]

[0116]

[0117] in, 、 、 is the system matrix, is the observer gain matrix, is the actual measured output current, To estimate the output current, the output of the second-order generalized integrator SOGI is used as an input of the observer to participate in the estimation of the motor state variables and the total disturbance.

[0118] According to the estimation results of the improved linear extended state observer, the estimated values ​​of the motor speed and rotor position are calculated. The speed and position information are obtained by estimating the back electromotive force using the relationship between the motor's back electromotive force and the speed and position.

[0119] The speed estimation value is compared with the given speed to obtain the speed error, which is then adjusted using a PI controller to output a control voltage.

[0120] According to the output of the speed controller and the mathematical model of the motor, the current reference value is calculated, and the actual measured current is compared with the current reference value to obtain the current error. The current error is adjusted using a PI controller, and the d-axis and q-axis components of the control voltage are output to achieve closed-loop control of the motor current.

[0121] The control voltage output by the current controller is converted into a space vector pulse width modulation (SVPWM) signal, which drives the inverter to output the corresponding voltage to the permanent magnet synchronous motor, achieving motor control. During the SVPWM signal generation process, the motor state estimated by the observer and the required control voltage are combined with the current signal processing results of the second-order generalized integrator (SOGI) to more accurately calculate the SVPWM reference voltage vector, improving voltage utilization and the smoothness of the motor output torque. A flexible SVPWM modulation strategy switching mechanism is designed to achieve sensorless speed control of the permanent magnet synchronous motor based on different motor operating conditions and control requirements.

[0122] In the sensorless control method for a permanent magnet synchronous motor based on an improved linear extended state observer incorporating a second-order generalized integrator (SOGI), the method for setting the initial motor speed and rotor position estimation values ​​is as follows:

[0123] The stator voltage equation in the model is:

[0124] Assuming that the three-phase stator winding of the motor is symmetrical, the waveform of the induced back electromotive force on the stator side is a standard sine wave when the motor rotates, and the core saturation is ignored, then the PMSM is stationary. , The stator voltage equation in the axis coordinate system is:

[0125]

[0126] In the formula 、 、 、 、 、 They are Voltage, current and back electromotive force in the coordinate system, is the stator phase resistance, is the stator phase inductance. Voltage, current and back electromotive force in the coordinate system, is the stator phase resistance, is the stator phase inductance;

[0127] The back electromotive force induced by the permanent magnet when the motor rotates is:

[0128]

[0129] In the formula is the permanent magnet flux, is the motor electrical angular velocity, For the rotor Extreme and Angle between the phase axes; , The two voltage equations are decoupled from each other in the coordinate system, and the two voltage equations are exactly the same except for the subscripts. The shaft voltage equation is used as an example. The back electromotive force is regarded as an unknown quantity and the expansion state is introduced:

[0130]

[0131] Initialize the system according to the rated parameters of the motor and set the initial motor speed and rotor position estimation values.

[0132] The sensorless control method for a permanent magnet synchronous motor based on an improved linear extended state observer incorporating a second-order generalized integrator (SOGI) is described. The method for designing the second-order generalized integrator (SOGI) is as follows:

[0133] is the input signal, and are the filtered output signal and its orthogonal signal respectively, and the corresponding transfer function is the transfer function of the second-order generalized integrator SOGI:

[0134]

[0135]

[0136] Where, is the gain coefficient of the second-order generalized integrator SOGI, is the center frequency of the second-order generalized integrator SOGI. When the system is in steady state, let the Laplace operator , it can be written as:

[0137]

[0138] Adjust according to the operating frequency range of the motor and the interference frequency that needs to be suppressed and The value of makes the second-order generalized integrator SOGI have good selectivity for signals of specific frequencies. Adjust according to the operating frequency range of the motor and the interference frequency that needs to be suppressed and The value of makes the second-order generalized integrator SOGI have good selectivity for signals of specific frequencies.

[0139] The sensorless control method for a permanent magnet synchronous motor based on an improved linear extended state observer incorporating a second-order generalized integrator is described as follows:

[0140] The state variables and total disturbance of the motor are estimated, and its state equation is:

[0141]

[0142]

[0143] In the formula 、 、 is the system matrix, is the observer gain matrix, is the actual measured output current, To estimate the output current, the output of the second-order generalized integrator SOGI is used as an input of the observer to participate in the estimation of the motor state variables and the total disturbance;

[0144] for The voltage equation in the coordinate system treats the back electromotive force as an unknown quantity and introduces an expansion state:

[0145]

[0146] In the formula for The shaft electromotive force component, and its derivative is .

[0147] According to formula 2, for The shaft electromotive force component, and its derivative is , construct the linear extended state observer LESO as shown in the formula:

[0148]

[0149] In the formula for Observed shaft current, 、 are the observer parameters, is the difference between the observed current and the actual current. The observer of the axis is configured by the frequency domain method. The 3w method is used to obtain the formula:

[0150] (in >0)

[0151] The 3W method is a method for configuring the parameters of a linearly expanded state observer. Within the framework of the frequency-domain method for configuring observer parameters, the 3W method selects an appropriate bandwidth value and utilizes frequency-domain characteristics and related mathematical derivations to determine the observer gain matrix. This ensures that the observer has good dynamic response performance while ensuring system stability.

[0152] The sensorless control method for a permanent magnet synchronous motor based on an improved linear extended state observer incorporating a second-order generalized integrator (SOGI) is described. The specific method for calculating the estimated values ​​of the motor speed and rotor position is as follows:

[0153] The normalized phase-locked loop (PLL) is used to extract the speed and position information from the back electromotive force observation value, which includes three links: phase detector, loop filter and voltage-controlled oscillator. The phase detector detects the phase difference between the input signal and the output signal. The mathematical expression of the back electromotive force in the coordinate system can be obtained as follows:

[0154]

[0155] In the formula is the position observation value output by the phase-locked loop PLL;

[0156] when , , that is,

[0157]

[0158] in ;

[0159] The loop filter is designed using the PI link. The position error is used to obtain the speed observation value through the PI link. The speed observation value is integrated to obtain the position observation value, and the angle error is derived. With input angle The transfer function between them is as follows:

[0160]

[0161] and It is the parameter of the PI link in the phase-locked loop PLL.

[0162] When the speed enters a steady state and is a fixed value, the rotor position signal is a ramp function. At this time, the steady-state error of the phase-locked loop PLL position observation is:

[0163]

[0164] Therefore, when the speed is constant, the phase-locked loop PLL designed above can achieve zero-error tracking of the rotor position;

[0165] After improving the normalization process, derive the output angle With input angle The transfer function between them is:

[0166]

[0167] The formula has low-pass characteristics and is reasonably configured and , smooth and accurate position estimation information can be obtained.

[0168] The method for implementing the output control voltage of the permanent magnet synchronous motor sensorless control method based on the improved linear extended state observer incorporating the second-order generalized integrator (SOGI) is as follows:

[0169] Compare the estimated speed with the given speed to obtain the speed error;

[0170] The PI controller is used to adjust the speed error and output the control voltage. The output formula of the speed controller is:

[0171]

[0172] In the formula For control voltage Axis component, For a given speed, is the proportionality coefficient, is the integration coefficient.

[0173] The sensorless control method for a permanent magnet synchronous motor based on an improved linear extended state observer incorporating a second-order generalized integrator (SOGI) realizes closed-loop control of the motor current as follows:

[0174] According to the output of the speed controller and the mathematical model of the motor, the current reference value is calculated, and the actual measured current is compared with the current reference value to obtain the current error;

[0175] The PI controller is used to adjust the current error and output the control voltage. Axis and axis component, the output formula of the current controller is

[0176]

[0177] For control voltage Axis component, 、 The current reference value Axis and Axis component, 、 The actual measured current Axis and Axis component, 、 、 、 are the corresponding proportional coefficient and integral coefficient, thereby realizing closed-loop control of the motor current.

[0178] The sensorless control method for a permanent magnet synchronous motor based on an improved linear extended state observer incorporating a second-order generalized integrator (SOGI) is implemented as follows:

[0179] Convert the control voltage output by the current controller into a space vector pulse width modulation (SVPWM) signal, drive the inverter to output the corresponding voltage to the permanent magnet synchronous motor, and realize the control of the motor;

[0180] In the SVPWM signal generation process, the motor state estimated by the observer and the required control voltage are combined with the current signal processing results of the second-order generalized integrator SOGI to more accurately calculate the SVPWM reference voltage vector;

[0181] According to different motor operating conditions and control requirements, a flexible SVPWM modulation strategy switching mechanism is designed to realize sensorless control of permanent magnet synchronous motor.

[0182] Example 2

[0183] 1) Magnetic levitation motor model and FOC system simulation

[0184] Use Simulink module to build magnetic levitation motor vector control system, and estimate the speed based on the square wave injection method of FOC system. The simulation construction diagram is as follows Figure 4This design is based on rotor flux oriented vector control. First, the correctness of the FOC system should be ensured, and then the observer model is built on this basis. The motor used in this design is rated at 3000rad / min, rated voltage U=380V, and rated frequency =50Hz, the specific magnetic levitation motor simulation parameters are shown in Table 1:

[0185] Table 1

[0186]

[0187] The parameters are adjusted according to the principle of adjusting the inner loop first and then the outer loop. The final parameter tuning results of the speed loop and current loop are shown in Table 2.

[0188] Table 2

[0189]

[0190] During the motor operation, the given speed is 10rad / s, that is, 95r / min, and the motor runs at low speed. Figure 5 The encoder's measured speed matches the set speed after 0.2 seconds, with a steady-state error of zero and excellent static and dynamic performance, verifying the feasibility of the FOC system. The simulation results demonstrate that the rotor-field-oriented vector control system model is accurate and parameter-matched. This model can be used to simulate and verify the high-frequency square wave injection speed estimation scheme.

[0191] 2) Simulation of sensorless control method for permanent magnet synchronous motor based on improved linear extended state observer incorporating second-order generalized integrator (SOGI)

[0192] First, it is verified that the linear extended state observer (LESO) and the phase-locked loop (PLL) can accurately observe the motor position and speed signals. The target speed is set to 750 r / min. At 0.1 s, the target speed suddenly changes to 1000 r / min. The motor runs at no load, and the parameter ω0 of the linear extended state observer is 5000.

[0193] After simulation verification, the system can operate normally. The specific analysis of the simulation waveform is as follows:

[0194] The angle signal obtained by the linear extended state observer LESO and the phase-locked loop PLL control algorithm is as follows Figure 4 As shown, the angle observation error is Figure 5 shown.

[0195] Figure 4The blue waveform in the middle represents the actual angle, and the red waveform represents the observed angle. During the initial phase (0-0.05 seconds), the observed angle (red waveform) rapidly rises and approaches the actual angle (blue waveform), demonstrating the system's rapid response and ability to accurately track the actual angle within a short period of time. Although the observed and actual angles deviate at certain moments, the overall trend shows consistent fluctuation frequency and general direction. This indicates that the observed angle closely reflects the changing trends of the actual angle, ensuring the long-term stability of the system.

[0196] Depend on Figure 5 As can be seen, the angle observation error remained within a small range, close to zero, for most of the time. This demonstrates that the system is able to observe angles very accurately during normal operation, with minimal deviation between the observed and actual angles, demonstrating the system's high precision. Although significant fluctuations occurred at specific time points (such as around 0.05 seconds and 0.15 seconds), these fluctuations were brief, and the system was able to quickly recover to a low-error state. This demonstrates that the system has a certain degree of self-regulation and control over errors, and can quickly recover stability after disturbances.

[0197] In summary, the linear extended state observer and phase-locked loop (PLL) model exhibits advantages such as fast convergence, high precision, and controllable error in angle observation. These advantages make the model have good application value in fields such as motor control.

[0198] The speed signal obtained by the linear extended state observer LESO and the phase-locked loop PLL control algorithm is as follows Figure 6 As shown in the figure, the speed observation error is as follows: Figure 7 shown.

[0199] Depend on Figure 6 It can be seen that during the initial phase (0-0.1s), both the actual speed (blue waveform) and the observed speed (red waveform) remain stable at around 300 r / min. This demonstrates that, in the initial state, the linear extended state observer and phase-locked loop (PLL) control algorithm enable the observed speed to track the actual speed well. The target speed (yellow waveform) is also 300 r / min during this phase, indicating that the system is operating in a stable state. At 0.1s, the target speed (yellow waveform) suddenly changes to 1000 r / min. Both the actual speed (blue waveform) and the observed speed (red waveform) quickly respond to this change and begin to rise. This demonstrates that the linear extended state observer and phase-locked loop (PLL) control algorithm have good dynamic response capabilities and can quickly track changes in the target speed.

[0200] At 0.2 seconds, a load is added. Despite the addition of the load, both the actual speed (blue waveform) and the observed speed (red waveform) remain near the target speed (1000 r / min). This demonstrates that the control algorithm has strong resistance to load disturbances and can maintain speed stability despite load changes.

[0201] Depend on Figure 7 As can be seen, the speed observation error (red waveform) remains within a small range for most of the entire process. Although the observation error briefly increases during the sudden speed change (0.1s) and load addition (0.2s), the system is able to quickly adjust, reducing the error and maintaining it at a low level. This demonstrates that the linear extended state observer and phase-locked loop (PLL) control algorithm are not only able to quickly respond to speed changes and load disturbances, but also maintain high observation accuracy during these dynamic processes.

[0202] In summary, the linear extended state observer and phase-locked loop (PLL) control algorithm can rapidly adjust the actual and observed speeds when the target speed changes suddenly, allowing them to quickly track the target speed. When a load is applied, the actual and observed speeds remain stable, ensuring system robustness. Throughout operation, especially during dynamic changes, the speed observation error is kept small, ensuring observation accuracy.

[0203] Next, a second-order generalized integrator (SOGI) is incorporated into the linear extended state observer LESO and the phase-locked loop PLL control algorithm, and the resulting angle signal is as follows: Figure 8 As shown, the angle observation error is Figure 9 As shown, the speed signal obtained is Figure 10 As shown in the figure, the speed observation error is as follows: Figure 11 shown.

[0204] Depend on Figure 8 As can be seen, before the improvement, there was a significant deviation between the observed angle and the actual angle at certain moments. After the improvement, the red and blue waveforms are much closer, indicating that the observed angle tracks the actual angle more accurately. The overall waveform fluctuations are also more stable, demonstrating the improved stability of the system in angle observation.

[0205] Depend on Figure 9 It can be seen that before the improvement, the angle observation error would fluctuate greatly at certain moments. After the improvement, the error waveform is more stable and remains at a low level most of the time, indicating that the improved algorithm has significantly improved the angle observation accuracy.

[0206] Depend on Figure 10As can be seen, during the sudden speed change (0.1s) and load addition (0.2s), while the actual and observed speeds before the improvement responded to the changes, there was still some deviation at the moment of the sudden change. After the improvement, the blue and red waveforms are more closely aligned at these key nodes, demonstrating that the system responds more quickly and accurately to speed changes. Throughout the entire process, the speed signal fluctuates less, enhancing system stability.

[0207] Depend on Figure 11 It can be seen that before the improvement, the speed observation error would fluctuate significantly during dynamic changes. After the improvement, the error waveform remained at a low level throughout the process, especially during sudden speed changes and load addition, where the peak error was significantly reduced, indicating that the system's anti-interference ability and observation accuracy have been improved.

[0208] In summary, the linear extended state observer and phase-locked loop (PLL) control algorithm, incorporating a second-order generalized integrator (SOGI), significantly reduces errors in both angle and speed observations, enabling the system to more accurately track actual values. During dynamic changes (such as sudden speed changes and load additions), the system exhibits less fluctuation and recovers to a stable state more quickly. In the face of external disturbances (such as load changes), the system is able to better maintain observation accuracy and reduce errors.

[0209] Unless otherwise specified, the device components involved in the above embodiments are all conventional device components, and the structural settings, working modes or control modes involved are all conventional settings, working modes or control modes in the art unless otherwise specified.

[0210] Finally, it should be noted that the above embodiments are only used to illustrate the technical solution of the present invention and are not limiting. Other modifications or equivalent substitutions made to the technical solution of the present invention by ordinary technicians in this field should be included in the scope of the claims of the present invention as long as they do not depart from the spirit and scope of the technical solution of the present invention.

Claims

1. A control system for a magnetic levitation motor, characterized in that: include: The linear extended state observer is used to input two stationary frame currents , and voltage 、 , output estimated back EMF 、 ; The linear extended state observer has a second-order generalized integrator; Phase-locked loop, used to input the output of the linear extended state observer and output the speed estimate and the rotor position estimate ; Function calculator for calculating speed estimates and The product of The first subtractor is used to calculate the output result of the function calculator and the reference speed difference; The first PI controller is used to input the result of the first subtractor and output a q-axis current reference value. ; The second subtractor is used to calculate the q-axis current reference value The difference between the d-axis current id; a third subtractor, configured to calculate a difference between the q-axis current iq and the d-axis reference current idref; a second PI controller, configured to input a result of the second subtractor and output a q-axis voltage Uq; a third PI controller, configured to input a result of the third subtractor and output a d-axis voltage Ud; Inverse Park transformer, which takes as input the q-axis voltage Uq, the d-axis voltage Ud, and the estimated rotor position , output voltage of two-phase stationary coordinate system 、 ; Space vector pulse width modulator, used to input voltage in two-phase stationary coordinate system 、 , output three-phase voltage 、 、 To magnetic levitation motor; Clarke converter is used to convert the three-phase current of the magnetic levitation motor 、 、 Converted to the current in the two-phase stationary coordinate system , ; Park converter, used to input current in a two-phase stationary coordinate system , and the rotor position estimate , output d-axis current id and q-axis current iq; The phase-locked loop includes: a first phase-locked loop multiplier for calculating Shaft electromotive force component The product with the cosine function; The second multiplier of the phase-locked loop is used to calculate the β-axis electromotive force component Multiplication with the sine function; The first subtractor of the phase-locked loop is used to calculate the difference between the results of the first multiplier of the phase-locked loop and the second multiplier of the phase-locked loop to obtain the phase difference ; Phase-locked loop function calculator for inputting phase difference , output the normalized phase difference result; Phase-locked loop PI controller, used to input the result of the phase-locked loop function calculator and output the control signal and phase estimate ; Output angle With input angle The transfer function between is: in, and is the parameter of the PI link in the phase-locked loop PLL; s is the complex frequency variable.

2. The control system of a magnetic levitation motor according to claim 1, characterized in that: The second-order generalized integrator comprises: The internal first subtractor is used to calculate the input signal With feedback filter output signal The difference between A gain device, used for gaining the result of the internal first subtractor; An internal second subtractor for calculating the difference between the gain result and the gain result after a specific delay or phase shift process; The first integrator is used to perform integration operation on the processed signal and output the filtered output signal ; Quadrature signal generator for input, filtered and output signals , output quadrature signal .

3. The control system of a magnetic levitation motor according to claim 2, characterized in that: The second-order generalized integrator further includes an internal frequency adjuster for dynamically adapting the frequency of a signal input to the integrator according to the real-time frequency condition of the motor operation.

4. The control system of a magnetic levitation motor according to claim 1, characterized in that: The transfer function of the second-order generalized integrator is: In the formula is the gain coefficient of the second-order generalized integrator SOGI, is the center frequency of the second-order generalized integrator SOGI, which is adjusted according to the operating frequency range of the motor and the interference frequency that needs to be suppressed and The value of .

5. The control system of a magnetic levitation motor according to claim 1, characterized in that: The phase-locked loop also includes an adder for converting the phase estimate With the compensated phase estimate Add together to get the phase estimate .

6. The method for using the control system of a magnetic levitation motor according to any one of claims 1 to 5, characterized in that: The following steps are involved: Input two stationary coordinate system currents , and voltage 、 , output estimated back EMF 、 ; To the linear extended state observer, the output is the estimated back EMF 、 ; Input the output of the linear extended state observer to the phase-locked loop and output the speed estimate and the rotor position estimate ; Calculate the estimated speed using the function calculator and The product of The output result of the function calculator and the reference speed are calculated by the first subtractor difference; Input the result of the first subtractor to the first PI controller and output the q-axis current reference value ; The q-axis current reference value is calculated by the second subtractor The difference between the d-axis current id; Calculating the difference between the q-axis current iq and the d-axis reference current idref by a third subtractor; The result of the second subtractor is input to the second PI controller, and the q-axis voltage Uq is output; The result of the third subtractor is input to the third PI controller, and the d-axis voltage Ud is output; Input q-axis voltage Uq, d-axis voltage Ud and rotor position estimation value To the inverse Park converter, output voltage in the two-phase stationary coordinate system 、 ; Input voltage in two-phase stationary coordinate system 、 To the space vector pulse width modulator, output three-phase voltage 、 、 Control magnetic levitation motor; The three-phase current of the magnetic levitation motor is converted by the Clarke converter 、 、 Converted to the current in the two-phase stationary coordinate system , ; Input current in the two-phase stationary coordinate system , and the rotor position estimate To the Park converter, output the d-axis current id to the second subtractor, and output the q-axis current iq to the third subtractor, completing the cycle of the closed-loop system.

7. A magnetic levitation motor, characterized in that: A control system for a magnetic levitation motor comprising any one of claims 1 to 5.

8. The magnetic levitation motor according to claim 7, characterized in that: The magnetic levitation motor further includes a drive circuit, which is electrically connected to the magnetic levitation motor.

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