Controller and control method for efficiency optimization of four-switch buck-boost converter

Abstract

The invention provides a controller and a control method for efficiency optimization of a four-switch buck-boost converter, and the controller comprises a voltage sampler and a current sampler which are respectively used for obtaining an output voltage vf2 signal of the four-switch buck-boost converter, namely an inductive current vifeed signal in the four-switch buck-boost converter; the first loop compensator PID1 receives an error signal ve1 of a reference voltage vr2 signal and a reference voltage vf2 signal and generates a first compensation reference signal vr1; the second loop compensator PID2 is used for generating and outputting a vpid1 signal according to the first compensation reference signal vr1 and an error signal ve2 of the vifeed signal; the third loop compensator PID3 is used for generating and outputting a vpid2 signal according to the first compensation reference signal vr1 and an inverse signal ve3 of an error of the vifeed signal; and the nonlinear controller is used for generating four control signals according to the vpid1 signal, the vpid2 signal and the vifeed signal so as to control four switching tubes in the four-switch buck-boost converter. The control method is used for efficiency optimization.

Description

Technical Field

[0001] The present invention relates to a controller and a control method for a four-switch buck-boost converter, in particular to a controller and a control method for optimizing the efficiency of a four-switch buck-boost converter. Background Art

[0002] The information provided in this section is only background information related to the present disclosure, and it does not necessarily represent prior art.

[0003] With the continuous improvement of the performance requirements of modern power electronic devices for energy conversion systems, the traditional Buck or Boost topologies can no longer meet the dynamic regulation requirements for a wide input-output voltage range under complex working conditions. The four-switch Buck-Boost converter (FSBB) has become a key technical direction for achieving efficient energy management due to its unique symmetric structure and flexible operation mode. Compared with traditional unidirectional buck / boost strategies and other non-isolated topologies, the four-switch structure realizes continuously adjustable voltage gain through innovative control logic, effectively covering the voltage gain intervals where the input voltage is higher or lower than the output voltage.

[0004] The four-switch Buck-Boost converter has three operating modes: Buck, Boost, and Buck-Boost modes. When the input voltage changes, it is necessary to switch among the three modes. Obviously, in different operating modes, the conversion efficiency varies due to the different numbers of working switches. Taking the fixed-frequency PWM modulation method as an example, in the Buck-Boost mode, all four switching tubes operate in the high-frequency chopping mode; while in the Buck and Boost modes, the switching tubes of the bridge arm with the higher voltage operate at high frequency, and the high-side switching tube of the bridge arm with the lower voltage is in the through mode. The difference in the number of working switching tubes causes differences in switching losses, conduction losses, etc., resulting in different conversion efficiencies. Under the same working conditions, such as when the voltages on both sides are the same, there are multiple static operating points for the duty cycle of the converter. Under different static operating points, the duty cycle has a great influence on the switching losses and inductor current ripple of the converter, resulting in differences in losses under different static operating points.

[0005] In the prior art, the efficiency of the operating point of the converter is relatively low, and the converter has large losses at some operating points, and there is still a large room for optimizing the conversion efficiency.

[0006] It should be noted that the information disclosed in the above background art section is only used to enhance the understanding of the background of the present disclosure, and therefore may include information that does not constitute prior art known to those of ordinary skill in the art. Summary of the Invention

[0007] Objective of the Invention: The technical problem to be solved by the present invention is to provide a controller and a control method for optimizing the efficiency of a four-switch buck-boost converter in view of the deficiencies of the prior art.

[0008] To solve the above technical problem, the present invention discloses a controller and a control method for optimizing the efficiency of a four-switch buck-boost converter, wherein the controller includes:

[0009] A voltage sampler and a current sampler, which respectively obtain the output voltage vf2 signal of the four-switch buck-boost converter, that is, the inductor current vifed signal in the four-switch buck-boost converter;

[0010] A first loop compensator, with the input being the difference signal ve1 between the reference voltage vr2 signal and the vf2 signal, that is, ve1 = vr2 - vf2, and the output being the first compensation reference signal vr1;

[0011] A second loop compensator, with the input being the difference signal ve2 between the first compensation reference signal vr1 and the vifed signal, that is, ve2 = vr1 - vifed, and the output being the vpid1 signal;

[0012] A third loop compensator, with the input being the inverted signal ve3 of the difference between the first compensation reference signal vr1 and the vifed signal, that is, ve3 = -(vr1 - vifed), and the output being the vpid2 signal;

[0013] A non-linear controller, which generates 4 control signals according to the vpid1 signal, the vpid2 signal and the vifed signal to control the 4 switching tubes in the four-switch buck-boost converter.

[0014] Further, the controller further includes:

[0015] A drive amplifier circuit;

[0016] Assume that the 4 control signals generated by the non-linear controller are PWM_D1, PWM_D2, PWM_D3 and PWM_D4 respectively. After the drive amplifier circuit amplifies the above signals, it sends them to the gates of the 4 switching tubes in the four-switch buck-boost converter, namely the first switching tube, the second switching tube, the third switching tube and the fourth switching tube, for controlling the four-switch buck-boost converter to achieve the efficiency optimization.

[0017] The present invention also proposes a control method for optimizing the efficiency of a four-switch buck-boost converter, including the following steps:

[0018] Step 1, calculate the first input signal of the non-linear controller and the second input signal of the non-linear controller according to the output voltage signal of the four-switch buck-boost converter, the inductor current signal in the four-switch buck-boost converter and the reference voltage signal;

[0019] Step 2: Determine the maximum duty cycle of the switching transistors in the four-switch buck-boost converter according to the direction and magnitude of the inductor current in the four-switch buck-boost converter.

[0020] Step 3: Calculate and generate a control signal based on the maximum duty cycle of the switching transistors in the four-switch buck-boost converter, the first input signal of the non-linear controller, and the second input signal of the non-linear controller, and control the four-switch buck-boost converter to optimize the efficiency of the four-switch buck-boost converter.

[0021] Further, calculating the first input signal of the non-linear controller and the second input signal of the non-linear controller in Step 1 includes:

[0022] Step 1-1: Sample the output voltage of the four-switch buck-boost converter using a voltage sampler to obtain the vf2 signal.

[0023] Step 1-2: Sample the current of the four-switch buck-boost converter using a current sampler to obtain the vifed signal.

[0024] Step 1-3: Use a first loop compensator to perform error comparison and tracking compensation on the vf2 signal and the reference voltage vr2 signal. The input signal of the first loop compensator is ve1, and the output of the first loop compensator is the first compensated reference signal vr1.

[0025] Step 1-4: Use a second loop compensator to perform error comparison and tracking compensation on the first compensated reference signal vr1 and the vifed signal. The input signal of the second loop compensator is ve2, and the output of the second loop compensator is the first input signal vpid1 of the non-linear controller.

[0026] Step 1-5: Use a third loop compensator to perform error comparison and tracking compensation on the first compensated reference signal vr1 and the vifed signal. The input signal of the third loop compensator is ve3, and the output of the third loop compensator is the second input signal vpid2 of the non-linear controller.

[0027] Further, the input signal ve1 of the first loop compensator in Step 1-3 is expressed as follows:

[0028] ve1 = vr2 - vf2

[0029] The input signal ve2 of the second loop compensator in Step 1-4 is expressed as follows:

[0030] ve2 = vr1 - vife.

[0032] Further, the input signal ve3 of the third loop compensator described in steps 1-5 is expressed as follows:

[0033] ve3 = -(vr1 - vifed).

[0035] Further, determining the maximum duty cycle of the switching tubes in the four-switch buck-boost converter described in step 2 includes:

[0036] Step 2-1, set the current threshold vth;

[0037] Step 2-2, when the inductor current is in the positive direction of the associated direction, that is, the sign of the inductor current vifed signal is positive, and the value of the vifed signal satisfies the following condition:

[0038] vifed ≥ vth

[0039] Then set the maximum duty cycle D3max of the third switching tube in the four-switch buck-boost converter, which is expressed as follows:

[0040] D3max = Dmax<1

[0041] where Dmax is the maximum duty cycle of the four-switch buck-boost converter;

[0042] Otherwise, the maximum duty cycle D3max is not restricted, that is, the maximum duty cycle D3max takes the value of 1;

[0043] Step 2-3, when the inductor current is in the negative direction of the associated direction, that is, the sign of the inductor current vifed signal is negative, when the value of the vifed signal satisfies the following condition:

[0044] vifed ≤ -vth

[0045] Then set the maximum duty cycle D1max of the first switching tube in the four-switch buck-boost converter, which is expressed as follows:

[0046] D1max = Dmax<1

[0047] Otherwise, the maximum duty cycle D1max is not restricted, that is, the maximum duty cycle D1max takes the value of 1.

[0048] Further, calculating and generating the control signal described in step 3 includes:

[0049] Step 3-1, limit the first input signal vpid1 of the non-linear controller using the interval [0, D1max], specifically including:

[0050] When vpid1 > D1max, let vpid1 = D1max;

[0051] When vpid1 < 0, set vpid1 = 0;

[0052] Step 3-2: Limit the amplitude of the second input signal vpid2 of the non-linear controller using the interval [0, D3max], specifically including:

[0053] When vpid2 > D3max, set vpid2 = D3max;

[0054] When vpid2 < 0, set vpid2 = 0;

[0055] Step 3-3: Calculate the PWM control signal PWM_D1 of the first switching tube with a duty cycle of D1 according to the value of the amplitude-limited first input signal vpid1 of the non-linear controller, and its inverted signal is the PWM control signal PWM_D2 of the second switching tube with a duty cycle of D2;

[0056] Step 3-4: Calculate the PWM control signal PWM_D3 of the third switching tube with a duty cycle of D3 according to the value of the amplitude-limited second input signal vpid2 of the non-linear controller, and its inverted signal is the PWM control signal PWM_D4 of the fourth switching tube with a duty cycle of D4.

[0057] Further, the calculation of the PWM control signal PWM_D1 of the first switching tube with a duty cycle of D1 according to the value of the amplitude-limited first input signal vpid1 of the non-linear controller in Step 3-3 specifically includes:

[0058] Use the value of the amplitude-limited first input signal vpid1 of the non-linear controller as the modulation wave value, compare it with the triangular carrier wave Tsw, and obtain the PWM control signal PWM_D1 of the first switching tube with a duty cycle of D1.

[0059] Further, the calculation of the PWM control signal PWM_D3 of the third switching tube with a duty cycle of D3 according to the value of the amplitude-limited second input signal vpid2 of the non-linear controller in Step 3-4 specifically includes:

[0060] Use the value of the amplitude-limited second input signal vpid2 of the non-linear controller as the modulation wave value, compare it with the triangular carrier wave Tsw, and obtain the PWM control signal PWM_D3 of the third switching tube with a duty cycle of D3.

[0061] Beneficial effects:

[0062] 1. The present invention only adjusts the control method. Without affecting the dynamic performance, it adjusts the respective working voltage gain intervals in three modes, improving the conversion efficiency at some voltage operating points.

[0063] 2. In the charge and discharge application scenarios under typical energy storage applications, when the voltages on both sides are very close, the present invention can narrow the operating voltage range of the converter in the Buck-Boost mode, enabling the converter to operate in the Buck and Boost modes as much as possible, so as to reduce the losses of the converter, improve the overall conversion efficiency, and reduce the energy loss caused by power conversion, having clear application value. BRIEF DESCRIPTION OF THE DRAWINGS

[0064] The following further describes the present invention in detail with reference to the drawings and specific embodiments, and the above and / or other advantages of the present invention will become clearer.

[0065] Figure 1 It is a schematic diagram of the overall structure of the power part and the control part of the FSBB converter.

[0066] Figure 2 It is a schematic diagram of the control method of the non-linear controller of the controller part of the FSBB converter.

[0067] Figure 3 It is a schematic diagram of the third switch tube commuting to the fourth switch tube when the inductor current of the FSBB converter is in the positive direction.

[0068] Figure 4 It is a schematic diagram of the fourth switch tube commuting to the third switch tube when the inductor current of the FSBB converter is in the positive direction.

[0069] Figure 5 It is a schematic diagram of the third switch tube commuting to the fourth switch tube when the inductor current of the FSBB converter is in the negative direction.

[0070] Figure 6 It is a schematic diagram of the fourth switch tube commuting to the third switch tube when the inductor current of the FSBB converter is in the negative direction.

[0071] Figure 7 It is a simulation schematic diagram of the FSBB converter without using the control method of the present invention.

[0072] Figure 8 It is a simulation schematic diagram of the FSBB converter using the control method of the present invention.

[0073] Figure 9 It is a simulation schematic diagram of the FSBB converter without using the control method of the present invention in each mode.

[0074] Figure 10 It is a simulation schematic diagram of the FSBB converter using the control method of the present invention in each mode. DETAILED DESCRIPTION OF THE EMBODIMENTS

[0075] The general idea of the present invention is as follows: By only adjusting the control method, without affecting the dynamic performance, the respective working voltage gain ranges in three modes are adjusted to improve the conversion efficiency at some voltage operating points. In the charge-discharge application scenario of typical energy storage applications, when the voltages on both sides are very close, the present invention can narrow the working voltage range of the converter in the Buck-Boost mode, so that the converter works in the Buck and Boost modes as much as possible, thereby reducing the loss of the converter, improving the overall conversion efficiency, and reducing the energy loss caused by power conversion, which has clear application value.

[0076] The basic principle of the present invention is to limit the maximum duty cycle of the switching tubes on both bridge arms respectively according to the direction and magnitude of the inductor current of the four-switch Buck-Boost converter. The main considerations for this limitation are the freewheeling mode of the MOSFET, the conduction loss and reverse recovery loss of the body diode, the driving loss of the MOSFET, and the conduction loss in the linear region of the MOSFET. When the direction and magnitude of the inductor current are not conducive to the above losses, the maximum duty cycle is restricted; otherwise, the maximum duty cycle is not restricted, that is, the maximum duty cycle can be 1. According to this control method, the working voltage range of the four-switch Buck-Boost in the Buck mode and Boost mode can be expanded, and the efficiency of the converter can be further improved. The specific content is described in detail as follows:

[0077] The four-switch Buck-Boost converter described in the present invention, that is, the four-switch buck-boost converter, includes a first switching tube S1, a second switching tube S2, a third switching tube S3, a fourth switching tube S4, a first capacitor C1, a second capacitor C2, and a first inductor L1. The source of the first switching tube S1 and the drain of the second switching tube S2 are connected in series to form a first half-bridge. The source of the third switching tube S3 and the drain of the fourth switching tube S4 are connected in series to form a second half-bridge. Both ends of the inductor L1 are connected between the source of the first switching tube S1 and the source of the third switching tube S3. The first capacitor C1 is connected between the drain of the first switching tube S1 and the source of the second switching tube S2. The second capacitor C2 is connected between the drain of the third switching tube S3 and the source of the fourth switching tube S4. The source of the second switching tube S2 is connected to the source of the fourth switching tube S4. The voltage of the first capacitor C1 is V1, the voltage of the second capacitor C2 is V2, the duty cycle of the first switching tube S1 is D1, the duty cycle of the third switching tube S3 is D3, the duty cycle of the second switching tube S2 is D2, and the duty cycle of the fourth switching tube S4 is D4.

[0078] The controller of the four-switch Buck-Boost converter described in the present invention includes: a first loop compensator PID1, a second loop compensator PID2, a third loop compensator PID3, a current sampler, a voltage sampler, and a non-linear controller; the voltage sampler samples the voltage V2 and outputs the vf2 signal, and the current sampler samples the current of the inductor L1 and outputs the vifed signal; the first loop compensator PID1 samples the V2 voltage, performs error comparison and tracking compensation with the voltage reference value vr2, and the input error signal is expressed as: vr2 - vf2, generating a first compensation reference signal vr1. The current sampler samples the inductor current signal vifed, and the input of the second loop compensator PID2 is the error signal, expressed as: vr1 - vifed, and outputs the vpid1 signal; the input of the third loop compensator PID3 is the inverse signal of the error signal, expressed as: -(vr1 - vifed), and outputs the vpid2 signal. The vpid1 signal, the vpid2 signal, and the current sampling signal vifed are the input signals of the non-linear controller.

[0079] As Figure 2 shown, the vpid1 signal undergoes duty cycle transformation through the non-linear controller to output a PWM signal PWM_D1 with a duty cycle of D1 and a PWM signal PWM_D2 with a duty cycle of D2; the vpid2 signal undergoes duty cycle transformation through the non-linear controller to output a PWM signal PWM_D3 with a duty cycle of D3 and a PWM signal PWM_D4 with a duty cycle of D4. The duty cycle D2 of the second switch tube S2 is complementary to the duty cycle D1 of the first switch tube S1, and they satisfy D1 + D2 = 1. The duty cycle D3 of the third switch tube S3 is complementary to the duty cycle D4 of the fourth switch tube S4, and they satisfy D3 + D4 = 1.

[0080] The maximum duty cycle of the first switch tube S1 is Dmax1 respectively, and the minimum duty cycle is Dmin1; the maximum duty cycle of the third switch tube S3 is Dmax3 respectively, and the minimum duty cycle is Dmin3. According to the rising edge time and dead time of the hardware drive circuit of the FSBB converter, the limit value Dmax of the usually designed maximum duty cycle < 1, such as Dmax = 0.95. The non-linear controller contains a triangular carrier signal for comparing and generating PWM. The minimum value of the triangular carrier Tsw is 0, the maximum value is 1, and the frequency is fsw.

[0081] When the duty cycle D1 of the first switch tube S1 < 1 and the duty cycle D3 of the third switch tube S3 = 1, the FSBB converter is in the Buck mode;

[0082] When the duty cycle D1 of the first switch tube S1 = 1 and the duty cycle D3 of the third switch tube S3 < 1, the FSBB converter is in the Boost mode;

[0083] When the duty cycle D1 of the first switching transistor S1 < 1 and the duty cycle D3 of the third switching transistor S3 < 1, the FSBB converter operates in Buck - Boost mode.

[0084] The non - linear controller needs to transform the vpid1 signal into the driving PWM signal PWM_D1 of the first switching transistor with a duty cycle of D1, and the vpid2 signal into the driving PWM signal PWM_D3 of the third switching transistor with a duty cycle of D3 according to the associated direction of the inductor current, that is, the sign of the inductor current sampling signal vifed. The control method adopted by the non - linear controller is as follows:

[0085] 1) When the inductor current is in the positive direction of the associated direction, the sign of vifed is positive. When vifed is greater than the current threshold vth, vifed ≥ vth, then set the maximum duty cycle D3max of the duty cycle D3 to be less than 1, that is, D3max = Dmax < 1; otherwise, the maximum duty cycle D3max of the duty cycle D3 is not restricted, and the maximum duty cycle D3max takes the value of 1.

[0086] 2) When the inductor current is in the negative direction of the associated direction and the sign of vifed is negative. When vifed is less than the current threshold - vth, vifed ≤ - vth, then set the maximum duty cycle D1max of the duty cycle D1 to be less than 1, that is, D1max = Dmax < 1; otherwise, the maximum duty cycle D1max of the duty cycle D1 is not restricted, and the maximum duty cycle D1max takes the value of 1.

[0087] 3) Limit the vpid1 signal using the interval [0, D1max]. When vpid1 > D1max, vpid1 = D1max; when vpid1 < 0, vpid1 = 0.

[0088] 4) Limit the vpid2 signal using the interval [0, D3max]. When vpid2 > D3max, vpid2 = D3max; when vpid2 < 0, vpid2 = 0.

[0089] 5) Take the value corresponding to the limited vpid1 as the modulation wave value, compare it with the triangular carrier wave Tsw, and obtain the PWM signal PWM_D1 of the first switching transistor with a duty cycle of D1. Its inverted signal is the PWM signal PWM_D2 of the second switching transistor with a duty cycle of D2.

[0090] 6) Take the value corresponding to the limited vpid2 as the modulation wave value, compare it with the triangular carrier wave Tsw, and obtain the PWM signal PWM_D3 of the third switching transistor with a duty cycle of D3. Its inverted signal is the PWM signal PWM_D4 of the fourth switching transistor with a duty cycle of D4.

[0091] 7) The non - linear controller outputs the PWM signal PWM_D1 of the first switching tube with a duty cycle of D1, the PWM signal PWM_D2 of the second switching tube with a duty cycle of D2, the PWM signal PWM_D3 of the third switching tube with a duty cycle of D3, and the PWM signal PWM_D4 of the fourth switching tube with a duty cycle of D4.

[0092] By implementing the control process and control method of the non - linear controller, when the converter operates bidirectionally, the operating voltage range in the single - buck (Buck) mode and single - boost (Boost) mode can be broadened according to the power flow direction and magnitude, that is, the direction and magnitude of the inductor current, and the conversion efficiency of the four - switch converter can be improved.

[0093] Embodiment:

[0094] The four - switch Buck - Boost converter operates as follows Figure 1 As shown, when the voltage on the V1 side is lower than the voltage on the V2 side, it operates in the Buck mode, the duty cycle D1 of the first switching tube is less than 1, and the duty cycle D3 of the third switching tube is equal to 1; when V1 gradually approaches the V2 voltage, the duty cycle of D1 gradually increases and approaches 1. When the voltage on the V1 side is higher than the voltage on the V2 side, it operates in the Boost mode, the duty cycle D3 of the third switching tube is less than 1, and the duty cycle D1 of the first switching tube is equal to 1; when V1 approaches the V2 voltage, the converter operates in the Buck - Boost mode. Since the number of high - frequency switches in the Buck - Boost mode is 4, which is greater than the number of high - frequency switches in the Buck or Boost mode, under the same operating conditions, the conversion efficiency of Buck - Boost is usually lower than that of the Buck mode or the Boost mode, and as the duty cycles D1 and D3 of Buck - Boost become smaller, the losses increase more. In practical applications, with high efficiency as the optimization goal, it is more desirable to cover a wider voltage gain range in the Buck mode or the Boost mode.

[0095] Buck - Boost is usually designed to operate within a relatively narrow voltage gain range (such as V1 / V2 = 0.9 - 1.1), which is reflected in the maximum duty cycles of D1 and D3 in the bridge arm. However, under different inductor current directions, the maximum duty cycle constraints of D1 and D3 are different, and the following analysis will be carried out.

[0096] The third switching tube S3 and the fourth switching tube S4 conduct complementarily. When S3 conducts, the inductor current flows through the S3 channel; when S3 turns off and S4 conducts, the inductor current flows through the S4 channel. Usually, there is a dead - time between S3 and S4, and the driving speeds of S3 and S4 also affect the process of the inductor current commuting from S3 to S4.

[0097] Assume that the inductor current is in the positive direction. Consider the commutation process from S3 to S4 as shown in Figure 3 and Figure 4。From t1 to t2, the voltage of the S3 switch driving signal PWM_D3 gradually drops to zero, and the current commutates between the channel and the body diode of S3. During the period from t2 to t3, it is the dead time for the driving of S3 and S4, and the current flows through the body diode of S3. During the period from t3 to t4, the voltage of the S4 driving signal PWM_D4 gradually rises, and the current gradually commutates from the body diode of S3 to the channel of S4, and the current commutation is completed at t4. During the period from t3 to t4, before the driving voltage of S4 slowly rises to the driving gate saturation voltage (usually <8V), S4 operates in the MOSFET linear region rather than the saturation region, and the on-resistance of S4 is relatively high; after the driving voltage of S4 reaches the gate saturation voltage, S4 operates in the saturation region and the on-resistance is small. At t5, the S4 switch driving signal gradually drops to zero, and the current commutates from the S4 switch channel to the S3 switch body diode. The period from t6 to t7 is the dead time period. When designing, sufficient dead time is reserved to ensure that the current can complete the commutation process during the period from t5 to t7. From t7 to t8, the voltage of the S3 switch driving signal gradually rises, and the current gradually commutates from the S3 body diode to the S3 channel, and S3 is turned on with zero voltage switching (ZVS). From the above analysis, in order to avoid large conduction losses and switching losses caused by S4 operating in the linear region during the period from t3 to t4, the minimum value of the duty cycle of S4 should be limited, and the maximum value of the duty cycle of S3 should be limited.

[0098] When the inductor current is in the negative direction, consider the same commutation process as Figure 5 and Figure 6 。From t1 to t2, the voltage of the S3 switch driving signal PWM_D3 gradually drops to zero. Since the inductor current is negative, the current commutates between the channel of S3 and the body diode of S4. During the period from t2 to t3, it is the dead time for the driving of S3 and S4, and the current flows through the body diode of S4. During the period from t3 to t4, the voltage of the S4 driving signal PWM_D4 gradually rises, and the current gradually commutates from the body diode of S4 to the channel of S4, and the current commutation is completed at t4. The turn-on of S4 is zero voltage switching (ZVS). At t5, the S4 switch driving signal gradually drops to zero, and the current commutates from the S4 switch channel to the S4 switch body diode. The period from t6 to t7 is the dead time period, and the current flows through the body diode of S4. From t7 to t8, the voltage of the S3 switch driving signal gradually rises, and the current gradually commutates from the body diode of S4 to the channel of S3. Before the driving voltage of S3 slowly rises to the driving gate saturation voltage (usually <8V), S3 operates in the MOSFET linear region rather than the saturation region, and the on-resistance of S3 is relatively high; after the driving voltage of S3 reaches the gate saturation voltage, S4 operates in the saturation region and the on-resistance is small. From the above analysis, in order to avoid large conduction losses and switching losses caused by S3 operating in the linear region during the period from t7 to t8, the minimum value of the duty cycle of S3 should be limited, and the maximum value of the duty cycle of S4 should be limited.

[0099] In the above two commutation processes, the positive and negative directions of the inductor current determine the need to design a minimum duty cycle constraint for S3 or S4 to avoid linear conduction losses and switching losses during the commutation process. Therefore, considering a symmetric structure, when the inductor current direction is positive, the maximum duty cycle of S3 switch needs to be constrained to be less than 1, while the maximum duty cycle of S1 switch can be 1.0; when the inductor current is negative, the maximum duty cycle of S1 switch needs to be constrained to be less than 1, while the maximum duty cycle of S3 switch can be 1.0. Therefore, when operating in Buck mode and the power flow direction is from V1 to V2, the inductor current is positive. Since there is no need to limit the maximum duty cycle of S1 switch, D1→1.0, and the voltage gain range of Buck mode can be extended from V1 / V2>1 to approaching 1, that is, V1 / V2→1. When operating in Boost mode and the power flow direction is from V2 to V1, the inductor current is negative. Since there is no need to limit the maximum duty cycle of S3 switch, D3→1.0, and the operating range of Boost mode can be extended from V2 / V1>1 to approaching 1, that is, V2 / V1→1. Through this control method, the voltage gain range and operating range of Buck and Boost modes are extended.

[0100] Take the 48V energy storage system as an example. There is a lithium battery connected to the V1 side, and the voltage range is typically 40.5 - 54.75V (15 series 3.2V single cells). The V2 side is connected to the DC bus and regulated to 48V. During discharge, the power flow direction is from V1 to V2. In the section where the battery SOC>80%, V1 = 54.75 - 48V. Using the above control strategy, it can work in single Buck mode as much as possible when the SOC is close to full charge state, reducing the power loss during battery discharge and equivalently increasing the discharge capacity of the battery; when charging in reverse, the power flow direction is from V2 to V1. In the voltage range of V1 side voltage V1 = 45 - 48V, using the above control strategy, it can work in single Boost mode as much as possible in this voltage range, reducing the power loss during battery charging. To sum up, using the control strategy in this scheme, when the battery SOC is around 80% and above, the overall charge-discharge efficiency is improved, thereby increasing the overall revenue of the energy storage system.

[0101] Next, the control method described in the present invention is simulated to verify its application effect.

[0102] The four-switch Buck-Boost system proposed in the present invention is simulated. The simulation example used is a bidirectional conversion scenario under typical 48V energy storage applications, using the power stage and control stage structures as shown in Figure 1 The voltage V1 = 35 - 65V, and the voltage V2 = 48V is regulated. The maximum duty cycles of S1 switch and S3 switch are fixed at Dmax = 0.95.

[0103] Simulate the situation of battery discharging. The inductor current is controlled at a constant current of 20 A, the voltage of V2 is a constant voltage of 48 V, and the voltage on the V1 side changes slowly from 65 V to 35 V to verify the dynamic process of three-mode switching caused by the change of the V1 voltage:

[0104] Figure 7 For the four-switch Buck-Boost converter in this example Not adopted The mode-switching simulation of the control method of the present invention. Simulate the battery discharging state. When the battery voltage (input voltage) gradually decreases and the direction of the inductor current is positive, the key waveforms of each stage from Buck to Buck-Boost to Boost mode are covered in sequence; Figure 7 The time-domain dynamic waveforms of the mode-switching process are shown; when the V1 voltage changes slowly from 65 V to 52.1 V, the converter operates in the Buck mode; when V1 changes slowly from 52.1 V to 46.8 V, the converter operates in the Buck-Boost mode; when V1 is below 46.8 V, the converter operates in the Boost mode. Figure 8 For the mode-switching simulation of the four-switch Buck-Boost converter in the example when adopting the control method of the present invention. Simulate the battery discharging state. When the battery voltage (input voltage) gradually decreases and the direction of the inductor current is positive, the key waveforms of each stage from Buck to Buck-Boost to Boost mode are covered in sequence; Figure 8 The time-domain dynamic waveforms of the mode-switching process are shown; when the V1 voltage changes slowly from 65 V to 49.6 V, the converter operates in the Buck mode; when V1 changes slowly from 49.6 V to 46.8 V, the converter operates in the Buck-Boost mode; when V1 is below 46.8 V, the converter operates in the Boost mode. It can be known from the simulation comparison that by adopting the control method designed by the present invention, the working voltage range of Buck is extended from 52.1 V to 49.6 V. Considering the factors of line impedance and voltage drop, at this time V1 is already very close to the V2 voltage of 48 V. During mode switching, the control method of this scheme does not affect the dynamic process of the original controller, and the inductor current is controlled smoothly.

[0105] Simulate the situation of battery charging. The inductor current is controlled at a constant current of -20 A, the voltage of V2 is a constant voltage of 48 V, and the voltage on the V1 side changes slowly from 35 V to 65 V to verify the dynamic process of three-mode switching caused by the change of the V1 voltage:

[0106] Figure 9The simulation of mode switching when the four-switch Buck-Boost converter in the example does not adopt the control method of the present invention. Simulating the battery charging state, when the battery voltage (input voltage) gradually increases and the inductor current direction is negative, the key waveforms of each stage from Boost to Buck-Boost to Buck mode are covered in sequence. Figure 9 It shows the time-domain dynamic waveforms during the mode switching process; when the voltage of V1 slowly changes from 35V to 45.0V, the converter operates in the Boost mode; when V1 slowly changes from 45.0V to 50.8V, the converter operates in the Buck-Boost mode; when V1 is above 50.8V, the converter operates in the Buck mode. Figure 10 For the four-switch Buck-Boost converter in the example Adopted The simulation of mode switching when adopting the control method of the present invention. Simulating the battery charging state, when the battery voltage (input voltage) gradually increases and the inductor current direction is negative, the key waveforms of each stage from Boost to Buck-Boost to Buck mode are covered in sequence. Figure 10 It shows the time-domain dynamic waveforms during the mode switching process; when the voltage of V1 slowly changes from 35V to 46.9V, the converter operates in the Boost mode; when V1 slowly changes from 46.9V to 50.8V, the converter operates in the Buck-Boost mode; when V1 is above 50.8V, the converter operates in the Buck mode. It can be known from the simulation comparison that by adopting the control method designed by the present invention, the working voltage range of Boost is extended from 45.0V to 46.9V. Considering the factors of line impedance and voltage drop, at this time V1 is already very close to the V2 voltage of 48V. During the mode switching, the control method of this solution does not affect the dynamic process of the original controller, and the inductor current is controlled smoothly.

[0107] It can be proved by simulation that:

[0108] 1. By adopting the control method described in the present invention, the working ranges of the four-switch Buck-Boost converter in the single Buck mode and the single Boost mode can be extended, and the overall power conversion efficiency of the converter can be improved.

[0109] 2. The control method described in the present invention does not affect the control accuracy and response speed of the original controller and the compensation loop.

[0110] 3. In typical application scenarios such as energy storage, the control method described in the present invention has clear application value.

[0111] In a specific implementation, the present application provides a computer storage medium and a corresponding data processing unit. Among them, the computer storage medium can store a computer program, and when the computer program is executed by the data processing unit, it can run the content of the invention of a controller and a control method for optimizing the efficiency of a four-switch buck-boost converter and some or all of the steps in each embodiment. The storage medium can be a magnetic disk, an optical disk, a read-only memory (ROM), a random access memory (RAM), or the like.

[0112] Those skilled in the art can clearly understand that the technical solutions in the embodiments of the present invention can be implemented by means of a computer program and its corresponding general hardware platform. Based on such an understanding, the essence of the technical solutions in the embodiments of the present invention, or the part that contributes to the prior art, can be embodied in the form of a computer program, that is, a software product. The computer program software product can be stored in the storage medium and includes several instructions for causing a device including a data processing unit (which can be a personal computer, a server, a single-chip microcomputer, an MCU, or a network device, etc.) to execute the methods described in each embodiment or some parts of the embodiments of the present invention.

[0113] The present invention provides an idea and method for a controller and a control method for optimizing the efficiency of a four-switch buck-boost converter. There are many methods and ways to specifically implement this technical solution. The above is only the preferred implementation manner of the present invention. It should be noted that for those of ordinary skill in the art in this technical field, without departing from the principle of the present invention, several improvements and refinements can be made, and these improvements and refinements should also be regarded as the protection scope of the present invention. Each component not clearly defined in this embodiment can be implemented by the prior art.

Claims

1. A controller for optimizing the efficiency of a four-switch buck-boost converter, characterized in that: include: A voltage sampler and a current sampler, respectively acquiring an output voltage vf2 signal of the four-switch buck-boost converter, that is, an inductor current vifed signal in the four-switch buck-boost converter; The first loop compensator PID1, whose input is the error signal ve1 of the reference voltage vr2 signal and the vf2 signal, and whose output is the first compensated reference signal vr1; The second loop compensator PID2 has an input of the first compensation reference signal vr1 and an error signal ve2 of the vifed signal, and an output of the output vpid1 signal; The third loop compensator PID3, whose input is the first compensation reference signal vr1 and the inverted signal ve3 of the error signal of the vifed signal, and whose output is the output vpid2 signal; The nonlinear controller generates four control signals according to the vpid1 signal, the vpid2 signal and the vifed signal to control four switch tubes in the four-switch buck-boost converter.

2. A controller for optimizing efficiency of a four-switch buck-boost converter according to claim 1, characterized in that: The controller further comprises: Driving amplifier circuit; Assume that the four control signals generated by the nonlinear controller are PWM_D1, PWM_D2, PWM_D3 and PWM_D4 respectively. After the driving amplifier circuit amplifies the above signals, they are sent to the gates of the four switch tubes in the four-switch buck-boost converter, namely the first switch tube S1, the second switch tube S2, the third switch tube S3 and the fourth switch tube S4, respectively, for controlling the four-switch buck-boost converter to achieve the efficiency optimization.

3. A control method for optimizing the efficiency of a four-switch buck-boost converter, characterized in that: The following steps are involved: Step 1, calculating a first input signal of a nonlinear controller and a second input signal of a nonlinear controller according to an output voltage signal of the four-switch buck-boost converter, an inductor current signal in the four-switch buck-boost converter and a reference voltage signal; Step 2, determining the maximum value of the duty cycle of the switch tube in the four-switch buck-boost converter according to the direction of the inductor current in the four-switch buck-boost converter; Step 3, calculate and generate a control signal based on the maximum value of the duty cycle of the switch tube in the four-switch buck-boost converter, the first input signal of the nonlinear controller and the second input signal of the nonlinear controller, control the four-switch buck-boost converter, and achieve efficiency optimization of the four-switch buck-boost converter.

4. The control method for optimizing the efficiency of a four-switch buck-boost converter according to claim 3, characterized in that: The step 1 of calculating the first input signal of the nonlinear controller and the second input signal of the nonlinear controller comprises: Step 1-1, using a voltage sampler to sample the output voltage of the four-switch buck-boost converter to obtain a vf2 signal; Step 1-2, using a current sampler to sample the current of the four-switch buck-boost converter to obtain a vifed signal; Step 1-3, the first loop compensator PID1, the input signal is the error signal ve1 between the vf2 signal and the reference voltage vr2 signal, and the output is the first compensated reference signal vr1; Step 1-4, the second loop compensator PID2, the input signal is the error signal ve2 of the first compensation reference signal vr1 and the vifed signal, and the output is the first input signal vpid1 of the nonlinear controller; Step 1-5, the third loop compensator PID3, the input signal is the inverted signal ve3 of the error signal of the first compensation reference signal vr1 and the vifed signal, and the output is the second input signal vpid2 of the nonlinear controller.

5. The control method for optimizing the efficiency of a four-switch buck-boost converter according to claim 4, characterized in that: The error signal ve1 described in step 1-3 is expressed as follows: ve1=vr2-vf2 The error signal ve2 described in step 1-4 is expressed as follows: ve2=vr1-vifed.

6. A control method for optimizing the efficiency of a four-switch buck-boost converter according to claim 5, characterized in that: The signal ve3 described in steps 1-5 is represented as follows: ve3 = -(vr1 - vifed).

7. A control method for optimizing the efficiency of a four-switch buck-boost converter according to claim 6, characterized in that: Determining the maximum value of the duty cycle of the switch tubes in the four-switch buck-boost converter in step 2 includes: Step 2-1, setting the current threshold vth; Step 2-2, when the inductor current is in the positive direction of the associated direction, that is, the sign of the inductor current vifed signal is positive, the value of the vifed signal satisfies the following conditions: vifed ≥ vth Then the maximum duty cycle D3max of the third switch tube S3 in the four-switch buck-boost converter is set, which is expressed as follows: D3max=Dmax<1 Wherein, Dmax is the maximum duty cycle of the four-switch buck-boost converter; Otherwise, there is no limit on the maximum duty cycle D3max, that is, the maximum duty cycle D3max is 1; Step 2-3, when the inductor current is in the negative direction of the associated direction, that is, the sign of the inductor current vifed signal is negative, when the value of the vifed signal meets the following conditions: vifed≤-vth Then the maximum duty cycle D1max of the first switch tube S1 in the four-switch buck-boost converter is set, which is expressed as follows: D1max=Dmax<1 Otherwise, there is no limit on the maximum duty cycle D1max, that is, the maximum duty cycle D1max is 1.

8. The control method for optimizing the efficiency of a four-switch buck-boost converter according to claim 7, characterized in that: The calculations described in step 3 and the generation of control signals include: Step 3-1, limiting the first input signal vpid1 of the nonlinear controller using the interval [0, D1max], specifically includes: When vpid1>D1max, let vpid1=D1max; When vpid1<0, set vpid1=0; Step 3-2, limiting the second input signal vpid2 of the nonlinear controller using the interval [0, D3max], specifically includes: When vpid2>D3max, let vpid2=D3max; When vpid2<0, set vpid2=0; Step 3-3, calculating the PWM control signal PWM_D1 of the first switch tube with a duty cycle of D1 according to the value of the first input signal vpid1 of the limited nonlinear controller, and its inverse signal is the PWM control signal PWM_D2 of the second switch tube with a duty cycle of D2; Step 3-4, calculate the PWM control signal PWM_D3 of the third switch tube with a duty cycle of D3 according to the value of the second input signal vpid2 of the limited nonlinear controller, and its inverse signal is the PWM control signal PWM_D4 of the fourth switch tube with a duty cycle of D4.

9. A control method for optimizing the efficiency of a four-switch buck-boost converter according to claim 8, characterized in that: The step 3-3 of calculating the PWM control signal PWM_D1 of the first switch tube with a duty cycle of D1 according to the value of the first input signal vpid1 of the nonlinear controller after limiting specifically includes: The value of the first input signal vpid1 of the nonlinear controller after limiting is used as the modulation wave value, and is compared with the triangular carrier Tsw to obtain the PWM control signal PWM_D1 of the first switch tube with a duty cycle of D1.

10. The control method for optimizing the efficiency of a four-switch buck-boost converter according to claim 9, characterized in that: The step 3-4 of calculating the PWM control signal PWM_D3 of the third switch tube with a duty cycle of D3 according to the value of the second input signal vpid2 of the nonlinear controller after limiting specifically includes: The value of the second input signal vpid2 of the nonlinear controller after limiting is used as the modulation wave value, and is compared with the triangular carrier Tsw to obtain the PWM control signal PWM_D3 of the third switch tube with a duty cycle of D3.

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