Symbol synchronization method suitable for THP wired communication system

Abstract

The invention relates to the field of digital signal processing, in particular to a symbol synchronization method suitable for a THP wired communication system, which comprises the following steps of: preprocessing a baseband receiving symbol by using a feed-forward transverse filter (FFE), performing phase discrimination processing on the symbol processed by the FFE, obtaining an estimated value of the current receiving symbol through modulus and judgment in a phase discrimination processing part, and performing signal synchronization on the estimated value. Calculating an error between the received symbol and the estimated value, performing threshold comparison and compensation on the error, and correlating the compensated signal with an expected symbol to obtain sampling phase deviation information; and finally, the sampling deviation information is processed by using a direct path and an integral path of the loop filter and then is used for controlling the NCO to adjust the next sampling position. By processing the sampling phase error information in the linear domain, the influence of nonlinear modulus operation of THP on the phase discrimination algorithm is avoided, and low calculation complexity is maintained while the stability of the symbol synchronization method is improved.

Description

Technical Field

[0001] The present invention relates to the field of digital signal processing. For a wired communication system using THP (T-H precoder), such as SHDSL, 10Gbase-T, etc., a symbol synchronization method is proposed, which has the characteristics of strong stability and low complexity. Background Art

[0002] In a communication system, symbol synchronization is one of the key links; due to factors such as line shape changes and temperature differences in the wired channel, clock jitter and offset occur, and symbol synchronization processing needs to be continuously performed so that the receiving end can lock the peak position of the baseband symbol waveform to achieve the best sampling effect. In particular, a wired communication system using a high-order baseband modulation method, such as 16-PAM, 32-PAM, etc., is more sensitive to the symbol synchronization effect and is more likely to cause symbol decision errors due to sampling position deviation.

[0003] The wired channel is a typical band-limited channel with serious waveform distortion, and an equalizer needs to be introduced to handle inter-symbol interference (ISI). Wired communication systems such as 10Gbase-T and SHDSL use THP (T-H precoder) technology as a time-domain equalizer. While this technology avoids the delay and error propagation problems of the DFE at the receiving end, due to the non-linear modulo operation performed by THP, the symbol synchronization method requires additional design. In addition, considering reducing hardware costs and computational complexity, the symbol synchronization method should minimize the number of sampling points required per symbol period. The classic M-M algorithm extracts the sampling phase deviation by comparing the ISI values at one symbol period ahead of the sampling position and one period behind the sampling position, and can complete synchronization with only the sampling rate equal to the baud rate, but cannot be directly applied to wired communication systems that require time-domain equalization. Scholars such as S.Haar, D.Daecke, and R.Zukunft proposed an M-M method applicable to wired communication systems with time-domain equalizers for symbol synchronization in the literature "Equalizer-based symbol-rate timing recovery for digital subscriber line systems", but did not consider the impact of the use of THP on the symbol synchronization method. Y.R.Chien, W.L.Mao, and H.W.Tsao proposed a symbol synchronization method applicable to THP in the literature "A Novel Baud-Rate Timing Error Detector Design for Baseband Transmission System Using Tomlinson-Harashima Precoder", but this method is expectation-dependent, and the variance of the phase detector output is large, affecting the stability of the feedback loop and reducing the tracking range of the clock offset. Therefore, it is necessary to study a symbol synchronization method with a more stable phase detector output distribution, lower complexity, and optimized tracking range. Summary of the Invention

[0004] In view of this, the present invention proposes a symbol synchronization method for a wired communication system applicable to THP, which improves its stability while maintaining low complexity.

[0005] The technical solution adopted by the present invention is as follows:

[0006] A symbol synchronization method for a wired communication system applicable to THP, comprising the following steps:

[0007] Step 1, preprocess the baseband received symbols, and use a feedforward transversal filter (FFE) during processing to handle the precursor inter-symbol interference (ISI) caused by subsequent symbols to the current symbol;

[0008] Step 2: Perform phase discrimination on the preprocessed baseband received symbols, and calculate the sampled phase deviation information;

[0009] Step 3: Process the sampled deviation information using the direct path and the integral path of the loop filter, and the loop filter outputs θ out [n];

[0010] Step 4: Calculate the sampling position according to the loop filter output θ out [n] to control the moment when the numerically controlled oscillator NCO performs the next sampling.

[0011] Furthermore, the process of using the feedforward transversal filter to preprocess the baseband received symbols in Step 1 includes:

[0012] Suppose the received baseband symbol sampled at time t[n] = (nT s +φ) is y[n], and y[n] passes through a feedforward transversal filter (FFE) with a tap length of M1, and the output is

[0013]

[0014] where y[n] represents the received baseband symbol sequence, ω[n] is the tap coefficient vector of the FFE, * is the convolution operation, is the output of the feedforward transversal filter, and the order M1 of the FFE is determined by the length of the previous ISI of the specific channel.

[0015] Furthermore, the process of performing phase discrimination on the preprocessed baseband received symbols in Step 2 and extracting the sampled phase deviation information from the preprocessed received symbols includes:

[0016] Step 2.1: Take the modulus of the output of the feedforward transversal filter to make map to the range of the original PAM symbol set used in the baseband of the communication system, and obtain

[0017]

[0018] where is the output of the modulus operator, is the output of the FFE in Step 1, and [-M, M] is the value range of the original PAM symbol;

[0019] Step 2.2: Use a decision device to make a decision on the output of the modulus operator to obtain the estimated value of the current received symbol and calculate the error value e a [n]:

[0020]

[0021] Wherein, is the estimated value of the current symbol, decide() is the nearest symbol decision operation with the original PAM symbol set as the decision reference, and e a [n] is the error between the actually received symbol and the estimated symbol;

[0022] Step 2.3, Subtract from the expected received symbol error e c [n] = e a [n] to obtain the expected received symbol. Subtract the expected received symbol from the expected received symbol two symbol periods ago and then take the reciprocal to obtain Compare with the threshold value. If is less than the threshold value, the phase discrimination output is invalid, χ[n] = 0; if is greater than or equal to the threshold value τ 0 , then the phase discrimination output is valid, and χ[n] is the ratio of the error e c [n - 1] passing through a delay unit to :

[0023]

[0024] Wherein, χ[n] is the phase discrimination output, that is, the estimation of the sampling position deviation; is the expected received signal, is the expected received symbol two symbol periods ago, and e c [n - 1] is the expected received symbol error one symbol period ago, and τ 0 is the threshold value.

[0025] Furthermore, in step 2.3, the expected received symbol two symbol periods ago is implemented by a delay device composed of a shift register, and the symbol error e c [n - 1] one symbol period ago is implemented by a delay device composed of a shift register.

[0026] Further, the process of using the direct path and the integral path of the loop filter to process the sampling deviation information in step 3 includes:

[0027] Multiply the phase discrimination output χ[n] by the coefficient α 1 in the direct path to obtain the direct path output θ out1 [n], θ out1 [n] = α 1 ·χ[n];

[0028] Multiply the phase discrimination output χ[n] by the coefficient α 2After multiplication, add it to the output θ of the loop filter integration path in the previous symbol period out2 [n - 1] to obtain θ out2 [n], where θ out2 [n]=α 2 ·χ[n]+θ out2 [n - 1];

[0029] Take the sum of the direct path output and the integration path output as the loop filter output θ out [n], where θ out [n]=θ out1 [n]+θ out2 [n].

[0030] Furthermore, the process of calculating the sampling position based on the loop filter output θ out [n] to control the moment of the next sampling of the numerically controlled oscillator NCO is as follows:

[0031] t[n + 1]=t[n]+T s +θ out [n]

[0032] In the formula, t[n + 1] is the next sampling moment, T s is the baseband symbol period, and θ out [n] is the output of the current received symbol loop filter.

[0033] The symbol synchronization method proposed by the present invention first pre - processes the baseband received symbols using a feed - forward transversal filter FFE to eliminate the pre - cursor inter - symbol interference ISI included in the received symbol sampling values; then performs phase discrimination processing on the symbols after FFE processing. In the phase discrimination processing part, the estimated value of the current received symbol is obtained through modulo - taking and decision - making, the error between the received symbol and the estimated value is calculated, and the error is subjected to threshold comparison and compensation to obtain the sampling phase deviation information related to the desired symbol after compensation; finally, the direct path and integration path of the loop filter are used to process the sampling deviation information and are used to control the numerically controlled oscillator NCO to adjust the next sampling position.

[0034] Compared with the prior art, the present invention processes the sampling phase error information in the linear domain, avoiding the influence of the non - linear modulo - taking operation of THP on the phase discrimination algorithm; at the same time, threshold comparison and compensation are introduced in the phase discrimination algorithm to reduce the variance of the phase deviation information output by the phase discriminator, maintaining low computational complexity while improving the stability of the symbol synchronization method. Description of the Drawings

[0035] Figure 1 It is the overall flowchart of a symbol synchronization method suitable for THP - based wired communication according to the present invention;

[0036] Figure 2 It is the structure diagram of the loop filter LF in step 3 of the present invention;

[0037] Figure 3 It is the schematic diagram of the original equivalent discrete channel impulse response h total [n] of the embodiment;

[0038] Figure 4 It is the schematic diagram of the total discrete impulse response g 0 [n] after the THP and FFE in the training stage of the embodiment have converged;

[0039] Figure 5 It is the schematic diagram of the total discrete impulse response g[n] after the deviation of the new sampling time appears in the data stage;

[0040] Figure 6 It is the comparison diagram of the phase discriminator output - sampling time deviation curve of the symbol synchronization method of the present invention and the existing method, where (a) is the comparison of the mean value and (b) is the comparison of the standard deviation;

[0041] Figure 7 It is the tracking process of the symbol synchronization method of the present invention for the +1000ppm clock frequency deviation in the SHDSL channel environment. Specific embodiments

[0042] To better demonstrate the effect of the present invention, the content of the present invention will be described in detail below in conjunction with the drawings and embodiments.

[0043] Before elaborating on the symbol synchronization method provided by the present invention in detail, the principle of the present invention will be expounded as follows:

[0044] Assume that the equivalent channel impulse response is:

[0045] h total (t) = g T (t) * h channel (t) * g R (t)

[0046] In the formula, h channel (t) is the channel impulse response, g T (t) is the impulse response of the transmitter shaping filter, and g R (t) is the impulse response of the receiver matched filter;

[0047] Assume that the sampling time of the nth symbol is (nT s + φ), then the sampling value of the baseband received signal y(t) at the receiver for the nth symbol is:

[0048]

[0049]

[0050] In the formula, T s is the symbol period, φ is the sampling time deviation caused by various interference factors, a[n] is the transmitted symbol sequence, n(t) is the noise, and * represents the convolution operation;

[0051] Define the equivalent discrete channel impulse response h total,φ [n]:

[0052] h total,φ [0] = h total,φ (φ), h total,φ [1] = h total,φ (T s + φ), h total,φ [2] =

[0053] h total,φ (2T s + φ), h total,φ [-1] = h total,φ (-T s + φ), …… and so on, then there is:

[0054]

[0055] Common physical media of wired channels, including twisted pairs, covered wires, coaxial cables, etc., generally have a band-limiting effect, showing time-domain distortion and causing ISI. h total,φ [n] ≠ impulse function δ[n], as Figure 3 shown. In the above summation formula, the term with k = n is the value of the nth symbol itself at the receiving end, the terms with k < n represent the backward ISI superimposed by the previous symbols on the nth symbol, and the terms with k > n represent the forward ISI superimposed by the subsequent symbols on the nth symbol. y n [n] is the discrete additive noise after sampling of the additive Gaussian noise n(t).

[0056] It can be seen from the above formula that h total,φ [n] is related to φ. Suppose the sampling time is (nT s + φ 0 ) after the symbol synchronization is initially carried out in the channel training stage. Then, after the THP and the feed-forward transversal filter FFE converge through the adaptive algorithm, it satisfies:

[0057]

[0058] In the formula is the total impulse response of THP, the equivalent discrete channel, and FFE. h THP [n] is the impulse response of THP, h FFE[n] is the impulse response of the FFE. Since the parameters of the THP and FFE are fixed, if no new sampling time deviation occurs during the data phase after the channel training phase, the ISI can still be equalized by the THP and FFE at this time, and the output of the feedforward transversal filter FFE does not contain ISI, and has a relationship with the transmitted symbol a[n] that is an integer multiple of 2M plus an additive noise:

[0059]

[0060] where d[n] = 2kM, is the integer multiple (2M) addition and subtraction of the modulo operation of mod(2M), and c n [n] is the discrete additive noise after y n [n] passes through the FFE.

[0061] Define that in the ideal symbol synchronization state, that is, in the state where no new sampling time deviation occurs during the data phase after the channel training phase, the ideal output of the FFE is c[n]:

[0062] c[n] = a[n] + d[n]

[0063] In actual situations, the receiving end cannot predict the actual transmitted symbol a[n], so the decision value is used to replace a[n]. In the case of no decision error, at this time, the ideal output of the FFE at the receiving end is:

[0064]

[0065] Define the expected received symbol error e c [n] as the error between the symbol actually equalized by the FFE and the ideal FFE output:

[0066]

[0067] When a new sampling time deviation occurs during the data phase, let the sampling time be (nT s + φ 0 + φ). Since the parameters of the THP and FFE remain fixed while at this time, the THP and FFE cannot completely eliminate the ISI, and the total response g φ [n] is no longer the impulse function δ[n]:

[0068]

[0069] Figures 3 - 5 shows the above effect, Figure 3 which is the equivalent discrete channel impulse response As can be seen from the figure, there is inter-symbol interference (ISI) caused by waveform distortion in the wired channel. Figure 3 is not the impulse function δ[n]; Figure 4 is the total impulse response including THP, channel, and FFE after the channel training phase is completed It can be seen that at this time, the total impulse response is the impulse function δ[n], that is, THP and FFE can well complete the equalization function; Figure 5 is the total impulse response when a new sampling time deviation φ = 0.1T appears after the channel training phase is completed s It can be seen that at this time, the total response g φ [n] is no longer the impulse function δ[n].

[0070] After a new sampling time deviation φ appears in the data phase, includes ISI caused by adjacent symbols. Therefore, the sampling phase deviation information can be extracted from e c [n]. The classical M-M algorithm uses as the phase discrimination output. However, due to the use of THP, modulo operation is required before decision-making, and this non-linear operation makes the classical M-M algorithm no longer feasible. To solve this problem, EDS-MM can be used to optimize the phase discrimination variance output, that is, it is processed based on the least mean square error idea, as follows:

[0071] Suppose that when the sampling phase deviation and frequency deviation are small, there is the following relationship:

[0072]

[0073] In the formula, represents taking the partial derivative of the mean square value E{e c 2 [n]} with respect to the sampling time deviation φ. The meaning of this formula is that when the sampling time deviation φ = 0, the mean square value E{e c 2 [n]} is the smallest.

[0074] Taking as the phase discrimination algorithm, by gradually approaching, the value of φ when E{e c 2 [n]} takes the minimum value, that is, φ = 0, to achieve the purpose of symbol synchronization. However, after processing according to the above steps, the phase discrimination output is a random variable. The specific value of each phase discrimination output is related not only to the current φ but also to multiple symbols before and after ​The influence of the value does not conform to the expected property that the output of the ideal phase discriminator is only a function of the deviation φ at the sampling moment and the variance is as small as possible, which is not conducive to the stable convergence of the feedback loop. Therefore, it is necessary to improve the above phase discrimination method and enhance the stability of the symbol synchronization method, which is specifically described as follows:

[0075] Since e c [n] contains additive noise and ISI from adjacent symbols:

[0076]

[0077] For the case of a small φ, such as the line having a deviation of at most ±32 ppm in the data phase as specified in the SHDSL protocol, Satisfy:

[0078]

[0079] The above equation shows that the ISI part in e c [n] is mainly caused by and rewrite its form as:

[0080]

[0081]

[0082] In the formula, c n [n] is the discrete noise after the received-end time-domain noise n(t) is sampled and processed by FFE, and c n+ISI [n] is the combined noise of the ISI generated by symbols earlier than the previous symbol c[n - 1] and symbols later than the next symbol c[n + 1] on e c [n] and c n [n].

[0083] In the above formula is only affected by φ and is positively correlated with φ. It is the output that the ideal phase discrimination algorithm should have. The large variance of the phase discrimination output of the EDS-MM method is because it is affected by the coefficient (c[n - 1] - c[n + 1]) and the additive noise c n+ISI [n]. When the baseband modulation order is relatively high, the range of (c[n - 1] - c[n + 1]) becomes larger, and this phenomenon becomes more obvious.

[0084] Table 1: Statistical correlation coefficient table of e c [n] and (c[n - 1] - c[n + 1]) in the simulation experiments under different baseband PAM modulation orders

[0085] Baseband PAM modulation order 2 4 8 16 32 64 Correlation coefficient 0.7875 0.7942 0.7931 0.796 0.7942 0.7937

[0086] As can be seen from Table 1, e c[n] and (c[n - 1] - c[n + 1]) have a high correlation. Based on this feature, the improved phase discrimination algorithm is obtained as follows:

[0087]

[0088] In the above formula, since the transmitted symbol is unknown, it is used under the assumption that there is no decision error to replace c[n], and due to causality, each term is uniformly increased by a delay of 1 symbol period.

[0089] When is small, e c [n - 1] is mainly composed of c n+ISI [n - 1], that is, mainly composed of additive noise and ISI caused by symbols farther away, which is interference independent of φ. Therefore, the phase discrimination output is invalid at this time, and χ[n] = 0; when is large enough, e c [n - 1] is mainly composed of and the product of its coefficient independent of φ , so the phase discrimination output at this time is By this method, the randomness of the phase discrimination algorithm is reduced. From Figure 6 a and Figure 6 b, it can be seen that the improved method proposed by the present invention has a smaller standard deviation compared with EDS-MM.

[0090] Embodiment 1

[0091] As Figure 1 shown, the symbol synchronization method applicable to a wired communication system using THP provided in this embodiment includes the following steps:

[0092] Step 1, preprocess the baseband received symbols: use a feedforward filter FFE to process the previous ISI of subsequent symbols in the current symbol. Let the received baseband symbol y[n] sampled at time t[n] = (nT s +φ), y[n] passes through a feedforward transversal filter FFE with a tap length of M1, and the output is

[0093]

[0094] In the formula, y[n] represents the received baseband symbol sequence, ω[n] is the tap coefficient vector of the FFE, * is the convolution operation, is the output of the FFE. The order M1 of the FFE is determined by the length of the previous ISI of the specific channel;

[0095] Step 2, perform phase discrimination to extract sampling phase deviation information from the received symbol error:

[0096] Step 2.1, modulo operation is performed on to map into the range of the original PAM symbol set, obtaining

[0097]

[0098] wherein, is the output of the modulo operator, is the FFE output of Step 1, and [-M, M] is the value range of the original PAM symbol;

[0099] Step 2.2, decision is made on to obtain the estimation and error of the current received symbol. After passing through the decision device, the output is the estimated symbol and the error value e a [n] is calculated:

[0100]

[0101] wherein, is the estimated value of the current symbol, decide() is the nearest symbol decision operation with the original PAM symbol set as the decision reference, and e a [n] is the error between the actual received symbol and the estimated symbol;

[0102] Step 2.3, is subtracted from the error e c [n] = e a [n] of the expected received symbol to obtain the expected received symbol which is then subtracted from the expected received symbol two symbol periods before, and the reciprocal is taken to obtain which is compared with the threshold value τ 0 . If is less than τ 0 , the phase discrimination output is invalid, χ[n] = 0. If is greater than or equal to τ_0, the phase discrimination output is valid, and χ[n] is the ratio of the error e c [n - 1] passing through a delay unit to :

[0103]

[0104] wherein, χ[n] is the phase discrimination output, i.e., the estimation of the sampling position deviation, is the expected received signal. Since the characteristic of the modulo operation in Step 2 is to perform addition and subtraction of integer multiples of 2M on , therefore can be obtained from is obtained by taking the difference with e c [n], and is the expected received symbol two symbol periods before, implemented by a delay element composed of a shift register, e is the symbol error one symbol period before, implemented by a delay element composed of a shift register, τ c [n - 1], and τ 0 is the comparison threshold, which makes the phase discrimination output valid when e c [n - 1] is mainly composed of the previous and next 1 item of ISI related to , and the phase discrimination output is invalid when e c [n - 1] is mainly composed of other previous and next items of ISI unrelated to .

[0105] Step 3, the output of the phase discriminator is processed by a loop filter, as Figure 2 shown. The loop filter includes a direct path and an integration path. The phase discrimination output χ[n] is multiplied by the coefficient α 1 in the direct path to obtain the direct path output θ out1 [n]. χ[n] is multiplied by the coefficient α 2 in the integration path and then added to the previous output θ out2 [n - 1] of the integration path to obtain θ out2 [n]. The output θ out [n] of the loop filter is the sum of the direct path output and the integration path output:

[0106] θ out1 [n] = α 1 ·χ[n]

[0107] θ out2 [n] = α 2 ·χ[n] + θ out2 [n - 1]

[0108] θ out [n] = θ out1 [n] + θ out2 [n]

[0109] where χ[n] is the output of the current symbol phase discriminator, and θ out2 [n - 1] is the output of the integration path of the loop filter in the previous symbol period, implemented by a delay element composed of a shift register.

[0110] Step 4, calculate the next sampling position: obtain the output θ out [n] of the loop filter, and control the time of the next sampling of the numerically controlled oscillator NCO:

[0111] t[n + 1] = t[n] + T s + θ out[n]

[0112] where t[n + 1] is the next sampling time, and T s is the baseband symbol period, and θ out [n] is the output of the current symbol loop filter.

[0113] Applying the symbol synchronization method of this embodiment to the tracking process of a +1000 ppm clock frequency deviation in the SHDSL channel environment, the tracking result is as Figure 7 shown. It can be seen from Figure 7 that the phase deviation gradually tends to 0 and stabilizes.

[0114] It can be understood that the present invention is described by means of some embodiments. Those skilled in the art know that, without departing from the spirit and scope of the present invention, various changes or equivalent replacements can be made to these features and embodiments. Additionally, under the teaching of the present invention, these features and embodiments can be modified to adapt to specific situations and materials without departing from the spirit and scope of the present invention. Therefore, the present invention is not limited by the specific embodiments disclosed herein, and all embodiments falling within the scope of the claims of this application belong to the scope protected by the present invention.

Claims

1. A symbol synchronization method for a wired communication system applicable to THP A symbol synchronization method for a wired communication system applicable to THP, characterized in that: The following steps are involved: Step 1: pre-process the baseband received symbols, and use a feed-forward transverse filter to process the preceding inter-symbol interference caused by the subsequent symbols to the current symbol; Step 2: performing phase discrimination on the pre-processed baseband received symbols to calculate sampling phase deviation information; Step 3: Use the direct path and integral path of the loop filter to process the sampling deviation information, and the loop filter output θ out [n]; Step 4: According to the loop filter output θ out [n] Calculate the sampling position to control the time when the numerically controlled oscillator NCO performs the next sampling.

2. A symbol synchronization method for a wired communication system suitable for THP according to claim 1, characterized in that: The process of using a feedforward transverse filter to preprocess the baseband received symbols in step 1 includes: Assume that at time t[n] = (nT s +φ) is sampled as the received baseband symbol y[n], which is outputted by a feedforward transversal filter (FFE) with a tap length of M1. Where y[n] represents the received baseband symbol sequence, ω[n] is the tap coefficient vector of FFE, and * is the convolution operation. is the output of the feedforward transversal filter, and the order M1 of the FFE is determined by the length of the preceding ISI of the specific channel.

3. A symbol synchronization method for a wired communication system suitable for THP according to claim 2, characterized in that: The process of performing phase discrimination on the preprocessed baseband received symbols and extracting sampling phase deviation information from the preprocessed received symbols in step 2 includes: Step 2.1: Modulo the output of the feedforward transverse filter so that Mapped to the original PAM symbol set used by the communication system baseband, we get In the formula, is the output of the modulo, is the FFE output of step 1, [-M,M] is the value range of the original PAM symbol; Step 2.2: Use the decision device to decide the output of the modulus detector to obtain the estimated value of the current received symbol. And calculate the error value e a [n]: In the formula, is the estimated value of the current symbol, decide() is the nearest symbol decision operation based on the original PAM symbol set as the decision reference, e a [n] is the error between the actual received symbol and the estimated symbol after modulo; Step 2.3: The output of the feedforward transverse filter The expected received symbol error e c [n] = e a [n] Subtract to get the expected received symbol Expect to receive symbol The expected received symbol two symbol periods ago After subtracting, take the reciprocal and get Compared with the threshold value, if If the phase detection output is invalid, X[n] = 0; If it is greater than or equal to the threshold value, the phase-comparison output is valid, and χ[n] is the error e after a delay unit. c [n-1] and Ratio of: Where X[n] is the phase-detection output, i.e., the estimation of the sampling position deviation; e c [n-1] is the symbol error before one symbol period, and τ0 is the threshold value.

4. The symbol synchronization method for a wired communication system suitable for THP according to claim 3, characterized in that: The expected received symbol two symbol periods before in step 2.3 The delay device composed of shift registers is used to implement the symbol error e before one symbol period. c [n-1] is implemented by a delay device composed of a shift register. Furthermore, the process of using the direct path and the integral path of the loop filter to process the sampling deviation information in step 3 includes: The phase detection output X[n] is multiplied by the coefficient α1 in the direct path to obtain the direct path output θ out1 [n], θ out1 [n] = α1·χ[n]; after multiplying the phase detection output χ[n] by the coefficient α2 in the integral path, it is combined with the output θ of the loop filter integral path of the previous symbol period. out2 [n-1] add, and we get θ out2 [n], θ out2 [n] = α2·χ[n] + θ out2 [n-1]; The sum of the direct path output and the integral path output is taken as the loop filter output θ out [n], θ out [n] = θ out1 [n]+θ out2 [n].

5. The symbol synchronization method for a wired communication system suitable for THP according to claim 4, characterized in that: The loop filter output θ out [n] The process of calculating the sampling position to control the time when the numerically controlled oscillator NCO performs the next sampling is: t[n+1]=t[n]+T s +θ out [n] Where t[n+1] is the next sampling time, T s is the baseband symbol period, θ out [n] is the loop filter output of the current received symbol.

Images