A symbol synchronization method for a wired communication system suitable for THP

Abstract

The application relates to the field of digital signal processing, in particular to a symbol synchronization method suitable for a wired communication system of THP, a feedforward transversal filter FFE is used to pre-process baseband receiving symbols, phase discrimination processing is carried out on the symbols processed by the FFE, in the phase discrimination processing part, an estimated value of the current receiving symbol is obtained through modulo operation and decision, an error between the receiving symbol and the estimated value is calculated, threshold comparison and compensation are carried out on the error, sampling phase deviation information is obtained by correlating the compensated error with an expected symbol; finally, the sampling deviation information is processed by using a direct path and an integral path of a loop filter, and is used for controlling a numerically controlled oscillator NCO to adjust a next sampling position. The sampling phase error information is processed in a linear domain, the influence of nonlinear modulo operation of THP on the phase discrimination algorithm is avoided, the stability of the symbol synchronization method is improved, and low calculation complexity is maintained.

Description

TECHNICAL FIELD

[0001] The present application relates to the field of digital signal processing, and provides a symbol synchronization method for a wired communication system using a THP (T-H precoder), such as SHDSL, 10Gbase-T, etc. BACKGROUND

[0002] In a communication system, symbol synchronization is one of the key links. Due to clock jitter and offset caused by factors such as line form change and temperature difference of a wired channel, symbol synchronization needs to be continuously processed, so that the receiving end can lock the peak position of a baseband symbol waveform and achieve the best sampling effect. In particular, a wired communication system using a high-order baseband modulation mode, such as 16-PAM, 32-PAM, etc. is more sensitive to symbol synchronization effect and more likely to cause symbol decision errors due to sampling position deviation.

[0003] The wired channel is a typical band-limited channel, and there is serious waveform distortion, and an equalizer needs to be introduced to process inter-symbol interference (ISI). The wired communication system such as 10Gbase-T and SHDSL adopts THP (T-H precoder) technology as a time domain equalizer. This technology avoids the time delay and error propagation problem of the DFE at the receiving end, but due to the nonlinear operation of the modulo operation by THP, the symbol synchronization method needs to be additionally designed. In addition, in order to reduce the hardware cost and the calculation complexity, the symbol synchronization method should reduce the number of sampling points required for each symbol period as much as possible. The classic M-M algorithm extracts the sampling phase deviation by comparing the ISI values of the leading sampling position and the lagging sampling position of one symbol period, and only needs a sampling rate same as the baud rate to complete synchronization, but it cannot be directly applied to the wired communication system which needs time domain equalization. S. Haar, D. Daecke and R. Zukunft and other scholars proposed an M-M method suitable for wired communication systems containing time domain equalizers in the document “Equalizer-based symbol-rate timing recovery for digital subscriber line systems”, which is used for symbol synchronization, but does not consider the influence of the use of THP on the symbol synchronization method. Y. R. Chien, W. L. Mao and H. W. Tsao proposed a symbol synchronization method suitable for THP in the document “A Novel Baud-Rate Timing Error Detector Design for Baseband Transmission System Using Tomlinson-Harashima Precoder”, but this method is expectation-dependent, and the output variance of the phase detector is large, which affects the stability of the feedback loop and reduces the tracking range of the clock offset. Therefore, it is necessary to study a symbol synchronization method with more stable phase detector output distribution, lower complexity and optimized tracking range. SUMMARY

[0004] In view of this, the present application proposes a symbol synchronization method suitable for a wired communication system with THP, which improves its stability while maintaining low complexity.

[0005] The technical scheme adopted by the present application is:

[0006] A symbol synchronization method suitable for a wired communication system with THP, comprising the following steps:

[0007] Step 1, pre-processing the baseband received symbol, using a feedforward transversal filter (FFE) to process the previous inter-symbol interference (ISI) caused by the subsequent symbol to the current symbol;

[0008] Step 2, phase detection is performed on the pre-processed baseband received symbol, and sampling phase deviation information is calculated;

[0009] Step 3, the sampling deviation information is processed using the direct path and integral path of the loop filter, and the loop filter outputs θ out [n];

[0010] Step 4, the sampling position is calculated according to the loop filter output θ out [n], to control the time of the next sampling of the numerically controlled oscillator NCO.

[0011] Further, the step 1 includes the process of pre-processing the baseband received symbol using the feedforward transversal filter, which comprises:

[0012] Suppose that the received baseband symbol sampled at time t[n] = (nT s + φ) is y[n], and y[n] is processed by the feedforward transversal filter (FFE) with a tap length of M1, and the output is

[0013]

[0014] In the formula, y[n] represents the received baseband symbol sequence, ω[n] is the tap coefficient vector of the FFE, * is the convolution operation, is the output of the feedforward transversal filter, and the order M1 of the FFE is determined by the pre-ISI length of the specific channel.

[0015] Further, the step 2 includes the process of phase detection on the pre-processed baseband received symbol to extract the sampling phase deviation information from the pre-processed received symbol, which comprises:

[0016] Step 2.1, the output of the feedforward transversal filter is taken modulo, so that is mapped to the range of the original PAM symbol set used in the baseband of the communication system, and

[0017]

[0018] In the formula, is the output of the modulo operator, is the output of the FFE in step 1, and [-M, M] is the value range of the original PAM symbol;

[0019] Step 2.2, the output of the modulo operator is judged using a decision device, and the estimated value of the current received symbol is obtained and the error value e a [n] is calculated.

[0020]

[0021] In the formula, The value is the estimated value of the current symbol. `decide()` is the nearest symbol decision operation performed with the original PAM symbol set as the decision reference. a [n] represents the error between the actual received symbol and the estimated symbol;

[0022] Step 2.3, With expected received symbol error e c [n] = e a Subtracting [n] yields the expected received symbol. Subtracting the expected received symbol from the expected received symbol two symbol periods ago and taking the reciprocal, we get... Compared with the threshold value, if If the value is less than the threshold, the phase detection output is invalid, and χ[n] = 0; if If the phase detection output is greater than or equal to the threshold value τ0, then the phase detection output is valid, and χ[n] is the error e after one delay unit. c [n-1] and The ratio:

[0023]

[0024] In the formula, χ[n] is the phase detection output, which is the estimate of the sampling position deviation; In order to receive the signal, For the expected received symbol two symbol periods prior, e c [n-1] represents the expected received symbol error one symbol period prior, and τ0 is the threshold.

[0025] Furthermore, the expected received symbol two symbol periods prior to step 2.3. Implemented by a delay unit composed of shift registers, the symbol error e before one symbol period is... c [n-1] is implemented using a delay circuit consisting of shift registers.

[0026] Furthermore, step 3, which involves processing the sampling deviation information using the direct path and integral path of the loop filter, includes:

[0027] The phase detection output χ[n] is multiplied by the coefficient α1 in the direct path to obtain the direct path output θ. out1 [n], θ out1 [n] = α1·χ[n];

[0028] The phase detection output χ[n] is multiplied by the coefficient α2 in the integration path and then combined with the output θ of the loop filter integration path of the previous symbol period. out2 Adding [n-1] together yields θ. out2 [n], θout2 [n] = a2x[n] + θ out2 [n-1];

[0029] The sum of the direct path output and the integral path output is taken as the loop filter output θ out [n], θ out [n] = θ out1 [n] + θ out2 [n].

[0030] Further, the process of calculating the sampling position according to the loop filter output θ out [n] to control the time of the next sampling of the numerically controlled oscillator NCO is as follows:

[0031] t[n+1] = t[n] + T s + θ out [n]

[0032] In the formula, t[n+1] is the time of the next sampling, T s is the baseband symbol period, θ out [n] is the current received symbol loop filter output.

[0033] The symbol synchronization method provided by the application first uses the feedforward transverse filter FFE to pre-process the baseband received symbol to eliminate the previous intersymbol interference ISI contained in the sampling value of the received symbol; then performs phase discrimination processing on the symbol processed by the FFE, obtains the estimated value of the current received symbol through the modulo operation and decision in the phase discrimination processing part, calculates the error between the received symbol and the estimated value, and performs threshold comparison and compensation on the error, obtains the sampling phase deviation information related to the expected symbol after compensation; finally, the sampling deviation information is processed by the direct path and the integral path of the loop filter, and is used to control the numerically controlled oscillator NCO to adjust the position of the next sampling.

[0034] Compared with the prior art, the application processes the sampling phase error information in the linear domain, avoids the influence of the nonlinear modulo operation of the THP on the phase discrimination algorithm, introduces threshold comparison and compensation in the phase discrimination algorithm to reduce the variance of the phase deviation information output by the phase discriminator, improves the stability of the symbol synchronization method, and maintains low computational complexity. BRIEF DESCRIPTION OF DRAWINGS

[0035] Figure 1 is a whole flowchart of the symbol synchronization method suitable for THP wired communication according to the application;

[0036] Figure 2 is a structure diagram of the loop filter LF in step 3 according to the application;

[0037] Figure 3Figure 1 is a schematic diagram of the original equivalent discrete channel impulse response h for an embodiment total Figure 1 is a schematic diagram of the original equivalent discrete channel impulse response h for an embodiment

[0038] Figure 4 Figure 2 is a schematic diagram of the total discrete impulse response g0[n] after the training phase THP and FFE have converged for an embodiment

[0039] Figure 5 Figure 3 is a schematic diagram of the total discrete impulse response g[n] after a new sampling time offset occurs during the data phase for an embodiment

[0040] Figure 6 Figure 4 is a comparison of the phase discriminator output-sampling time offset curve for the symbol synchronization method of the present application and the prior art, where (a) is a comparison of the mean and (b) is a comparison of the standard deviation

[0041] Figure 7 Figure 5 is a tracking process for a +1000 ppm clock frequency offset for the symbol synchronization method of the present application in a SHDSL channel environment DETAILED DESCRIPTION

[0042] To better illustrate the effects of the present application, the following detailed description of the present application is provided in conjunction with the accompanying drawings and embodiments.

[0043] Before the symbol synchronization method provided by the present application is described in detail, the principles of the present application are first described as follows:

[0044] Let the equivalent channel impulse response be:

[0045] h total (t) = g T (t) * h channel (t) * g R (t)

[0046] In the formula, h channel (t) is the channel impulse response, g T (t) is the transmit end shaping filter impulse response, and g R (t) is the receive end matching filter impulse response.

[0047] Let the sampling time of the nth symbol be (nT s + φ), then the sampling value of the nth symbol of the baseband received signal y(t) at the receive end is:

[0048]

[0049]

[0050] In the formula, T sis the symbol period, φ is the sampling time deviation caused by various interference factors, a[n] is the transmitted symbol sequence, n(t) is the noise, * represents convolution operation;

[0051] Define the equivalent discrete channel impulse response h total,φ [n]:

[0052] h total,φ [0] = h total,φ (φ), h total,φ [1] = h total,φ (T s + φ), h total,φ [2] =

[0053] h total,φ (2T s + φ), h total,φ [-1] = h total,φ (-T s + φ), and so on, then we have:

[0054]

[0055] The common physical media of wired channel, including twisted pair, covered wire, coaxial cable, etc. generally have band-limited effect, showing time domain distortion and leading to ISI, h total,φ [n] ≠ impulse function δ[n], as shown in Figure 3 In the above summation, the term k = n is the value of the nth symbol itself at the receiving end, the term k < n represents the backward ISI of the previous symbol superimposed on the nth symbol, and the term k > n represents the forward ISI of the subsequent symbol superimposed on the nth symbol, y n [n] is the discrete additive noise after sampling of the additive Gaussian noise n(t).

[0056] From the above formula, it can be seen that h total,φ [n] is related to φ. Assuming that the sampling time is (nT s + φ0) after preliminary symbol synchronization in the channel training stage, then after the THP and the feedforward transverse filter FFE are converged by the adaptive algorithm, they satisfy:

[0057]

[0058] In the formula, h is the total impulse response of THP, equivalent discrete channel, and FFE, h THP [n] is the impulse response of THP, h FFE [n] is the impulse response of FFE. Since the parameters of THP and FFE are fixed, if no new sampling time deviation is generated in the data stage after the channel training stage, then at this time, the ISI can still be equalized by THP and FFE, and the output of the feedforward transverse filter FFE without ISI, with the transmitted symbol a[n] with an integer multiple of 2M and additive noise:

[0059]

[0060] where d[n] = 2kM, is an integer multiple of mod(2M) plus or minus, c n [n] is y n [n] is the discrete additive noise after FFE.

[0061] The ideal output of FFE is c[n] under the ideal symbol synchronization state, i.e. the state in which no new sampling time deviation is generated in the data stage after the channel training stage:

[0062] c[n] = a[n] + d[n]

[0063] In actual situations, the actual transmitted symbol a[n] cannot be predicted by the receiving end, and therefore the decision value is used instead of a[n], and in the absence of decision errors, At this time, the ideal output of the receiving end FFE is:

[0064]

[0065] Define the expected received symbol error e c [n] as the error between the actual symbol after FFE equalization and the ideal FFE output:

[0066]

[0067] When a new sampling time deviation occurs in the data stage, let the sampling time be (nT s + φ0+ φ), since the parameters of THP and FFE remain fixed and At this time, THP and FFE cannot completely eliminate ISI, and the total response g φ [n] is no longer an impulse function δ[n]:

[0068]

[0069] Figures 3-5 The above effects are shown, Figure 3 is the equivalent discrete channel impulse response As can be seen from the figure, there is inter-symbol interference (ISI) due to waveform distortion in the wired channel. Figure 3 is not an impulse function δ[n]; Figure 4 is the total impulse response of THP, channel, and FFE after completion of the channel training stage It can be seen that the total impulse response is impulse function δ[n] at this time, that is, THP and FFE can complete the equalization function well. Figure 5 When the new sampling time deviation φ = 0.1T occurs at the data stage after the channel training stage is completed s The total impulse response at this time It can be seen that the total response g φ [n] is no longer an impulse function δ[n].

[0070] When the new sampling time deviation φ occurs at the data stage, The ISI caused by adjacent symbols is contained in e c [n], so the sampling phase deviation information can be extracted from e As a phase discrimination output, but because THP is used, it is necessary to take the modulus before judgment, and this nonlinear operation makes the classical M-M algorithm no longer feasible. To solve this problem, the EDS-MM optimized phase discrimination output can be used, that is, based on the minimum mean square error idea, the specific process is as follows:

[0071] It is assumed that when the sampling phase deviation and the frequency deviation are small, the following relationship exists:

[0072]

[0073] In the formula, represents the partial derivative of the mean square value E{e c 2 [n]} of the expected received symbol error with respect to the sampling time deviation φ, and the meaning of the formula is that when the sampling time deviation φ = 0, the mean square value E{e c 2 [n]} of the expected received symbol error is minimum.

[0074] Taking as a phase discrimination algorithm, the value of φ when E{e c 2 [n]} is minimum is gradually approached, that is, φ = 0, so that the purpose of symbol synchronization is achieved. However, after the above steps are processed, the phase discrimination output is a random variable, and the specific value of each phase discrimination output is related to the current φ, and is also affected by the values of multiple symbols before and after, which is not consistent with the expected property that the output of an ideal phase discriminator is only a function of the sampling time deviation φ and the variance is as small as possible, which is not conducive to the stable convergence of the feedback loop. Therefore, it is necessary to improve the above phase discrimination method and enhance the stability of the symbol synchronization method, and the specific description is as follows:

[0075] Because e c [n] contains additive noise and ISI of adjacent symbols:

[0076]

[0077] For the case of small φ, as specified in SHDSL protocol, the line has at most ±32ppm deviation in data phase, satisfies:

[0078]

[0079] The above equation shows that e c The ISI part in e is mainly caused by

[0080]

[0081]

[0082] where c n is the discrete noise after sampling and FFE processing of the received time-domain noise n(t), c n+ISI is the combined noise of the ISI caused by the symbol earlier than c[n-1] and the symbol later than c[n+1] on e c and c n [n].

[0083] In the above equation, e is only affected by φ and positively related to φ, which is the output of the ideal phase-detecting algorithm. The large variance of the phase-detecting output of EDS-MM method is caused by the influence of the coefficient (c[n-1]-c[n+1]) and the additive noise c n+ISI [n]. When the modulation order of baseband PAM is high, the range of (c[n-1]-c[n+1]) becomes large, and this phenomenon becomes more obvious.

[0084] Table 1: Statistical correlation coefficient table of e c [n] and (c[n-1]-c[n+1]) in simulation experiments under different baseband PAM modulation orders

[0085] Baseband PAM modulation order 2 4 8 16 32 64 Correlation coefficient 0.7875 0.7942 0.7931 0.796 0.7942 0.7937

[0086] From Table 1, it can be seen that e c [n] and (c[n-1]-c[n+1]) have a high correlation. Based on this feature, the improved phase-detecting algorithm is obtained as follows:

[0087]

[0088] In the above equation, since the transmitted symbol is unknown, c Instead of c[n], and due to causality, each term is increased by a delay of one symbol period.

[0089] When is small, e c [n-1] is mainly composed of c n+ISI [n-1], i.e. mainly composed of additive noise and ISI caused by more distant symbols, is a disturbance independent of φ, so the phase detection output is invalid at this time, χ[n] = 0; when is large enough, e c [n-1] is mainly composed of and the product of its coefficient independent of φ , so the phase detection output at this time is By this method, the randomness of the phase detection algorithm is reduced, and Figure 6 a and Figure 6 b show that the improved method proposed in the present application has a smaller standard deviation than EDS-MM.

[0090] Embodiment 1

[0091] As shown in Figure 1 , the symbol synchronization method suitable for a wired communication system using THP provided by the present embodiment comprises the following steps:

[0092] Step 1, pre-processing the baseband received symbol: using a feedforward filter FFE to process the pre-ISI of the subsequent symbol. Let the received baseband symbol y[n] sampled at time t[n] = (nT s + φ) be y[n], y[n] is processed by a feedforward transverse filter FFE with a tap length of M1, and the output is

[0093]

[0094] In the formula, y[n] represents the received baseband symbol sequence, ω[n] is the tap coefficient vector of FFE, * is convolution operation, and y[n] is the output of FFE. The order M1 of FFE is determined by the pre-ISI length of the specific channel;

[0095] Step 2, phase detection, extract the sampling phase deviation information from the received symbol error:

[0096] Step 2.1, take the modulus of e , so that e is mapped into the range of the original PAM symbol set, and e

[0097]

[0098] In the formula, For the output of the mold extractor, The FFE output of step 1, where [-M,M] is the range of values ​​for the original PAM symbol;

[0099] Step 2.2, for A decision is made to obtain an estimate and error of the currently received symbol. After passing through the decision unit, the output is the estimated symbol. And calculate the error value e a [n]:

[0100]

[0101] In the formula, The value is the estimated value of the current symbol. `decide()` is the nearest symbol decision operation performed with the original PAM symbol set as the decision reference. a [n] represents the error between the actual received symbol and the estimated symbol;

[0102] Step 2.3, With expected received symbol error e c [n] = e a Subtracting [n] yields the desired received symbol. Compared to the expected received symbol two symbol periods ago Subtract and take the reciprocal to get Compared with the threshold value τ0, if If the value is less than τ0, the phase detection output is invalid, and χ[n] = 0. If the value is greater than or equal to τ_0, then the phase detection output is valid, and χ[n] is the error e after one delay unit. c [n-1] and The ratio:

[0103]

[0104] In the formula, χ[n] is the phase detection output, which is the estimate of the sampling position deviation. To receive the desired signal, due to the characteristics of the modulo operation in step 2, Based on this, we perform addition and subtraction operations on integer multiples of 2M, therefore can be With e c [n] is obtained by subtracting from it. The expected received symbol two symbol periods in advance is implemented using a delay function consisting of shift registers, e c [n-1] represents the symbol error one symbol period prior, implemented by a delay circuit composed of shift registers, and τ0 is the comparison threshold, which ensures that when e c [n-1] is mainly composed of and The phase-locked output is valid when the relevant pre- and post-1 ISI components are present, e c [n-1] are mainly composed of the unrelated pre- and post-1 ISI components.

[0105] Step 3, the phase detector output is processed by a loop filter, as shown in Figure 2 The loop filter includes a direct path and an integral path. The phase detector output χ[n] is multiplied by a coefficient α1 in the direct path to obtain the direct path output θ out1 [n], and χ[n] is multiplied by a coefficient α2 in the integral path and then added with the last output θ out2 [n-1] of the integral path to obtain θ out2 [n], the loop filter output θ out [n] is the sum of the direct path output and the integral path output:

[0106] θ out1 [n] = α1 · χ[n]

[0107] θ out2 [n] = α2 · χ[n] + θ out2 [n-1]

[0108] θ out [n] = θ out1 [n] + θ out2 [n]

[0109] wherein χ[n] is the current symbol phase detector output, θ out2 [n-1] is the last symbol period loop filter integral path output, which is realized by a delay register.

[0110] Step 4, calculate the next sampling position: obtain the loop filter output θ out [n], control the time of the next sampling of the numerically controlled oscillator NCO:

[0111] t[n+1] = t[n] + T s + θ out [n]

[0112] wherein t[n+1] is the next sampling time, T s is the baseband symbol period, and θ out [n] is the current symbol loop filter output.

[0113] The symbol synchronization method of the embodiment is applied to the SHDSL channel environment, and the tracking process of +1000ppm clock frequency deviation is shown in Figure 7 From Figure 7 it can be seen that the phase deviation gradually tends to 0 and is stable.

[0114] It is to be understood that the present application is described by way of example only, and that modifications or alterations can be made to the features and embodiments described without departing from the spirit and scope of the application. In addition, modifications can be made to the features and embodiments described to accommodate specific situations and materials without departing from the spirit and scope of the application. Accordingly, the application is not limited to the specific embodiments disclosed herein, but rather, the scope of the application includes all embodiments falling within the scope of the claims.

Claims

1. A symbol synchronization method for a wired communication system suitable for THP. The symbol synchronization method for a wired communication system suitable for THP is characterized in that... Includes the following steps: Step 1: Preprocess the baseband received symbols. During processing, a feedforward transverse filter is used to handle the inter-symbol interference caused by subsequent symbols to the current symbol. Step 2: Perform phase detection on the preprocessed baseband received symbols and calculate the sampling phase deviation information; Its implementation process includes: Step 2.1: Take the modulus of the output of the feedforward transverse filter, so that... Mapping to the original PAM symbol set used by the baseband of the communication system, we obtain... In the formula, For the output of the mold extractor, The FFE output of step 1, where [-M,M] is the range of values ​​for the original PAM symbol; Step 2.2: Use the decision unit to make a decision on the output of the mode extractor to obtain the estimated value of the currently received symbol. And calculate the error value e a [n]: In the formula, The value is the estimated value of the current symbol. `decide()` is the nearest symbol decision operation performed with the original PAM symbol set as the decision reference. a [n] represents the error between the actual received symbol modulo the estimated symbol; Step 2.3: Convert the output of the feedforward transverse filter... With expected received symbol error e c [n] = e a Subtracting [n] yields the desired received symbol. Expected to receive symbols Compared to the expected received symbol two symbol periods ago Subtract and take the reciprocal to get Compared with the threshold value, if If the value is less than the threshold, the phase detection output is invalid, and χ[n] = 0; if If the value is greater than or equal to the threshold, the phase detection output is valid, and χ[n] is the error e after one delay unit. c [n-1] and The ratio: In the formula, χ[n] is the phase detection output, i.e., the estimate of the sampling position deviation; e c [n-1] represents the symbol error one symbol period prior, and τ0 is the threshold value; the expected received symbol two symbol periods prior in step 2.

3. Implemented by a delay unit composed of shift registers, the symbol error e before one symbol period is... c [n-1] is implemented using a delay circuit composed of shift registers; Step 3: Process the sampling bias information using the direct path and integral path of the loop filter. The loop filter outputs θ. out [n]; Step 4: Based on the output θ of the loop filter out [n] Calculates the sampling position to control the timing of the next sampling by the numerically controlled oscillator NCO.

2. A symbol synchronization method for a wired communication system suitable for THP according to claim 1, characterized in that, The process of preprocessing the baseband received symbols using a feedforward transverse filter in step 1 includes: Let at time t[n] = (nT) s The received baseband symbol sampled by +φ is y[n]. y[n] passes through a feedforward transverse filter (FFE) with a tap length of M1, and the output is... In the formula, y[n] represents the received baseband symbol sequence, ω[n] is the FFE tap coefficient vector, and * represents the convolution operation. For the output of the feedforward transverse filter, the order M1 of the FFE is determined by the length of the preceding ISI term of the specific channel.

3. The symbol synchronization method for a wired communication system suitable for THP according to claim 2, characterized in that, Step 3, which involves processing the sampling deviation information using the direct path and integral path of the loop filter, includes: The phase detection output χ[n] is multiplied by the coefficient α1 in the direct path to obtain the direct path output θ. out1 [n], θ out1 [n] = α1·χ[n]; After multiplying the phase detection output χ[n] by the coefficient α2 in the integration path, it is then combined with the output θ of the loop filter integration path of the previous symbol period. out2 Adding [n-1] together yields θ. out2 [n], θ out2 [n] = α²·χ[n] + θ out2 [n-1]; The sum of the direct path output and the integral path output is used as the loop filter output θ. out [n], θ out [n] = θ out1 [n]+θ out2 [n].

4. The symbol synchronization method for a wired communication system suitable for THP according to claim 3, characterized in that, The output θ based on the loop filter out The process of calculating the sampling position [n] to control the timing of the next sampling by the numerically controlled oscillator (NCO) is as follows: t[n+1]=t[n]+T s +θ out [n] In the formula, t[n+1] is the next sampling time, and T s For the baseband symbol period, θ out [n] represents the output of the current received symbol loop filter.

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