High-frequency torque ripple suppression method for electrically-driven reconfiguration type vehicle-mounted charging system

Abstract

The invention discloses a high-frequency torque ripple suppression method for an electrically-driven reconfiguration type vehicle-mounted charging system, and the method comprises the steps: firstly, building a mathematical expression formula of a phase current of the system in a charging mode; secondly, a high-precision torque model of the motor is established, and a high-frequency torque component mathematical expression formula is obtained; thirdly, calculating a current value needing to be compensated for by a d axis according to the high-frequency torque component; then, the d-axis current compensation value, the current reference value and the actual value serve as input of a d-axis current feedback loop; and finally, realizing high-frequency torque ripple suppression of the system in a charging mode through a vector control and feedforward control scheme. According to the method, under the condition that electromagnetic torque is generated after zero-sequence current injection is considered, the torque generation mechanism is analyzed, a torque ripple suppression scheme is designed based on a torque component formula, the compensation component is injected into the subspace current, the torque ripple of the vehicle-mounted charger during charging is effectively suppressed, and the system safety is improved.

Description

Technical Field

[0001] The present invention relates to a method for suppressing high-frequency torque ripple of an electric drive reconstruction type in-vehicle charging system. Background Art

[0002] The electric drive reconstruction type in-vehicle charging system provides a new lightweight solution for the existing in-vehicle charging system. By using the original electric drive unit to realize the functions of electric drive and charging, the manufacturing cost can be reduced. However, its endurance ability remains the key concern. Based on this, introducing a new energy port to enable the electric vehicle to charge while driving, thereby improving the endurance ability, has become the mainstream research direction of electric vehicles at present. However, the charging current of the electric drive reconstruction type in-vehicle system with multiple energy ports adopts the zero-sequence current injection method, resulting in high-frequency ripple in the motor current and electromagnetic torque, thereby reducing the motor performance and increasing the power loss. Therefore, suppressing the high-frequency torque ripple of the motor has important application value.

[0003] Currently, in the torque ripple minimization technology, the torque ripple minimization control method that does not require modifying the motor body design is the current mainstream solution. However, there are few analysis and suppression schemes for the high-frequency torque ripple of the in-vehicle charging system, and the generation cause of the high-frequency torque ripple has not been analyzed, and there are contradictions in electric drive control and charging control. Summary of the Invention

[0004] Object of the Invention: Aiming at the above-mentioned existing technologies, a method for suppressing high-frequency torque ripple of an electric drive reconstruction type in-vehicle charging system is proposed to effectively suppress the torque ripple of the in-vehicle charger during charging.

[0005] Technical Problem: A method for suppressing high-frequency torque ripple of an electric drive reconstruction type in-vehicle charging system includes:

[0006] Step 1: Establish a mathematical expression formula of the phase current of the system in the charging mode according to the vector space decoupling matrix;

[0007] Step 2: Establish a high-precision torque model of the motor by the method of co-energy and obtain the mathematical expression formula of the high-frequency torque component;

[0008] Step 3: Compensate the high-frequency torque by the reluctance torque of the motor, and then calculate the current component to be compensated on the d-axis;

[0009] Step 4: Take the d-axis current compensation value, current reference value and actual value as the inputs of the current feedback loop, and adjust through a PI controller to obtain the d-axis reference voltage;

[0010] Step 5: Realize the suppression of high-frequency torque ripple of the system in the charging mode through a vector control and feed-forward control scheme.

[0011] Furthermore, the electric drive reconfiguration type vehicle-mounted charging system adopts a six-phase symmetrical permanent magnet synchronous motor. In the vector control scheme of the system, the output torque of the motor is adjusted by controlling the q-axis current, and the charging power of the battery is adjusted by controlling the zero-sequence current of the 02 axis. The zero-sequence current I 02 has a charging current ripple. Combining with the vector space decoupling matrix, the mathematical expression of the phase current of the system in the charging mode is established as follows:

[0012]

[0013] In the formula, I A 、I B 、I C 、I U 、I V 、I W are the phase currents of the six-phase symmetrical permanent magnet synchronous motor respectively, I q 、I 02 are the q-axis current component and the zero-sequence current component of the 02 axis respectively, θ e is the electrical angle of the motor rotor, g X (t) is the high-frequency charging current ripple of each phase of the motor within a switching period, X ∈ {A, B, C, U, V, W}, and t represents time; g X (t) is specifically expressed as:

[0014]

[0015] In the formula, V dc is the DC source voltage, V b is the battery voltage, L X is the inductance of winding X, D X is the duty cycle of the bridge arm corresponding to winding X, T s is the switching period.

[0016] Furthermore, in step 2, under the condition of charging while driving, due to the injection of the zero-sequence current I 02 of the 02 axis, a DC bias is generated in the phase flux linkage, and the zero-sequence current I 02 of the 02 axis is related to the third harmonic flux linkage. Therefore, the flux linkage ψ X of each phase of the motor consists of a fundamental wave flux linkage, a third harmonic flux linkage, and a DC component. The expression of the flux linkage of each phase under the charging condition is obtained by using the finite element method, and then a high-precision torque mathematical model of the motor is established through the co-energy method as:

[0017] T e =3p n [I q ψ f1 +I q I d (L d -Lq ) + T0 (3)

[0018] Wherein, T e is the electromagnetic torque of the motor, p n is the number of pole pairs of the motor, I d is the d-axis current component, L d and L q are the d-axis and q-axis inductances of the motor respectively, ψ f1 is the fundamental wave amplitude of the phase magnetic flux linkage, and T0 is the high-frequency torque caused by the high-frequency charging current pulsation generated after the injection of the 02-axis zero-sequence current I 02 :

[0019] T0 = 3p n K(t)ψ f3 sin3θ e (4)

[0020] Wherein, ψ f3 is the third harmonic amplitude of the phase magnetic flux linkage, and K(t) is the sum of the components of each phase current under the charging condition.

[0021] Furthermore, in step 3, the current component to be compensated on the d-axis is calculated according to the high-frequency torque T0, that is, the reluctance torque T generated by the d-axis current er is used to suppress the high-frequency torque pulsation caused by the zero-sequence current, and the compensation value of the injected d-axis current is expressed as:

[0022]

[0023] Beneficial effects: Considering the electromagnetic torque generated after the injection of the zero-sequence current, the present invention analyzes the mechanism of torque generation, and designs a torque pulsation suppression scheme based on the torque component formula. By injecting the compensation component of the subspace current, the torque pulsation of the on-vehicle charger during charging is effectively suppressed, and the system safety is improved. Brief Description of the Drawings

[0024] Figure 1 is a simplified circuit topology schematic diagram of the electric drive reconfigured on-vehicle charging system in the present invention;

[0025] Figure 2 is a control block diagram of a method for suppressing high-frequency torque pulsation of an electric drive reconfigured on-vehicle charging system in the present invention;

[0026] Figure 3 is an equivalent circuit schematic diagram of the DC charging mode of the electric drive reconfigured on-vehicle charging system in the present invention;

[0027] Figure 4 is a schematic diagram of the result of not suppressing the high-frequency torque in the present invention;

[0028] Figure 5Schematic diagram of the high-frequency torque suppression result of the present invention; Specific embodiments

[0029] The present invention will be further described in detail below in conjunction with the accompanying drawings and through embodiments. The following embodiments are explanations of the present invention, and the present invention is not limited to the following embodiments.

[0030] A method for suppressing high-frequency torque ripple in an electric drive reconfiguration type on-vehicle charging system. The circuit topology is as Figure 1 shown. The DC source is connected between the neutral point of the six-phase symmetrical permanent magnet synchronous motor winding and the negative pole of the battery. At this time, the two neutral points O1 and O2 of the six-phase symmetrical permanent magnet synchronous motor are connected together. As Figure 2 shown, this method builds a high-precision torque model through the actual expression of the phase current and the actual flux linkage matrix, and calculates the current component required to be compensated on the d-axis according to the high-frequency torque component; then, the d-axis current compensation value, the current reference value, and the actual value are used as the inputs of the current feedback loop; finally, the high-frequency torque component in the motor is suppressed through feedforward compensation control and vector control.

[0031] This method includes the following specific steps:

[0032] Step 1: Establish the mathematical expressions of the phase current and each space current in the charging mode through the vector space decoupling matrix, as follows:

[0033] Calculate the motor phase currents I A , I B , I C , I U , I V , I W and the subspace currents I ɑ , I β , I x , I y , I 01 , I 02 mathematical relationship:

[0034]

[0035] Among them, I X is the phase current of winding X, X ∈ {A, B, C, U, V, W}, I ɑ , I β respectively represent the ɑ-axis and β-axis current components of the motor in the stationary coordinate system, I x , I y respectively represent the current components of the motor in the x-axis and y-axis directions, I 01 , I 02 respectively represent the zero-sequence current components of the motor on the 01-axis and 02-axis.

[0036] Taking the phase A current as an example, its specific expression is as follows:

[0037] I A = I α + I x + I 01 + I 02 (2)

[0038] In the vector control scheme of the electric drive reconfiguration type vehicle-mounted charging system, the output torque of the six-phase symmetrical permanent magnet synchronous motor can be adjusted by controlling the q-axis current. Therefore, using the inverse Park transformation matrix, the relationship between the phase A current and the d, q-axis currents I d 、I q is as follows:

[0039]

[0040] I A = I α + I x + I 01 + I 02 = I d cos(θ e ) - I q sin(θ e ) + I x + I 01 + I 02 (4)

[0041] Where θ e is the electrical angle of the motor rotor. In the ideal control case of a single neutral point, the currents I d 、I x 、I 01 of the subspace currents d, x, and 01 axes are 0, and in practical applications, affected by the control accuracy and the sensor acquisition accuracy, there is a ripple I A_ri1 . Therefore, the phase A current can be simplified as:

[0042] I A = -I q sin(θ e ) + I 02 + I A_ri1 (5)

[0043] However, in the DC charging mode, the charging current is adjusted through the 02-axis zero-sequence current I 02 to adjust the charging power of the battery. At this time, the motor inductor and the inverter are equivalent to a boost chopper circuit, as shown in Figure 3 . Due to the inherent characteristics of the boost circuit, the current I 02 has a charging current ripple, and this ripple can be calculated by the volt-second balance principle within one switching period:

[0044]

[0045] Wherein, V dc is the DC source voltage, and V b is the battery voltage; L X is the inductance of winding X ∈ {A, B, C, U, V, W}; D X is the duty cycle of the bridge arm corresponding to winding X, and T s is the switching period.

[0046] According to Equation (6), it can be known that the high-frequency charging current pulsation g X (t) of each phase of the motor within a switching period can be expressed as:

[0047]

[0048] Wherein, t represents time.

[0049] Therefore, the specific expression of the motor phase current is:

[0050]

[0051] Step 2: Establish a high-precision torque model through the method of co-energy and obtain the mathematical expression formula of the high-frequency torque component, as follows:

[0052] In the charging topology proposed in the present invention, the high-frequency charging current pulsation generated after the injection of the 02-axis zero-sequence current I 02 will inevitably cause high-frequency torque. From the perspective of electromechanical energy conversion, the electromagnetic torque T e of the motor can be calculated according to the co-energy formula, that is, the partial derivative of the magnetic field energy storage W m with respect to the mechanical angle θ e :

[0053]

[0054] Wherein, p n is the number of pole pairs of the motor, and ψ X is the magnetic flux linkage of winding X.

[0055] Under the condition of charging while running, due to the injection of the 02-axis zero-sequence current I 02 , the phase magnetic flux linkage generates a DC bias, and the 02-axis zero-sequence current I 02 is related to the third-harmonic magnetic flux linkage. Therefore, the magnetic flux linkage ψ X of each phase of the motor is composed of the fundamental magnetic flux linkage, the third-harmonic magnetic flux linkage and the DC component. The expression of the magnetic flux linkage of each phase under the charging condition is obtained by using the finite element scheme:

[0056]

[0057] where ψ dc is the DC component of the phase flux linkage; ψ f1 is the fundamental amplitude of the phase flux linkage; ψ f3 is the third harmonic amplitude of the phase flux linkage;

[0058] Based on Equation (9), the expression formula for the high-precision electromagnetic torque model of the motor is:

[0059] T e = 3p n [I q ψ f1 + I q I d (L d - L q )] + T0 (11)

[0060] where the high-frequency torque component T0 is expressed as:

[0061] T0 = 3p n K(t)ψ f3 sin3θ e

[0062] The high-precision torque model can be further expressed as:

[0063] T e = T em + T er + T0 = 3p n [I q ψ f1 + I q I d (L d - L q )] + 3p n K(t)ψ f3 sin3θ e (12)

[0064] where T em is the permanent magnet torque, T er is the reluctance torque, L d , L q are the d-axis and q-axis inductances of the motor respectively, K(t) is the sum of the components of each phase current under the charging condition, and K(t) = (g A (t) + g B (t) + g C (t)) + (g U (t) + g V (t) + g W (t)).

[0065] Step 3: Compensate the high-frequency torque through the reluctance torque of the motor, and then calculate the current value to be compensated on the d-axis. Let:

[0066]

[0067] The d-axis current to be compensated is obtained according to the formula:

[0068]

[0069] Step 4: Take the d-axis current compensation value I dr , the current reference value I d * and the actual value I d as the inputs of the d-axis current loop, and obtain the d-axis reference voltage through the regulation of the PI controller.

[0070] Step 5: After taking the difference between the actual values and the reference values of the q, x, y, 01-axis and 02-axis currents, respectively obtain the relevant reference voltages through the PI controller. After passing the reference voltages through the inverse Park transformation matrix and the inverse VSD matrix, input them into the PWM signal generation unit, and then obtain the duty cycle of each phase, so as to realize the normal operation of the system.

[0071] The present invention introduces a magnetic co-energy analysis and feed-forward compensation scheme to solve the problem of high-frequency torque ripple existing in the electric drive reconfiguration type on-vehicle charging system. To establish a high-precision torque model, suppress high-frequency torque ripple, quantitatively analyze the relationship between high-frequency torque ripple and current ripple, and then determine the compensation current value of the feed-forward compensation scheme. To verify the method of the present invention, based on Figure 1 the control block diagram of a method for suppressing high-frequency torque ripple of an electric drive reconfiguration type on-vehicle charging system shown, a simulation model of dual-motor synchronous control is built. In this embodiment, the high-frequency torque ripple of a six-phase permanent magnet synchronous motor in the running and charging state is simulated and verified.

[0072] In the simulation verification of the present invention, the photovoltaic input condition is set as the standard light intensity (1000W / m 2 ), and the typical working condition of DC charging (zero speed / zero load) is simulated. Figure 4 And Figure 5 shows the comparative simulation results of the high-frequency electromagnetic torque suppression strategy in the DC / photovoltaic charging mode. Figure 4 The simulation data shows that: the photovoltaic input current IVRPPs generates significant high-frequency ripple (about ±1.5A) due to the inherent switching frequency characteristics, exciting high-frequency torque ripple with an amplitude of up to ±0.5N·m. This ripple component will exacerbate mechanical vibration noise, indicating the inherent defect of the traditional modulation strategy in high-frequency harmonic suppression. Figure 5The simulation results show that the amplitude of the high-frequency ripple of the input current is attenuated to ±0.3 A, and the corresponding high-frequency torque ripple is suppressed to ±0.1 N·m. Based on the above simulation results, the high-frequency torque suppression scheme of the present invention can effectively suppress the high-frequency torque ripple during the charging process and avoid high-frequency vibration and noise in the system during the charging process.

[0073] The above are only the preferred embodiments of the present invention. It should be noted that for those of ordinary skill in the art, without departing from the principle of the present invention, several improvements and refinements can be made, and these improvements and refinements should also be regarded as the protection scope of the present invention.

Claims

1. A method for suppressing high-frequency torque pulsation in an electric drive reconfigurable vehicle charging system, characterized in that: include: Step 1: Establish a mathematical expression formula of the phase current of the system in charging mode according to the vector space decoupling matrix; Step 2: Establish a high-precision torque model of the motor through the magnetic energy method, and obtain the mathematical expression formula of the high-frequency torque component; Step 3: Compensate the high-frequency torque by the reluctance torque of the motor, and then calculate the current component required to compensate for the d-axis; Step 4: The d-axis current compensation value, the current reference value and the actual value are used as the input of the current feedback loop, and the d-axis reference voltage is obtained by adjusting the PI controller; Step 5: Realize high-frequency torque pulsation suppression of the system in charging mode through vector control and feedforward control schemes.

2. The high-frequency torque pulsation suppression method of the electric drive reconfigurable vehicle charging system according to claim 1 is characterized in that: The electric drive reconfigurable on-board charging system adopts a six-phase symmetrical permanent magnet synchronous motor. In the vector control scheme of the system, the output torque of the motor is adjusted by controlling the q-axis current, and the charging power of the battery is adjusted by controlling the 02-axis zero-sequence current. The 02-axis zero-sequence current I 02 There is charging current pulsation. Combined with the vector space decoupling matrix, the mathematical expression of the phase current of the system in charging mode is established as follows: In the formula, I A ,I B ,I C ,I U ,I V ,I W are the phase currents of the six-phase symmetrical permanent magnet synchronous motor, I q ,I 02 are the q-axis current component and the 02-axis zero-sequence current component, θ e is the electrical angle of the motor rotor, g X (t) is the high-frequency charging current pulsation of each phase of the motor in one switching cycle, X∈{A,B,C,U,V,W}, t represents time; g X (t) is specifically expressed as: Where V dc is the DC source voltage, V b is the battery voltage, L X is the inductance of winding X, D X is the duty cycle of the bridge arm corresponding to winding X, T s is the switching cycle.

3. The high-frequency torque pulsation suppression method of the electric drive reconfigurable vehicle charging system according to claim 2 is characterized in that: In step 2, under the condition of running and charging, due to the zero-sequence current I 02 The injection of the phase flux leads to a DC bias, and the zero-sequence current I 02 Related to the third harmonic flux, the flux per phase of the motor is X It is composed of fundamental flux, third harmonic flux and DC component. The expression of each phase flux under charging condition is obtained by finite element method. Then the high-precision torque mathematical model of the motor is established by magnetic energy scheme: T e =3p n [I q ψ f1 +I q I d (L d -L q )]+T0 (3) Where, T e is the electromagnetic torque of the motor, p n is the number of motor pole pairs, I d is the d-axis current component, L d With L q are the d-axis and q-axis inductances of the motor, ψ f1 is the fundamental amplitude of the phase flux, T0 is the zero-sequence current I 02 The high-frequency torque caused by the high-frequency charging current pulsation generated after injection: T0=3p n K(t)ψ f3 sin3θ e (4) In the formula, ψ f3 is the third harmonic amplitude of the phase flux, and K(t) is the sum of the components of each phase current under charging conditions.

4. The high-frequency torque pulsation suppression method of the electric drive reconfigurable vehicle charging system according to claim 3 is characterized in that: In step 3, the current component required to be compensated for the d-axis is calculated based on the high-frequency torque T0, that is, the reluctance torque T generated by the d-axis current is calculated. er To suppress the high-frequency torque ripple caused by the zero-sequence current, the compensation value of the injected d-axis current is expressed as:

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