Controller and control method for smooth mode switching of four-switch buck-boost converter

Abstract

The invention discloses a controller and a control method for smooth mode switching of a four-switch buck-boost converter, and the controller comprises a voltage sampler and a current sampler which are respectively used for obtaining the voltage at the two ends of a second capacitor C2 and the current of an inductor L1; the voltage sampler and the current sampler are respectively connected with the second loop compensator PID2 and the first loop compensator PID1, and then are in cascade connection and output connection with the nonlinear controller; an output signal of the nonlinear controller controls the first switch tube S1, the second switch tube S2, the third switch tube S3 and the fourth switch tube S4 through a driving amplification module; according to the control method, the controller is adopted to control the four-switch buck-boost converter, so that smooth mode switching of the four-switch buck-boost converter is realized. Inductive current and output voltage fluctuation in the mode switching process can be restrained, control boundary circulation jitter is restrained, smoothness of the mode switching process is achieved, and the stability and the response speed of a system are improved.

Description

Technical Field

[0001] The present invention relates to a controller and a control method for a four-switch buck-boost converter, and particularly to a controller and a control method for smooth mode switching of a four-switch buck-boost converter. Background Art

[0002] The information provided in this part is only background information related to the present disclosure, and it is not necessarily prior art.

[0003] As an efficient DC-DC converter, the four-switch Buck-Boost (FSBB) converter is widely used in various electronic devices. Its core advantage lies in being able to achieve an output voltage higher than or lower than the input voltage, having good voltage regulation performance and high conversion efficiency. This technology is applicable to fields such as bidirectional power conversion, energy storage, electric vehicles, solar photovoltaic power generation systems, communication power supplies, and LED lighting.

[0004] The four-switch Buck-Boost converter has three operating modes: Buck, Boost, and Buck-Boost modes. When the input voltage changes, it is necessary to switch among the three modes. The existing control methods have the following problems:

[0005] 1. The existing switching control strategies are relatively complex. Currently, most switching schemes design the boundaries of mode switching in the form of a hysteresis loop, and there are obvious dynamic steps and fluctuations at the switching boundaries; non-linear and advanced control methods such as sliding mode control and model predictive control pose high challenges to the speed and accuracy of the controller when the load switches quickly or the load changes rapidly.

[0006] 2. The small-signal models in the three modes are slightly different, which will cause instability problems at the switching boundaries, resulting in large fluctuations in voltage and inductor current.

[0007] Current research has conducted many studies on the control and mode switching problems of the FSBB converter. In the application of the actual switching control method of the FSBB, the maximum and minimum duty cycle constraints and the dead time of the switching tubes will affect the stability of the converter during the three-mode switching, causing fluctuations in the output voltage and inductor current, and thus affecting the dynamic performance and stability of the converter.

[0008] It should be noted that the information disclosed in the above background art section is only used to enhance the understanding of the background of the present disclosure, and thus may include information that does not constitute prior art known to those of ordinary skill in the art. Summary of the Invention

[0009] Objective of the Invention: The technical problem to be solved by the present invention is to provide a controller and a control method for smooth mode switching of a four-switch buck-boost converter in view of the deficiencies of the prior art.

[0010] To solve the above technical problem, the present invention discloses a controller and a control method for smooth mode switching of a four-switch buck-boost converter. The four-switch buck-boost converter includes a first switch tube, a second switch tube, a third switch tube, a fourth switch tube, a first capacitor, a second capacitor, and an inductor. Among them, the source electrode of the first switch tube and the drain electrode of the second switch tube are connected in series to form a first half-bridge, the source electrode of the third switch tube and the drain electrode of the fourth switch tube are connected in series to form a second half-bridge, both ends of the inductor are connected between the source electrode of the first switch tube and the source electrode of the third switch tube, the first capacitor is connected between the drain electrode of the first switch tube and the source electrode of the second switch tube, the second capacitor is connected between the drain electrode of the third switch tube and the source electrode of the fourth switch tube S4, and the source electrode of the second switch tube is connected to the source electrode of the fourth switch tube; the voltage across the first capacitor is, the voltage across the second capacitor is, and the controller includes:

[0011] A voltage sampler and a current sampler, which are respectively used to obtain the voltage across the second capacitor and the current of the inductor;

[0012] After the voltage sampler and the current sampler are respectively connected to a second loop compensator and a first loop compensator, they are cascaded and the output is connected to a non-linear controller;

[0013] The output signal of the non-linear controller controls the first switch tube, the second switch tube, the third switch tube, and the fourth switch tube through a drive amplification module.

[0014] Further, the controller specifically includes:

[0015] The second loop compensator PID2, the input of which is the error signal ve2 between the vf2 signal output by the voltage sampler and the voltage reference signal vr2, that is, ve2 = vr2 - vf2, and the output is the vr1 signal;

[0016] The first loop compensator PID1, the input of which is the error signal ve1 between the vf1 signal output by the current sampler and the vr1 signal, that is, ve1 = vr1 - vf1, and the output is the vpid1 signal;

[0017] The vpid1 signal is input to the non-linear controller to generate four PWM signals, namely the PWM_D1 signal, the PWM_D2 signal, the PWM_D3 signal, and the PWM_D4 signal, for controlling the first switch tube, the second switch tube, the third switch tube, and the fourth switch tube;

[0018] The four-channel PWM signals are respectively connected to the gates of the first switch tube, the second switch tube, the third switch tube, and the fourth switch tube after passing through the drive amplification module to complete the control.

[0019] The present invention also proposes a control method for smooth mode switching of a four-switch buck-boost converter, including the following steps:

[0020] Step 1: Calculate the vpid1 signal according to the output voltage signal of the four-switch buck-boost converter, the inductor current signal in the four-switch buck-boost converter, and the reference voltage signal.

[0021] Step 2: Use a non-linear controller to calculate the control signals of the four switch tubes in the four-switch buck-boost converter according to the vpid1 signal.

[0022] Step 3: Use the control signals obtained in Step 2 to control the four switch tubes to achieve smooth mode switching of the four-switch buck-boost converter.

[0023] Further, the calculation of the vpid1 signal in Step 1 includes:

[0024] Step 1-1: Use the second loop compensator PID2 to convert the error signal ve2 between the vf2 signal output by the voltage sampler and the voltage reference vr2 signal, that is, ve2 = vr2 - vf2, into a vr1 signal for output.

[0025] Step 1-2: Use the first loop compensator PID1 to convert the error signal ve1 between the vf1 signal output by the current sampler and the vr1 signal, that is, ve1 = vr1 - vf1, into a vpid1 signal for output.

[0026] Further, the use of the control signals obtained in Step 2 to control the four switch tubes in Step 3 includes:

[0027] Use a drive amplification module to amplify the control signals and then use them to control the gates of the four switch tubes in the four-switch buck-boost converter respectively.

[0028] Further, the calculation of the control signals of the four switch tubes in the four-switch buck-boost converter according to the vpid1 signal in Step 2 includes:

[0029] Step A1: Judge the vpid1 signal input to the non-linear controller, specifically including:

[0030] When vpid1 ≤ Dmax, execute Step 2, where Dmax is the maximum duty ratio value of the duty ratio D1 of the first switch tube and the duty ratio D3 of the third switch tube.

[0031] When vpid1 > Dmax and vpid1 < 1 + (1 - Dmax), perform step A3;

[0032] When vpid1 > 1 + (1 + Dmax), perform step A4;

[0033] Step A2, set:

[0034] D1 = vpid1

[0035] D3 = 1

[0036] Perform step A7;

[0037] Step A3, calculate the temporary values D11 and D31 of the duty cycles of the first switch tube and the third switch tube; make a judgment based on the temporary values D11 and D31, specifically including:

[0038] When D11 ≥ D31, perform step A5;

[0039] When D11 < D31, perform step A6;

[0040] Step A4, set:

[0041] D1 = 1

[0042] D3 = vpid1 / (2·vpid1max - vpid1)

[0043] Perform step A7;

[0044] Step A5, set:

[0045] D1 = Dmax

[0046] D3 = vpid1 / (2·vpid1max - vpid1)

[0047] Perform step A7;

[0048] Step A6, set:

[0049] D3 = Dmax

[0050] D1 = (2·vpid1max - vpid1) / vpid1

[0051] Perform step A7;

[0052] Step A7, use the values of D1 and D3 as modulation wave values respectively, compare with the triangular carrier wave Tsw1 to obtain the PWM signal PWM_D1 of the first switch tube and the PWM signal PWM_D3 of the third switch tube, and obtain the PWM signal PWM_D2 of the second switch tube and the PWM signal PWM_D4 of the fourth switch tube according to their inverted signals respectively.

[0053] Further, the specific method for calculating the temporary values D11 and D31 of the duty cycles of the first switching transistor and the third switching transistor described in step A3 includes:

[0054] D11 = vpid1

[0055] D31 = 2·vpid1max - vpid1

[0056] where vpid1max is the maximum output value of the first loop compensator.

[0057] Further, calculating the control signals of the four switching transistors in the four-switch buck-boost converter according to the vpid1 signal described in step 2 includes:

[0058] Step B1, calculating the dead-time equivalent loss duty cycle Dd according to the dead time Td of the driving and amplifying module, specifically as follows:

[0059]

[0060] where Ts is the switching cycle time;

[0061] Step B2, judging the vpid1 signal input to the non-linear controller, specifically including:

[0062] When vpid1 ≤ Dmax, execute step B3; where Dmax is the maximum duty cycle value of the duty cycle D1 of the first switching transistor and the duty cycle D3 of the third switching transistor;

[0063] When vpid1 > Dmax and vpid1 < 1 + (1 - Dmax), execute step B4;

[0064] When vpid1 > 1 + (1 + Dmax), execute step B5;

[0065] Step B3, set:

[0066] D1 = vpid1

[0067] D3 = 1

[0068] Execute step B8;

[0069] Step B4, calculating the temporary values D11 and D31 of the duty cycles of the first switching transistor and the third switching transistor; judging according to the temporary values D11 and D31, specifically including:

[0070] When D11 ≥ D31, execute step B6;

[0071] When D11 < D31, execute step B7;

[0072] Step B5, set:

[0073] D1 = 1

[0074] D3 = vpid1 / (2·vpid1max - vpid1)

[0075] Execute step B8;

[0076] Step B6; Set

[0077] D1 = Dmax

[0078]

[0079] Execute step B8;

[0080] Step B7, Set:

[0081] D3 = Dmax

[0082] D1 = (2·vpid1max - vpid1) / vpid1 + Dd

[0083] Execute step B8;

[0084] Step B8: Take the values of D1 and D3 as the modulation wave values respectively, compare them with the triangular carrier wave Tsw1 to obtain the PWM signal PWM_D1 of the first switching tube and the PWM signal PWM_D3 of the third switching tube, and take their reverse signals as the PWM signal PWM_D2 of the second switching tube and the PWM signal PWM_D4 of the fourth switching tube respectively.

[0085] Furthermore, the specific method for calculating the temporary values D11 and D31 of the duty cycles of the first switching tube and the third switching tube in step B4 includes:

[0086] D11 = vpid1 - Dd

[0087] D31 = 2·vpid1max - vpid1 - Dd

[0088] Wherein, vpid1max is the output maximum value of the first loop compensator.

[0089] Beneficial effects:

[0090] 1) The non - linear controller designed according to the present invention can achieve smooth output voltage and smooth inductor current during the transient process of Buck, Buck - Boost - Boost three - mode switching, and solve the instability problem caused by control boundary cyclic jitter.

[0091] 2) The present invention is applicable to different voltage and current control scenarios, suitable for use in two-way and one-way conversion scenarios of four-switch Buck-Boost, adaptable to constant current and constant voltage control, step-up and step-down scenarios, and suitable for typical application scenarios such as battery charging and discharging in energy storage scenarios.

[0092] 3) The theoretical analysis and simulation results in this paper both show that this controller has beneficial improvement effects and has good dynamic performance in the smooth switching of three modes. Brief Description of the Drawings

[0093] The following further specifically describes the present invention in conjunction with the drawings and specific embodiments, and the above and / or other advantages of the present invention will become clearer.

[0094] Figure 1 It is a schematic diagram of the overall structure of the power part and control part of the FSBB converter in the example.

[0095] Figure 2 It is a schematic diagram of the control method of the non-linear controller in Embodiment 1.

[0096] Figure 3 It is a schematic diagram of the control method of the non-linear controller in Embodiment 2.

[0097] Figure 4 It is a schematic diagram of the input signal vpid1 of the non-linear controller in Embodiment 1 to the duty ratios D1 and D3.

[0098] Figure 5 It is a schematic diagram of the input signal vpid1 of the non-linear controller in Embodiment 2 to the duty ratios D1 and D3.

[0099] Figure 6 It is a schematic diagram of the enlarged part of the intermediate waveform of the input signal vpid1 of the non-linear controller in Embodiment 2 to the duty ratios D1 and D3.

[0100] Figure 7 It is a schematic diagram of the time-domain dynamic waveform during the mode switching process without using this method.

[0101] Figure 8 It is a schematic diagram of the time-domain dynamic waveform during the mode switching process when using the non-linear controller in Embodiment 1.

[0102] Figure 9 It is a schematic diagram of the time-domain dynamic waveform during the mode switching process when using the non-linear controller in Embodiment 2.

[0103] Figure 10 It is a schematic diagram of the time-domain dynamic waveform during the mode switching process of the converter caused by the input voltage step in the time-domain simulation when using the non-linear controller in the embodiment. Detailed Description of the Embodiment

[0104] The general idea of the present invention is as follows: Based on a four-switch buck-boost converter (four-switch Buck-Boost converter), an improved control strategy for smooth three-mode switching is proposed to reduce the influence of the maximum and minimum duty cycle constraints and the switch drive dead time on the switching dynamic process during mode switching, and to reduce the output voltage fluctuation, current fluctuation and boundary cycle jitter problems during the switching process. This control method has the following advantages: significantly reducing the voltage and current fluctuations brought about during the three-mode switching, improving the stability of the controller; not requiring additional hardware circuits and having no cost increase; being easy to implement in a digital controller with a very low computational load and not significantly increasing the real-time loop calculation burden; achieving smooth three-mode switching and improving the dynamic response of the converter.

[0105] The present invention is based on a non-linear transformation method of carrier modulation with average current mode control. The idea of the specific technical solution is as follows:

[0106] The four-switch Buck-Boost converter described in the present invention, namely the FSBB converter, includes a first switching tube S1, a second switching tube S2, a third switching tube S3, a fourth switching tube S4, a first capacitor C1, a second capacitor C2, and a first inductor L1. The source of the first switching tube S1 and the drain of the second switching tube S2 are connected in series to form a first half-bridge, and the source of the third switching tube S3 and the drain of the fourth switching tube S4 are connected in series to form a second half-bridge. Both ends of the inductor L1 are connected between the source of the first switching tube S1 and the source of the third switching tube S3. The first capacitor C1 is connected between the drain of the first switching tube S1 and the source of the second switching tube S2, and the second capacitor C2 is connected between the drain of the third switching tube S3 and the source of the fourth switching tube S4. The source of the second switching tube S2 is connected to the source of the fourth switching tube S4. The voltage of the first capacitor C1 is V1, the voltage of the second capacitor C2 is V2, the duty cycle of the first switching tube S1 is D1, and the duty cycle of the third switching tube S3 is D3.

[0107] The controller of the four-switch Buck-Boost converter described in the present invention includes a first loop compensator PID1, a current sampler, a voltage sampler, a second loop compensator PID2, a non-linear controller, and a drive amplifier module; among them, the second loop compensator PID2 serves as the control outer loop for controlling the output voltage V2 to follow a given reference, and its output serves as the reference for the first loop compensator; the first loop compensator PID1 serves as the inner loop for controlling the inductor current to follow the output of the outer loop PID2; the non-linear controller is used to generate drive signals for the four switching tubes with different duty cycles according to the output of the inner loop PID1; the above devices in the controller finally output a duty cycle pulse width modulation signal (PWM) for the four switches through sampling, loop compensation, and non-linear transformation, so as to control the voltage V2 to follow the reference and control the FSBB to smoothly switch between three modes. The output ends of the drive amplifier module are respectively connected to the gates of the four switching tubes, and after amplifying the output control signal of the non-linear controller, they are used to control the 4 switching tubes; specifically as follows:

[0108] The voltage sampler samples the voltage V2 and outputs the vf2 signal, and the current sampler samples the inductor current and outputs the vf1 signal; the second loop compensator samples the V2 voltage, performs error comparison and tracking compensation with the voltage reference value vr2, generates the reference vr1 signal of the inner loop current compensator, that is, the first loop compensator. The inner loop current compensator performs error comparison and tracking compensation on the vf1 signal and the reference vr1, and outputs the vpid1 signal. The vpid1 signal undergoes a duty cycle transformation through the non-linear transformation controller to output a PWM signal PWM_D1 with a duty cycle of D1 and a PWM signal PWM_D3 with a duty cycle of D3.

[0109] The duty cycle D2 of the second switching tube S2 is complementary to the duty cycle D1 of the first switching tube S1, and they always satisfy D1 + D2 = 1. The PWM signal of the second switching tube S2 is PWM_D2. The duty cycle D3 of the third switching tube S3 is complementary to the duty cycle D4 of the fourth switching tube S4, and they always satisfy D3 + D4 = 1. The PWM signal of the fourth switching tube S4 is PWM_D4.

[0110] The maximum duty cycle of the first switching transistor S1 is D1, and the maximum duty cycle of the third switching transistor S3 is D3, with the minimum duty cycle being Dmin. The minimum value of the triangular carrier wave Tsw1 used by the first switching transistor S1 and the second switching transistor S2 to generate PWM is 0, the maximum value is 1, and the frequency is fsw; the minimum value of the triangular carrier wave Tsw2 used by the third switching transistor S3 and the fourth switching transistor S4 to generate PWM is 0, the maximum value is 1, and the frequency is fsw; there is no requirement for the phase difference between the Tsw1 and Tsw2 signals. The minimum output of the first loop compensator is 0, and the maximum value is vpid1max; the value range of vpid1max is (1, 2.0], and the value of vpid1max determines the working range of the Buck - Boost mode: when the output value of vpid1 is within the interval [0, vpid1max - 1), the duty cycle D1 of the first switching transistor S1 ≤ 1, the duty cycle D3 of the third switching transistor S3 = 1, and the converter operates in the Buck mode; when the output value of vpid1 is within the interval [vpid1max - 1, vpid1max), the duty cycle D1 of the first switching transistor S1 < 1, the duty cycle D3 of the third switching transistor S3 < 1, and the converter operates in the Buck - Boost mode; when the output value of vpid1 is within the interval [1+(1 - vpid1max), vpid1max], the duty cycle D1 of the first switching transistor S1 = 1, the duty cycle D3 of the third switching transistor S3 ≤ 1, and the FSBB converter operates in the Boost mode.

[0111] When the duty cycle D1 of the first switching transistor S1 < 1 and the duty cycle D3 of the third switching transistor S3 = 1, the FSBB converter is in the Buck mode; when the duty cycle D1 of the first switching transistor S1 = 1 and the duty cycle D3 of the third switching transistor S3 < 1, the FSBB converter is in the Boost mode; when the duty cycle D1 of the first switching transistor S1 < 1 and the duty cycle D3 of the third switching transistor S3 < 1, the FSBB converter is in the Buck - Boost mode.

[0112] When the non - linear controller generates a PWM signal PWM_D1 with a duty cycle of D1 and a PWM signal PWM_D3 with a duty cycle of D3 through the input signal vpid1, the inductor current loop compensation output vpid1 is subjected to the following non - linear transformation:

[0113] 1) When vpid1 ≤ Dmax, let D1 = vpid1 and let D3 = 1.

[0114] 2) When vpid1 > Dmax and vpid1 < 1+(1 - Dmax), first calculate according to the formulas D11 = vpid1 and D31 = 2·vpid1max - vpid1 to calculate the temporary values D11 and D31 of the duty cycles of the first switching transistor S1 and the third switching transistor S3; compare the maximum values of D11 and D31:

[0115] When D11 ≥ D31, let D1 = Dmax, and let D3 = vpid1 / (2·vpid1max - vpid1);

[0116] When D11 < D31, let D3 = Dmax, and let D1 = (2·vpid1max - vpid1) / vpid1.

[0117] 3) When vpid1 > 1 + (1 + Dmax), let D1 = 1, and let D3 = vpid1 / (2·vpid1max - vpid1).

[0118] 4) Use the value corresponding to D1 as the modulation wave value, compare it with the triangular carrier wave Tsw1 to obtain the PWM signal PWM_D1 of the first switching tube S1, and its inverted signal is the PWM signal PWM_D2 of the second switching tube S2; use the value corresponding to D3 as the modulation wave value, compare it with the triangular carrier wave Tsw2 to obtain the PWM signal PWM_D3 of the third switching tube S3, and its inverted signal is the PWM signal PWM_D4 of the second switching tube S2.

[0119] By implementing the control process and control method of the non - linear controller, with the first loop compensator as the input vpid1, the PWM signals PWM_D1 of the first switching tube S1, PWM_D2 of the second switching tube S2, PWM_D3 of the third switching tube S3, and PWM_D4 of the fourth switching tube S4 are used to drive the switching tubes, and the smooth switching process of the three - mode can be realized.

[0120] In a further technical solution, considering that the dead - time Td is added after the D1 and D3 signals pass through the drive circuit, the equivalent duty - cycle loss of the dead - time Td is Dd = Td / Ts, the actual conduction duty - cycle of the first switching tube S1 is D1d = D1 - Dd, and the dead - time signal of the second switching tube S2 is D3d = D3 - Dd. When the non - linear controller generates the duty - cycles D1 and D3 through the input signal vpid1, the following non - linear transformation is performed on the inductor current loop compensation output vpid1:

[0121] 1) Calculate the dead - time equivalent loss duty - cycle Dd = Td / Ts according to the dead - time Td of the drive circuit.

[0122] 2) When vpid1 ≤ Dmax, let D1 = vpid1, and let D3 = 1.

[0123] 3) When vpid1 > Dmax and vpid1 < 1 + (1 - Dmax), first calculate the temporary values D11 and D31 of the duty cycles of the first switch S1 and the third switch S3, and calculate them according to the formulas D11 = vpid1 - Dd and D31 = 2·vpid1max - vpid1 - Dd; compare the maximum values of the two:

[0124] When D11 ≥ D31, let D1 = Dmax, and let D3 = vpid1 / (2·vpid1max - vpid1) + Dd;

[0125] When D11 < D31, let D3 = Dmax, and let D1 = (2·vpid1max - vpid1) / vpid1 + Dd;

[0126] 4) When vpid1 > 1 + (1 + Dmax), let D1 = 1, and let D3 = vpid1 / (2·vpid1max - vpid1).

[0127] 5) Take the value corresponding to D1 as the modulation wave value, compare it with the triangular carrier wave Tsw1 to obtain the PWM signal PWM_D1 of the first switch S1, and its inverted signal is the PWM signal PWM_D2 of the second switch S2; take the value corresponding to D3 as the modulation wave value, compare it with the triangular carrier wave Tsw2 to obtain the PWM signal PWM_D3 of the third switch S3, and its inverted signal is the PWM signal PWM_D4 of the second switch S2.

[0128] By implementing the control process and control method of the non - linear controller, with the first loop compensator as the input vpid1, the PWM signals PWM_D1 of the first switch S1, PWM_D2 of the second switch S2, PWM_D3 of the third switch S3, and PWM_D4 of the fourth switch S4 are generated to drive the switch tubes, compensating for the influence of the dead - time on the three - mode switching process, and the smoothness of the three - mode switching process can be achieved.

[0129] Embodiment 1:

[0130] In a specific embodiment, a four - switch Buck - Boost converter, as Figure 1 shown, includes: The first loop compensator is for inner - loop current regulation, used to regulate the inductor current to follow the output of the second loop compensator, that is, the outer - loop regulator. The second loop compensator is for outer - loop regulation, used to control the modulation to make the target voltage stable. According to the circuit working direction, it can be the V2 or V1 voltage. In this example, the V2 voltage is used for analysis. The duty cycles D1 and D3 of the four switch tubes of the FSBB are generated by the inner - loop current regulator vpid1.

[0131] In the research literature "Multimode control for a four-switch buck-boost converter" by Bosheng Sun, it is pointed out that during the switching transient process of the three modes, when maintaining the volt-second balance of the original inductor voltage, the inductor current will not mutate within the switching cycle of mode switching, thereby reducing the stability of the output voltage and maintaining the smoothness of the inductor current. Therefore, based on the inductor volt-second balance, the following conclusions can be easily deduced:

[0132] Inference (1): When the converter is in a steady state or within any single switching cycle, it is considered that V1 and V2 are stable and unchanged. Regardless of whether it is operating in any mode of Buck, Boost, or Buck-Boost, the relationship between the capacitor voltages V1, V2 and the duty cycles D1, D3 satisfies: V1·D1 = V2·D3, that is, V1 / V2 = D3 / D1.

[0133] Inference (2): If the volt-second balance of the inductor current is maintained, the inductor volt-second relationship before and after the step should be the same. The duty cycles of the first switch tube and the third switch tube before and after the step are D1p, D3p and D1s, D3s respectively; the duty cycles maintaining the inductor volt-second balance should have the following relationship: V1·D1p - V2·D3p = V1·D1s - V2·D3s. Initially, V1 / V2 = D3p / D1p. Divide both sides of the equation by V2 to eliminate V1 / V2, and we can get: D3s / D1s = D3p / D1p. That is: Before and after the duty cycle step, to maintain the smoothness of the inductor volt-second product, the duty cycle D1 of the first switch tube and the duty cycle D3 of the third switch tube should be in a geometric ratio relationship.

[0134] Taking the switching from Buck mode to Buck - Boost mode in this embodiment as an example, the mode - switching process is analyzed. The first switching transistor and the second switching transistor are used to generate the triangular carrier wave Tsw1 for PWM. The minimum value of Tsw1 is 0, the maximum value is 1, and the frequency is fsw. The third switching transistor and the fourth switching transistor are used to generate the triangular carrier wave Tsw2 for PWM. The minimum value of Tsw2 is 0, the maximum value is 1, and the frequency is fsw. Considering the switching process of the converter mode from Buck to Buck - Boost mode, before switching, the output of the current compensator vpid1 is less than 1. Let the duty ratio of the third switching transistor be D3 = vpidmax - vpid1>1. At this time, the third switching transistor is in the always - on mode, and the duty ratio D3 is clamped to 1.0, operating in Buck mode. Suppose the voltage V2 slowly approaches V1, and the converter is about to switch to Buck - Boost mode. At this time, the output vpid1 of the current compensator increases. When D3 = vpidmax - vpid1<1 and D3>Dmax, the controller determines that it is necessary to switch from Buck to Buck - Boost mode, that is, the third switching transistor operates with a duty ratio less than 1. Since it is necessary to impose a maximum duty - ratio constraint on the duty ratio D3, the duty ratio D3 will step from 1.0 to Dmax (such as 0.9). In a switching period Ts, the voltage - second product at the connection point of the inductor and the third switching transistor is:

[0135] Before the step: Ts·V D3阶跃前 = Ts·V2·D3 阶跃前 = Ts·V2·1.0 = Ts·V2,

[0136] After the step: Ts·V D3阶跃后 = Ts·V2·D3 阶跃后 = Ts·V2·Dmax,

[0137] The voltage - second product at the connection point of the inductor and the first switching transistor has no step change, which is Ts·V D1 = Ts·V1·D1 = Ts·V1·vpid1. Therefore, the step amount of the voltage - second on the inductor is:

[0138] Ts·V D3阶跃后 - Ts·V D3阶跃前 = Ts·V2·(1 - Dmax)

[0139] The inductor volt-second integral is theoretically zero at steady state. As a result of a small voltage step occurring across the inductor, the inductor current will increase and change in the biasing direction within one period. According to the sign of the volt-second integral of the step, it can be known that Ts·V2·(1 - Dmax) > 0, and the inductor current will increase in the associated direction in the current period and several future periods. According to the negative feedback logic of the first compensator, when the inductor current increases in the associated direction, vf1 increases, and the negative error input vr1 - vifed of the first compensator increases negatively, that is, the error becomes larger; the P link of the first compensator will quickly make adjustments, resulting in a negative change in vpid1, that is, a decrease, which is reciprocal to the logic that the increase in the output of vpid1 causes the converter to switch from the Buck mode to the Buck-Boost mode as described above. The error compensation effect of the first compensator attempts to pull down vpid1, which may cross the boundary between the Buck and Buck-Boost modes, and pull the converter back to the Buck mode in the next period; this process may repeat cyclically. Therefore, the duty cycle step caused by the direct switching between the above two modes may cause control boundary cyclic jitter at this boundary, resulting in inductor current fluctuations and uneven output voltage. If the compensator parameters are not designed properly, it may cause the controller to become unstable.

[0140] As vpid1 increases, when the duty cycle D3 of the third switching transistor is D3 = vpid1max - vpid1 < Dmax, the controller crosses the maximum duty cycle constraint interval (Dmax, 1.0) of D3 for the Buck and Buck-Boost modes, and the duty cycle of D3 enters the linear regulation range of the first compensator.

[0141] When the control method of the non-linear controller of the present invention is adopted, as Figure 2 shown, the detailed process is as follows.

[0142] The voltage of the first inductor's volt-second before the step is: VL 阶跃前 = V1·D1 阶跃前 - V2·D3 阶跃前 = V1·vpid1 - V2·vpid1max - vpid1

[0143] The core idea of the control method described in the present invention is to keep the inductor volt-seconds consistent before and after the mode switching transient. Considering the same switching process as above, when a mode switch occurs, the duty cycle D3 of the third switching transistor undergoes a step. At the same time, since D3 is clamped at Dmax within the maximum duty cycle constraint interval (Dmax, 1.0), in order to keep the inductor volt-seconds consistent, a correction should be applied to the duty cycle of the first switching transistor. According to the conclusion in "Inference (1)", the duty cycle D1 of the first switching transistor and the duty cycle D3 of the third switching transistor that maintain the balance of the inductor volt-seconds before and after are in an equal proportion relationship with the voltages on both sides. The duty cycle D1 of the first switching transistor after transformation 阶跃后 and the duty cycle D3 of the first switching transistor after transformation阶跃后 The following equation holds:

[0144] Equation 1: V1 / V2 = D3 阶跃前 / D1 阶跃前 = D3 阶跃后 / D1 阶跃后

[0145] In "Equation 1", the duty cycle D3 before the step 阶跃前 = 1, D1 阶跃前 = vpid1; after the step, the duty cycle D3 is clamped, D3 阶跃后 = Dmax, so the duty cycle of D1 can be deduced:

[0146] Equation 2:

[0147] According to "Equation 1", for a fixed voltage operating point, there must be multiple solutions for the duty cycle, that is, the duty cycle D1 of solution 1 解1 , D3 解1 and the duty cycle D1 of solution 2 解2 , D3 解2 As long as the identity V1 / V2 = D3 解1 / D1 解1 = D3 解2 / D1 解2 is satisfied, that is, the inductor volt-second balance relationship is satisfied.

[0148] On the other hand, the duty cycle D1 deduced according to "Equation 2" 阶跃后 can satisfy the inductor volt-second balance before and after the step, but when the value of D1 阶跃后 is greater than Dmax, D1 阶跃后 must also be clamped by the maximum duty cycle Dmax. At this time, the calculation process of Equation 1 needs to be iterated repeatedly until both duty cycles satisfy D1 阶跃后 ≤ Dmax and D3 阶跃后 ≤ Dmax. To simplify the calculation process and reduce the number of iterations, according to the conclusion of "Equation 1", it is easy to obtain that for the general duty cycles D3 解n , D1 解n satisfy the inverse ratio relationship D3 解n / D1 解n . Just select the larger value of D1 解1 , D3 解1 for Dmax clamping during the first iteration, and calculate the volt-second relationship for the smaller value of the two, then the final solution result D3 解n ≤ Dmax and D1 解n ≤ Dmax. Let the duty cycles after the maximum value comparison and calculation be D1 变换后1 , D3 变换后1 , and the simplified calculation process is as follows:

[0149] When the non-linear controller generates duty cycles D1 and D3 through the input signal vpid1, the inductor current loop compensation output vpid1 is subjected to the following non-linear transformation:

[0150] 1) When vpid1 ≤ Dmax, let D1 变换后1 = Vpid2, and let D3 变换后1 = 1. At this time, it operates in the Buck mode.

[0151] 2) When vpid1 > Dmax and vpid1 < 1+(1 - Dmax), first calculate the temporary values D11 and D31 of the duty cycles of the first switch and the third switch according to the formulas D11 = Vpid1 and D31 = 2·vpid1max - vpid1; compare D11 and D31 for the maximum value:

[0152] When D11 ≥ D31, let D1 变换后1 = Dmax, and let D3 变换后1 = vpid1 / (2·vpid1max - vpid1);

[0153] When D11 < D31, let D3 变换后1 = Dmax, and let D1 变换后1 = (2·vpid1max - vpid1) / vpid1. At this time, the converter enters the Buck - Boost mode.

[0154] 3) When vpid1 > 1+(1 + Dmax), let D1 变换后1 = 1, and let D3 变换后1 = vpid1 / (2·vpid1max - vpid1).

[0155] After completing the above duty cycle calculation, D1 变换后1 and D3 变换后1 , as the modulation wave values, are compared with the triangular carrier waves Tsw1 and Tsw2 to output the pulse width modulation signal PWM_D1 of the first switch with a duty cycle of D1 变换后1 , and output the pulse width modulation signal PWM_D3 of the third switch with a duty cycle of D3 变换后1 ; the complementary signal of the pulse width modulation signal PWM_D1 of the first switch is used as the pulse width modulation signal PWM_D2 of the second switch; the complementary signal of the pulse width modulation signal PWM_D3 of the third switch is used as the pulse width modulation signal PWM_D4 of the fourth switch. PWM_D1, PWM_D2, PWM_D3, and PWM_D4 are amplified by the drive circuit to drive the switch tubes to operate.

[0156] Generally, when the converter operates in the steady state of Buck mode, Buck-Boost mode, and Boost mode, the duty cycles D1 and D3 are equivalently converted according to the non-linear controller, which will not affect the operation of the converter in the steady state; at the boundaries of Buck and Buck-Boost modes and at the boundaries of Buck-Boost and Boost modes, the duty cycle transformation based on the working principle of the non-linear controller maintains the volt-second balance before and after the transformation, thereby ensuring the smoothness of the switching process.

[0157] Embodiment 2:

[0158] Next, consider the influence of the driving dead time on the switching process and the working principle of the non-linear controller, as Figure 3 shown. Let the switching period be Ts and the driving dead time be Td. When the duty cycle D1 of the first switch tube is affected by the driving dead time, the actual conduction time is Ts·D1 - Td; the duty cycle loss corresponding to the dead time is Dd = Td / Ts; when the duty cycle of the third switch tube is D3, the actual conduction time is Ts·D3 - Td = Ts·(D3 - Dd). Affected by the dead time, the actual volt-second voltages at both ends of the inductor are smaller than the theoretical values. When mode switching occurs, the actual volt-second relationship and the theoretical duty cycle that satisfy "Equation 1" should consider the dead time, which is:

[0159] Equation 3: V1 / V2 = (D3 阶跃前 - Dd) / (D1 阶跃前 - Dd) = (D3 阶跃后 - Dd) / (D1 阶跃后 - Dd)

[0160] When a duty cycle step occurs, according to "Equation 3", duty cycle compensation and non-linear correction should be carried out according to the actual duty cycle. The correction process is the same as the working principle described in the non-linear controller in Embodiment 1, and its basic derivation will not be elaborated here. In actual calculation, in order to avoid repeatedly iterating and correcting D1 and D3, the correction is carried out by selecting the maximum value, and the smaller value must be less than Dmax. Let the duty cycles after comparing and calculating the maximum values be D1 变换后2 , D3 变换后2 , and the correction process after simplified calculation is:

[0161] 1) According to the driving circuit dead time Td, calculate the dead time equivalent loss duty cycle Dd = Td / Ts.

[0162] 2) When vpid1 ≤ Dmax, let D1 变换后2 = vpid1, and let D3 变换后2 = 1.

[0163] 3) When vpid1 > Dmax and vpid1 < 1+(1 - Dmax), first calculate the temporary values D11 and D31 of the duty cycles of the first switch and the third switch, and calculate according to the formulas D11 = vpid1 - Dd and D31 = 2·vpid1max - vpid1 - Dd; compare the maximum values of the two:

[0164] When D11 ≥ D31, let D1 变换后2 = Dmax, and let D3 变换后2 = vpid1 / (2·vpid1max - vpid1)+Dd;

[0165] When D11 < D31, let D3 变换后2 = Dmax, and let D1 变换后2 = (2·vpid1max - vpid1) / vpid1+Dd;

[0166] 4) When vpid1 > 1+(1 + Dmax), let D1 变换后2 = 1, and let D3 变换后2 = vpid1 / (2·vpid1max - vpid1).

[0167] When considering the dead time, equivalently deduct the dead time on the basis of the non - linear controller in Embodiment 1, and its transformation process is similar to that of the non - linear controller in Embodiment 1.

[0168] Next, simulate the control method of the present invention to verify its application effect.

[0169] Independently simulate the non - linear controller proposed in Embodiment 1 to obtain the function relationship diagram of the input vpid1 to the outputs D1 and D3, as shown in Figure 4 . The simulation set value is Dmax = 0.9 and vpid1max = 1.8. It can be seen from the figure that when vpid1 < 0.9, then D3 = 1, and the duty cycle of D1 is calculated according to the linear transformation; when Dmax < vpid1 < 1.0, D3 and D1 are respectively clamped at the maximum value according to the control strategy, and the duty cycle transformation of the minimum value for equivalent volt - second balance is performed; when vpid1 > 1.0, D1 = 1.0 is clamped, and the duty cycle of D3 is calculated according to the linear transformation.

[0170] Independently simulate the non - linear controller proposed in Embodiment 2 to obtain the function relationship diagram of the input vpid1 to the outputs D1 and D3, as shown in Figure 5 ; Figure 5 The locally enlarged waveform of the duty cycle in the range of 0.6 - 1.1 is as shown in Figure 6。The simulation set values are Dmax = 0.9, vpid1max = 1.8, and the dead - zone duty cycle Dd = 0.05. The transformation process in the figure takes into account the dead - zone time and consists of multiple linear segments. The meanings of each segment are the same as those described in Figure 4 the same as described in

[0171] According to the example simulations of the non - linear controllers in the two embodiments, the input vpid1 to the outputs D1 and D3 undergoes a piece - wise linear transformation, and this transformation is suitable for implementation in a digital controller.

[0172] The practical application effects of the non - linear controllers in the two embodiments proposed in the present invention are verified by simulation. The simulation example used is a four - switch Buck - Boost converter in a bidirectional conversion scenario under a typical 48V energy storage application, with a power - stage and control - stage structure as shown in Figure 1 shown. The voltage V1 = 35 - 65V, the voltage V2 = 48V is regulated; the power flows from V1 to V2, and the associated direction of the inductor L1 current is greater than zero.

[0173] First, connect V2 to a voltage - source - type load with a voltage of 48V, and the voltage on the V1 side changes slowly from 65V to 35V. The reference value of the inductor current is given as 20A to verify the dynamic process of three - mode switching caused by the change in the V1 voltage:

[0174] Figure 7 For the mode - switching simulation of the four - switch Buck - Boost converter in the example without using the control method of the present invention, the time - domain dynamic waveforms of the mode - switching process are shown. When the V1 voltage drops to around 62V, the converter switches from the Buck mode to the Buck - Boost mode, and boundary - cycle jitter appears during the switching process, and the inductor current iL oscillates, with a maximum peak - to - peak value of 10.1A; when the V1 voltage drops to around 47V, the converter switches from the Buck - Boost mode to the Boost mode, and boundary - cycle jitter appears during the switching process, and the inductor current iL oscillates, with a maximum peak - to - peak value of 10.7A.

[0175] Figure 8 For the time - domain simulation of mode - switching of the four - switch Buck - Boost converter using the non - linear controller in Embodiment 1, the time - domain dynamic waveforms of the mode - switching process are shown. When the V1 voltage drops to around 62V, the converter switches from the Buck mode to the Buck - Boost mode, and there is no boundary - cycle jitter during the switching process. The inductor loop has a short - term adjustment dynamic process due to mode - switching, with a maximum peak - to - peak value of 9.1A, but the adjustment process is rapid and stable; when the V1 voltage drops to around 43V, the converter switches from the Buck - Boost mode to the Boost mode, and there is no boundary - cycle jitter during the switching process. The inductor loop has a short - term adjustment dynamic process due to mode - switching, with a maximum peak - to - peak value of 9.6A, but the adjustment process is rapid and stable.

[0176] Figure 9 For the time-domain simulation of mode switching when the four-switch Buck-Boost converter in the example adopts the non-linear controller in Embodiment 2, it shows the time-domain dynamic waveforms during the mode switching process; when the voltage V1 drops to around 62V, the converter switches from Buck mode to Buck-Boost mode, and there is no boundary cycle jitter during the switching process. The inductor loop has a transient adjustment dynamic process due to mode switching, with a maximum peak-to-peak value of 7.6A, but the adjustment process is rapid and stable; when the voltage V1 drops to around 43V, the converter switches from Buck-Boost mode to Boost mode, and there is no boundary cycle jitter during the switching process. The inductor loop has a transient adjustment dynamic process due to mode switching, with a maximum peak-to-peak value of 7.8A, but the adjustment process is rapid and stable.

[0177] Adopting the non-linear controllers in these two embodiments, during the mode switching process, compared with the traditional mode: it improves the dynamic response during the mode switching process, solves the problem of boundary cycle jitter, and the switching process is stable; due to the loop dynamic adjustment caused by switching, the current peak-to-peak value is smaller. Among them, the current peak-to-peak value of the dynamic process using the non-linear controller in Embodiment 2 is smaller than that of the non-linear controller in Embodiment 1, but both have good dynamic performance and have obvious beneficial effects compared with the traditional scheme.

[0178] Figure 10 For the time-domain simulation of mode switching when the four-switch Buck-Boost converter adopts the non-linear controller in Embodiment 2, it shows that the input voltage step causes the converter to switch modes, and the time-domain dynamic waveforms during the switching process. At 6.0ms, 8.0ms, 10.0ms, and 12.0ms, when the voltage V1 steps from 35V to 47V, to 65V, to 47V, and to 35V respectively, the voltage V2 adopts voltage stabilization control and is stabilized at 48V, with a resistive load. When the voltage V1 steps, the converter works in Boost mode, Buck-Boost mode, Buck mode, Buck-Boost mode, and Boost mode switching respectively. Figure 10 The dynamic waveforms of the voltage V2 and the inductor current iL are observed, and it can be seen that the switching process is stable and rapid, the dynamic adjustment process of the inductor current is rapid, and the switching process does not affect the adjustment processes of the current and voltage loops.

[0179] The following conclusions are obtained through simulation:

[0180] 1. The controller and control method of the present invention, when applied in a four-switch Buck-Boost converter, can effectively smooth the switching process among Buck, Buck-Boost, and Boost modes, and suppress the boundary cycle jitter during the three-mode switching process;

[0181] 2. The controller and control method according to the present invention have good dynamic performance when the input voltage is disturbed and do not affect the control effects of the original voltage and current loop compensators.

[0182] In specific implementation, the present application provides a computer storage medium and a corresponding data processing unit. Among them, the computer storage medium can store a computer program, and when the computer program is executed by the data processing unit, it can run the inventive concepts and some or all of the steps in each embodiment of a controller and a control method for smooth mode switching of a four-switch buck-boost converter provided by the present invention. The storage medium can be a magnetic disk, an optical disk, a read-only memory (ROM), a random access memory (RAM), or the like.

[0183] Those skilled in the art can clearly understand that the technical solutions in the embodiments of the present invention can be implemented by means of a computer program and its corresponding general hardware platform. Based on such an understanding, the essence of the technical solutions in the embodiments of the present invention, or the part that contributes to the prior art, can be embodied in the form of a computer program, that is, a software product. The computer program software product can be stored in a storage medium and includes several instructions for causing a device (which can be a personal computer, a server, a single-chip microcomputer, an MCU, or a network device, etc.) including a data processing unit to execute the methods described in each embodiment or some parts of the embodiments of the present invention.

[0184] The present invention provides an idea and method for a controller and a control method for smooth mode switching of a four-switch buck-boost converter. There are many methods and ways to specifically implement this technical solution. The above description is only the preferred embodiment of the present invention. It should be noted that for those of ordinary skill in the art, without departing from the principle of the present invention, several improvements and refinements can be made, and these improvements and refinements should also be regarded as the protection scope of the present invention. Each component not clearly defined in this embodiment can be implemented by the prior art.

Claims

1. A controller for smooth mode switching of a four-switch buck-boost converter, the four-switch buck-boost converter comprising a first switch tube S1, a second switch tube S2, a third switch tube S3, a fourth switch tube S4, a first capacitor C1, a second capacitor C2 and an inductor L1, wherein: The source of the first switch tube S1 and the drain of the second switch tube S2 are connected in series to form a first half bridge, the source of the third switch tube S3 and the drain of the fourth switch tube S4 are connected in series to form a second half bridge, the two ends of the inductor L1 are connected between the source of the first switch tube S1 and the source of the third switch tube S3, the first capacitor C1 is connected between the drain of the first switch tube S1 and the source of the second switch tube S2, the second capacitor C2 is connected between the drain of the third switch tube S3 and the source of the fourth switch tube S4, and the source of the second switch tube S2 is connected to the source of the fourth switch tube S4; The voltage across the first capacitor C1 is V1, and the voltage across the second capacitor C2 is V2. The controller includes: A voltage sampler and a current sampler, used to obtain the voltage across the second capacitor C2 and the current of the inductor L1 respectively; The voltage sampler and the current sampler are connected to the second loop compensator PID2 and the first loop compensator PID1 respectively, and then cascaded and outputted to the nonlinear controller; The output signal of the nonlinear controller controls the first switch tube S1 , the second switch tube S2 , the third switch tube S3 and the fourth switch tube S4 through the driving amplifier module.

2. The controller for smooth mode switching of a four-switch buck-boost converter according to claim 1, characterized in that: The controller specifically comprises: The second loop compensator PID2 takes as input an error signal ve2 between the vf2 signal output by the voltage sampler and the voltage reference vr2 signal, that is, ve2=vr2-vf2, and outputs a vr1 signal; The first loop compensator PID1 receives as input an error signal ve1 between the vf1 signal output by the current sampler and the vr1 signal, that is, ve1=vr1-vf1, and outputs a vpid1 signal; The vpid1 signal is input to a nonlinear controller to generate four PWM signals PWM_D1, PWM_D2, PWM_D3 and PWM_D4 for controlling the first switch tube S1, the second switch tube S2, the third switch tube S3 and the fourth switch tube S4; The four PWM signals, after passing through the driving amplifier module, are respectively connected to the gates of the first switch tube S1 , the second switch tube S2 , the third switch tube S3 and the fourth switch tube S4 to complete the control.

3. A control method for smooth mode switching of a four-switch buck-boost converter, characterized in that: The following steps are involved: Step 1, calculating a vpid1 signal according to an output voltage signal of the four-switch buck-boost converter, an inductor current signal in the four-switch buck-boost converter, and a reference voltage signal; Step 2, using a nonlinear controller to calculate control signals of four switch tubes in the four-switch buck-boost converter according to the vpid1 signal; Step 3, using the control signal obtained in step 2 to control the four switch tubes to achieve smooth mode switching of the four-switch buck-boost converter.

4. The control method for smooth mode switching of a four-switch buck-boost converter according to claim 3, characterized in that: Calculate the vpid1 signal as described in step 1, including: Step 1-1, using the second loop compensator PID2, converting the error signal ve2 between the vf2 signal output by the voltage sampler and the voltage reference vr2 signal, that is, ve2=vr2-vf2, into a vr1 signal output; Step 1-2: Use the first loop compensator PID1 to convert the error signal ve1 between the vf1 signal output by the current sampler and the vr1 signal, i.e., ve1 = vr1 - vf1, into a vpid1 signal for output.

5. The control method for smooth mode switching of a four-switch buck-boost converter according to claim 3, characterized in that: In step 3, the use of the control signal obtained in step 2 to control the four switching tubes includes: Use a drive amplification module to amplify the control signal and then use it to control the gates of the four switching tubes in the four-switch buck-boost converter respectively.

6. The control method for smooth mode switching of a four-switch buck-boost converter according to claim 3, characterized in that: In step 2, calculating the control signals of the four switching tubes in the four-switch buck-boost converter according to the vpid1 signal includes: Step A1: Judge the vpid1 signal input to the nonlinear controller, specifically including: When vpid1 ≤ Dmax, execute step 2, where Dmax is the maximum duty cycle value of the duty cycle D1 of the first switching tube S1 and the duty cycle D3 of the third switching tube S3; When vpid1 > Dmax and vpid1 < 1+(1 - Dmax), execute step A3; When vpid1 > 1+(1 + Dmax), execute step A4; Step A2: Set: D1 = vpid1 D3＝1 Execute step A7; Step A3: Calculate the temporary values D11 and D31 of the duty cycles of the first switching tube S1 and the third switching tube S3; make a judgment according to the temporary values D11 and D31, specifically including: When D11 ≥ D31, execute step A5; When D11 < D31, execute step A6; Step A4: Set: D1＝1 D3 = vpid1 / (2·vpid1max - vpid1) Execute step A7; Step A5: Set: D1 = Dmax D3 = vpid1 / (2·vpid1max - vpid1) Execute step A7; Step A6: Set: D3 = Dmax D1 = (2·vpid1max - vpid1) / vpid1 Execute step A7; Step A7: Use the values of D1 and D3 as modulation wave values respectively, compare them with the triangular carrier wave Tsw1 to obtain the PWM signal PWM_D1 of the first switching tube S1 and the PWM signal PWM_D3 of the third switching tube S3, and obtain the PWM signal PWM_D2 of the second switching tube S2 and the PWM signal PWM_D4 of the fourth switching tube S4 according to their inverted phase signals respectively.

7. The control method for smooth mode switching of a four-switch buck-boost converter according to claim 6, characterized in that: In step A3, the specific method for calculating the temporary values D11 and D31 of the duty cycles of the first switching tube S1 and the third switching tube S3 includes: D11 = vpid1 D31 = 2·vpid1max - vpid1 Where vpid1max is the maximum output value of the first loop compensator PID1.

8. The control method for smooth mode switching of a four-switch buck-boost converter according to claim 7, characterized in that: In step 2, calculating the control signals of the four switching tubes in the four-switch buck-boost converter according to the vpid1 signal includes: Step B1: Calculate the dead-time equivalent loss duty cycle Dd according to the dead time Td of the drive amplification module; Step B2: Judge the vpid1 signal input to the nonlinear controller, specifically including: When vpid1 ≤ Dmax, step B3 is executed; where Dmax is the maximum duty cycle value of the duty cycle D1 of the first switching transistor S1 and the duty cycle D3 of the third switching transistor S3. When vpid1 > Dmax and vpid1 < 1 + (1 - Dmax), step B4 is executed. When vpid1 > 1 + (1 + Dmax), step B5 is executed. Step B3, set: D1 = vpid1 D3＝1 Execute step B8. Step B4, calculate the temporary duty cycle values D11 and D31 of the duty cycles of the first switching transistor S1 and the third switching transistor S3; make a judgment based on the temporary duty cycle values D11 and D31, specifically including: When D11 ≥ D31, step B6 is executed. When D11 < D31, step B7 is executed. Step B5, set: D1＝1 D3 = vpid1 / (2·vpid1max - vpid1) Execute step B8. Step B6; set D1 = Dmax Execute step B8. Step B7, set: D3 = Dmax D1 = (2·vpid1max - vpid1) / vpid1 + Dd Execute step B8. Step B8: Use the values of D1 and D3 as modulation wave values respectively, compare them with the triangular carrier wave Tsw1 to obtain the PWM signal PWM_D1 of the first switching transistor S1 and the PWM signal PWM_D3 of the third switching transistor S3, and take their inverted signals as the PWM signal PWM_D2 of the second switching transistor S2 and the PWM signal PWM_D4 of the fourth switching transistor S4 respectively.

9. The control method for smooth mode switching of a four-switch buck-boost converter according to claim 8, characterized in that: The calculation of the dead - zone equivalent loss duty cycle Dd described in step B1 is specifically as follows: Where Ts is the switching cycle time.

10. The control method for smooth mode switching of a four-switch buck-boost converter according to claim 9, characterized in that: The specific method for calculating the temporary duty cycle values D11 and D31 of the duty cycles of the first switching transistor S1 and the third switching transistor S3 described in step B4 includes: D11 = vpid1 - Dd D31 = 2·vpid1max - vpid1 - Dd Where vpid1max is the maximum output value of the first loop compensator PID1.

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