Method and device for online self-optimizing parameters of current regulator and control method and device

Abstract

The invention discloses a method and a device for online self-optimizing parameters of a current regulator and a control method and a device for online self-optimizing parameters of the current regulator, and the method for online self-optimizing parameters of the current regulator comprises the following steps: subtracting a given amplitude of a motor stator current from a current amplitude of the motor stator current, and carrying out differential calculation; according to the difference and the differential of the motor stator current amplitude and the optimal bandwidth of the current control loop at the previous moment, an adjustment rule is formulated, and an adjustment rule table is generated; the self-optimization bandwidth of the current control loop at the current moment is obtained by searching the adjustment rule table; and according to the current control loop self-optimization bandwidth at the current moment, acquiring parameters of an online self-optimization current controller of the current control loop at the current moment. According to the method, the static and dynamic performance of the motor driving system can be improved at the same time under the condition of ensuring the operation stability of the motor, the influence of the resonance phenomenon in the LC filtering type permanent magnet synchronous motor control system on the stability is effectively solved, and the static and dynamic performance of the motor driving system facing different working conditions is improved.

Description

Technical Field

[0001] The present invention belongs to the field of permanent magnet synchronous motor control, and particularly relates to a method and device for online self-optimizing the parameters of a current regulator, and a control method and device. Background Art

[0002] In industrial applications, the interior permanent magnet synchronous motor (IPMSM) has been widely used due to its advantages in power density, operating efficiency, and control performance. The IPMSM is usually directly powered by a voltage source inverter (VSI), and pulse width modulation (PWM) technology is used to regulate the output voltage of the VSI. However, the rapid switching of the power switching devices of the VSI will lead to a high voltage change rate (du / dt), which brings some problems, such as increasing the insulation pressure on the motor side, causing shaft voltage and bearing current, and shortening the service life of the motor. To eliminate the above problems, adding an LC sine filter between the VSI and the IPMSM is an effective method. However, the addition of the LC sine filter increases the order of the motor control system, and the operating stability of the current control loop in the flux-oriented controller (FOC) will inevitably be affected by the inherent resonance phenomenon.

[0003] Generally, in order to overcome the influence of the resonance phenomenon on the stability of the FOC, passive damping and active damping methods can be adopted in the control structure to increase the damping effect of the control system. However, most passive damping methods achieve the function of resonance suppression by adding appropriate damping resistors to the controlled object. And adding additional resistors will cause unnecessary power loss, which will have a serious impact on the operating efficiency of the motor drive system. The characteristic of active damping is that a damping resistor can be virtualized in the control system through the state variables measured in real time. Such a virtual resistor can eliminate the resonance problem of the control system without increasing additional power loss. However, the selection of the control parameters of active damping will affect the operating stability of the control system. Therefore, this method will put forward higher requirements for the selection of the control loop parameters. At the same time, compared with the traditional motor drive system, the control method of combining FOC with active damping can ensure the operating stability of the system, but it is more difficult to improve the static and dynamic performance of the motor drive system at the same time, which will reduce the operating performance of the motor under different working conditions. Summary of the Invention

[0004] The object of the present invention is to provide a method and device for online self-optimizing the parameters of a current regulator, and a control method and device, which are used to solve the problems that it is difficult to select the stable control parameters of a traditional LC-filtered permanent magnet synchronous motor, and it is difficult to balance the static and dynamic performances of the motor drive system under different working conditions. The present invention can not only adjust the parameters of the control loop according to different operating conditions to ensure the operating stability of the LC-filtered permanent magnet synchronous motor, but also improve the static and dynamic performances of the motor drive system at the same time.

[0005] The present invention is implemented according to the following technical solutions:

[0006] In the first aspect, the present invention provides a method for online self-optimizing the parameters of a current regulator, and the method includes:

[0007] Subtract the given amplitude of the motor stator current from the amplitude of the motor stator current, and perform differential calculation;

[0008] Formulate an adjustment rule based on the difference, differential of the motor stator current amplitude, and the optimal bandwidth of the current control loop at the previous moment, and generate an adjustment rule table;

[0009] Obtain the self-optimizing bandwidth of the current control loop at the current moment by looking up the adjustment rule table;

[0010] According to the self-optimizing bandwidth of the current control loop at the current moment, obtain the parameters of the online self-optimizing current controller of the current control loop at the current moment.

[0011] The adjustment rule formulation is:

[0012] When |Δi sd (k)| > E high And When E low <|Δi sd (k)| ≤ E high And When 0 < |Δi sd (k)| ≤ E low And When 0 < |Δi sd (k)| ≤ E low And When |Δi sq (k)| > E high And When E low <|Δi sq (k)| ≤ E high And When 0 < |Δi sq (k)| ≤ E low And When 0 < |Δi sq(k) | ≤ E low and at time k, the calculation rule for the optimal bandwidth of the current control loop is as follows:

[0013] ω bd (k) = ω bd (k - 1)

[0014] ω bq (k) = ω bq (k - 1)

[0015] When E low <|Δi sd (k)| ≤ E high and at time, when 0 < |Δi sd (k)| ≤ E low and at time, when E low <|Δi sq (k)| ≤ E high and at time and when 0 < |Δi sq (k)| ≤ E low and at time, the calculation rule for the optimal bandwidth of the current control loop is as follows:

[0016] ω bd (k) = 0.998ω bd (k - 1)

[0017] ω bq (k) = 0.998ω bq (k - 1)

[0018] When |Δi sd (k)| > E high and at time, when E low <|Δi sd (k)| ≤ E high and at time, when |Δi sq (k)| > E high and at time and when E low <|Δi sq (k)| ≤ E high and at time, the calculation rule for the optimal bandwidth of the current control loop is as follows:

[0019] ω bd (k) = 1.002ω bd (k - 1)

[0020] ω bq(k) = 1.002ω bq (k - 1)

[0021] When |Δi sd (k)| > E high and at this time, when |Δi sq (k)| > E high and at this time, the calculation rule of the optimal bandwidth of the current control loop corresponding to the k - th moment is as follows:

[0022] ω bd (k) = 1.004ω bd (k - 1)

[0023] ω bq (k) = 1.004ω bq (k - 1)

[0024] Among them, E high represents the upper limit of the absolute value of the difference Δi sd of the stator current amplitude of the deviation - value motor, and Δi sq ; E low represents the lower limit of the absolute value of the difference Δi sd of the stator current amplitude of the deviation - value motor, and Δi sq ; E chigh represents the absolute - value upper - limit value of the differential of the stator current amplitude difference of the motor ; E clow represents the absolute - value lower - limit value of the differential of the stator current amplitude difference of the motor ; ω bd (k), ω bd (k - 1), ω bq (k), ω bq (k - 1) are the optimal bandwidths of the d - axis and q - axis current control loops at the k - th moment and the (k - 1) - th moment respectively.

[0025] In an implementation manner, the online self - optimizing controller parameters include k pd (k), k id (k), k pq (k), k iq (k), and their expressions are:

[0026]

[0027] Among them, T d represents the time constant, T s represents the sampling period, R s represents the stator resistance of the motor, L f represents the inductance value of the LC filter, C f represents the capacitance value of the LC filter, Ld Represents the d-axis inductance value of the permanent magnet synchronous motor, L q Represents the q-axis inductance value of the permanent magnet synchronous motor.

[0028] In a second aspect, the present invention provides a device for online self-optimizing the parameters of a current regulator. The device includes:

[0029] A stator current deviation module, configured to calculate the difference in the amplitude of the motor stator current according to the given amplitude of the motor stator current and the amplitude of the motor stator current;

[0030] A stator current deviation differential module, configured to perform a differential calculation on the difference in the amplitude of the motor stator current;

[0031] An optimal adjustment rule module, configured to formulate an adjustment rule according to the difference in the amplitude of the motor stator current, the differential, and the optimal bandwidth of the current control loop at the previous moment, and generate an adjustment rule table;

[0032] A controller parameter tuning module, configured to obtain the self-optimizing bandwidth of the current control loop at the current moment by looking up the adjustment rule table, and obtain the parameters of the online self-optimizing current controller of the current control loop according to the self-optimizing bandwidth of the current control loop at the current moment.

[0033] In a third aspect, the present invention provides an online liberalized current regulator module. The module includes:

[0034] The above-mentioned device for online self-optimizing the parameters of a current regulator, configured to obtain the parameters of the online self-optimizing current controller of the current control loop at the current moment;

[0035] A PI controller module, configured to calculate a resonant voltage signal according to the parameters of the online self-optimizing controller of the current control loop at the current moment and the difference in the amplitude of the motor stator current;

[0036] An active damping parameter self-adjusting module, configured to obtain a resonant suppression signal according to the capacitor current and the parameters of the online self-optimizing controller of the current control loop at the current moment;

[0037] An inverter output voltage command module, configured to calculate the inverter output voltage in the two-phase rotating coordinate system at the kth moment through the resonant voltage signal and the resonant suppression signal.

[0038] In an implementation manner, the resonant voltage signal u idr (k), u iqr (k), and its expression is:

[0039]

[0040] Where k pd (k), k pq(k), k id (k), k iq (k) represents the parameter of the on-line self-optimizing current controller of the current control loop at time k, Δi sd (k), Δi sq (k) represents the difference in the amplitude of the stator current of the motor at time k, and s represents the Laplace operator.

[0041] In one embodiment, the resonance suppression signal u ids (k), u iqs (k), and its expression is:

[0042]

[0043] Among them, T s represents the sampling period, R s represents the stator resistance of the motor, L f represents the inductance value of the LC filter, C f represents the capacitance value of the LC filter, L d represents the d-axis inductance value of the permanent magnet synchronous motor, L q represents the q-axis inductance value of the permanent magnet synchronous motor, ω resd represents the d-axis resonance frequency, ω resq represents the q-axis resonance frequency, cos(ω resd T s ) represents the cosine value of the d-axis resonance component, sin(ω resd T s ) represents the sine value of the d-axis resonance component, cos(ω resq T s ) represents the cosine value of the q-axis resonance component, sin(ω resq T s ) represents the sine value of the q-axis resonance component.

[0044] Fourthly, the present invention provides an LC filter type permanent magnet synchronous motor control method, and the method includes:

[0045] Obtain the inverter-side current, the stator current of the motor, and the electrical angular velocity information of the motor at time k in the three-phase stationary rotating coordinate system, and calculate the stator current and the capacitor current of the motor in the two-phase rotating coordinate system;

[0046] Based on the electrical angular velocity of the motor and the given speed information of the motor at time k, the given amplitude of the stator current of the motor;

[0047] Adopt the above on-line liberalized current regulator module to calculate the output voltage of the inverter in the two-phase rotating coordinate system at time k;

[0048] According to the inverter output voltage command in the two-phase rotating coordinate system at time k, the inverter switching state at time k is obtained and applied to the switching devices of the three-phase voltage source inverter.

[0049] In a fifth aspect, the present invention provides an LC-filtered permanent magnet synchronous motor control device, which includes:

[0050] A signal acquisition module, configured to obtain the inverter-side current, the motor stator current, and the motor electrical angular velocity information at time k in the three-phase stationary rotating coordinate system;

[0051] A current transformation module, configured to calculate the motor stator current and the capacitor current in the two-phase rotating coordinate system;

[0052] A motor stator current command generation module, configured to obtain the given amplitude of the motor stator current based on the motor electrical angular velocity and the given motor speed information at time k;

[0053] The above-mentioned online liberalized current regulation module is configured to calculate the inverter output voltage in the two-phase rotating coordinate system at time k;

[0054] A three-phase voltage source inverter module, configured to obtain the inverter switching state at time k according to the inverter output voltage in the two-phase rotating coordinate system at time k and apply it to the switching devices of the three-phase voltage source inverter.

[0055] In a sixth aspect, the present invention provides an LC-filtered permanent magnet synchronous motor control system, which includes the above-mentioned LC-filtered permanent magnet synchronous motor control device and an LC filter module.

[0056] Advantages of the present invention:

[0057] The present invention can improve the static and dynamic performance of the motor drive system while ensuring the stability of the motor operation, effectively solving the influence of the resonance phenomenon on the stability in the LC-filtered permanent magnet synchronous motor control system, and improving the static and dynamic performance of the motor drive system under different working conditions. Description of the Drawings

[0058] The drawings, as a part of the present invention, are used to provide a further understanding of the present invention. The schematic embodiments and descriptions of the present invention are used to explain the present invention, but do not constitute an improper limitation to the present invention. Obviously, the drawings in the following description are only some embodiments, and those of ordinary skill in the art can obtain other drawings without creative efforts based on these drawings.

[0059] Figure 1 It is a flowchart of a method for online self-optimizing the parameters of a current regulator provided by an embodiment of the present invention;

[0060] Figure 2Flow chart of an LC filter type permanent magnet synchronous motor control method provided by an embodiment of the present invention;

[0061] Figure 3 Structure diagram of an LC filter type permanent magnet synchronous motor control system provided by an embodiment of the present invention;

[0062] Figure 4 Comparison chart of experimental results of traditional method and the method proposed by the present invention (motor suddenly adds 27 N·m load and suddenly reduces 27 N·m load);

[0063] Figure 5 Comparison chart of experimental results of traditional method and the method proposed by the present invention (motor runs at no load from 750 r / min to 1200 r / min and then decelerates to 750 r / min).

[0064] It should be noted that these drawings and text descriptions are not intended to limit the scope of the concept of the present invention in any way, but to illustrate the concept of the present invention to those skilled in the art by referring to specific embodiments. Detailed implementation manners

[0065] To make the objectives, technical solutions and advantages of the embodiments of the present invention clearer, the technical solutions in the embodiments of the present invention will be clearly and completely described below with reference to the accompanying drawings in the embodiments of the present invention. The following embodiments are used to illustrate the present invention, but are not used to limit the scope of the present invention.

[0066] To deepen the understanding and recognition of the present invention, the technical solutions of the present invention will be further introduced below with reference to the accompanying drawings and specific implementation manners.

[0067] The following further elaborates on the detailed implementation manners of the present invention:

[0068] As Figure 1 shown, an embodiment of the present invention provides a method for online self-optimizing the parameters of a current regulator, and the method specifically includes the following steps:

[0069] Step S311: Subtract the given amplitude of the motor stator current from the amplitude of the motor stator current, and perform differential calculation.

[0070] Specifically, the given amplitudes of the motor stator current i sd * 、i sq * can be calculated from the motor electrical angular velocity information ω e collected at time k, and the given motor speed N r * .

[0071] According to the calculated given amplitudes of the motor stator current (i sd* , i sq * ), and the stator current amplitude of the motor (i sd , i sq ) to obtain the difference in the stator current amplitude of the motor (Δi sd , Δi sq ), and calculate to obtain:

[0072]

[0073] The differential of the difference in the stator current amplitude of the motor is

[0074] Step S312: According to the difference, differential of the stator current amplitude of the motor, and the optimal bandwidth of the current control loop at the previous moment, formulate an adjustment rule and generate an adjustment rule table.

[0075] Furthermore, the adjustment rule is formulated as:

[0076] When |Δi sd (k)| > E high and , the calculation rule for the optimal bandwidth of the current control loop at the k-th moment is as follows:

[0077] ω bd (k) = ω bd (k - 1)

[0078] When E low < |Δi sd (k)| ≤ E high and , the calculation rule for the optimal bandwidth of the current control loop at the k-th moment is as follows:

[0079] ω bd (k) = 0.998ω bd (k - 1)

[0080] When 0 < |Δi sd (k)| ≤ E low and , the calculation rule for the optimal bandwidth of the current control loop at the k-th moment is as follows:

[0081] ω bd (k) = 0.998ω bd (k - 1)

[0082] When |Δi sd (k)| > E high and , the calculation rule for the optimal bandwidth of the current control loop at the k-th moment is as follows:

[0083] ω bdω(k) = 1.002ω(k - 1) bd (k - 1)

[0084] When E low <|Δi sd (k)| ≤ E high and the optimal bandwidth calculation rule for the current control loop at time k is as follows:

[0085] ω bd (k) = ω bd (k - 1)

[0086] When 0 < |Δi sd (k)| ≤ E low and the optimal bandwidth calculation rule for the current control loop at time k is as follows:

[0087] ω bd (k) = ω bd (k - 1)

[0088] When |Δi sd (k)| > E high and the optimal bandwidth calculation rule for the current control loop at time k is as follows:

[0089] ω bd (k) = 1.004ω bd (k - 1)

[0090] When E low <|Δi sd (k)| ≤ E high and the optimal bandwidth calculation rule for the current control loop at time k is as follows:

[0091] ω bd (k) = 1.002ω bd (k - 1)

[0092] When 0 < |Δi sd (k)| ≤ E low and the optimal bandwidth calculation rule for the current control loop at time k is as follows:

[0093] ω bd (k) = ω bd (k - 1)

[0094] When |Δi sq (k)| > E high and the optimal bandwidth calculation rule for the current control loop at time k is as follows:

[0095] ω bq (k) = ω bq (k - 1)

[0096] When E low <|Δi sq (k)| ≤ E high and at this time, the calculation rule of the optimal bandwidth of the current control loop at the corresponding k - moment is as follows:

[0097] ω bq (k) = 0.998ω bq (k - 1)

[0098] When 0 <|Δi sq (k)| ≤ E low and at this time, the calculation rule of the optimal bandwidth of the current control loop at the corresponding k - moment is as follows:

[0099] ω bq (k) = 0.998ω bq (k - 1)

[0100] When |Δi sq (k)| > E high and at this time, the calculation rule of the optimal bandwidth of the current control loop at the corresponding k - moment is as follows:

[0101] ω bq (k) = 1.002ω bq (k - 1)

[0102] When E low <|Δi sq (k)| ≤ E high and at this time, the calculation rule of the optimal bandwidth of the current control loop at the corresponding k - moment is as follows:

[0103] ω bq (k) = ω bq (k - 1)

[0104] When 0 <|Δi sq (k)| ≤ E low and at this time, the calculation rule of the optimal bandwidth of the current control loop at the corresponding k - moment is as follows:

[0105] ω bq (k) = ω bq (k - 1)

[0106] When |Δi sq (k)| > E high and At time \(k\), the calculation rule for the optimal bandwidth of the current control loop is as follows:

[0107] \(\omega\) bq \((k)=1.004\omega\) bq (k - 1)

[0108] When \(E\) low \(<|\Delta i\) sq (k)|\leq E\) high And At this time, the calculation rule for the optimal bandwidth of the current control loop at time \(k\) is as follows:

[0109] \(\omega\) bq (k)=1.002\omega\) bq (k - 1)

[0110] When \(0\lt|\Delta i\) sq (k)|\leq E\) low And At this time, the calculation rule for the optimal bandwidth of the current control loop at time \(k\) is as follows:

[0111] \(\omega\) bq (k)=\omega\) bq (k - 1)

[0112] Among them, \(E\) high Represents the upper limit of the absolute value of the difference in the amplitude of the stator current of the motor with deviation (\(\Delta i\) sd , \(\Delta i\) sq ), \(E\) low Represents the lower limit of the absolute value of the difference in the amplitude of the stator current of the motor with deviation (\(\Delta i\) sd , \(\Delta i\) sq ), \(E\) chigh Represents the upper limit of the absolute value of the differential of the difference in the amplitude of the stator current of the motor , \(E\) clow Represents the lower limit of the absolute value of the differential of the difference in the amplitude of the stator current of the motor .

[0113] According to the formulated adjustment rules, the following adjustment rule table is generated:

[0114]

[0115]

[0116] Specifically, the self-optimized bandwidth \(\omega\) bd (k), \(\omega\) bq (k) of the current control loop at the current moment can be obtained from the calculated difference in the amplitude of the stator current of the motor (\(\Delta i\) sd , \(\Delta i\) sq ) and the differential of the difference in the amplitude of the stator current of the motor Optimal bandwidth ω of the current control loop at the previous moment bd (k - 1), ω bq (k - 1), obtained by looking up the rules.

[0117] Step S313: Obtain the self - optimized bandwidth of the current control loop at the current moment by looking up the adjustment rule table.

[0118] Step S314: Obtain the parameters of the online self - optimized current controller of the current - moment current control loop according to the self - optimized bandwidth of the current - moment current control loop.

[0119] Specifically, the parameter k of the online self - optimized controller of the current - moment current control loop pd (k), k id (k), k pq (k), k iq (k) can be obtained from the self - optimized bandwidth ω of the current - moment current control loop calculated bd (k), ω bq (k), and the calculation is as follows:

[0120]

[0121] Among them, T d represents the time constant, T s represents the sampling period, R s represents the stator resistance of the motor, L f represents the inductance value of the LC filter, C f represents the capacitance value of the LC filter, L d represents the d - axis inductance value of the permanent - magnet synchronous motor, L q represents the q - axis inductance value of the permanent - magnet synchronous motor.

[0122] The following is an embodiment of the device for online self - optimizing the parameters of the current regulator of the present invention, which can be used to execute the method embodiment of online self - optimizing the parameters of the current regulator of the present invention. For the details not disclosed in the embodiment of the device for online self - optimizing the parameters of the current regulator of the present invention, please refer to the method embodiment of online self - optimizing the parameters of the current regulator of the present invention.

[0123] In one embodiment, a device for online self - optimizing the parameters of the current regulator is proposed. Referring to Figure 3 the content shown in the online self - optimizing current regulator module, this device includes: The online self - optimizing current regulator module 2 includes a stator current deviation module 201, a stator current deviation differential module 202, an optimal adjustment rule module 203, and a controller parameter tuning module 204. Among them,

[0124] The stator current deviation module 201 is used to calculate the difference between the stator current amplitude of the motor according to the stator current amplitude of the motor and the stator current of the motor;

[0125] The stator current deviation differential module 202 is used to perform differential calculation on the difference in the amplitude of the motor stator current;

[0126] The optimal adjustment rule module 203 is used to formulate an adjustment rule based on the difference in the amplitude of the motor stator current, the differential, and the optimal bandwidth of the current control loop at the previous moment, and generate an adjustment rule table;

[0127] The controller parameter tuning module 204 is used to obtain the self-optimized bandwidth of the current control loop at the current moment by looking up the adjustment rule table, and obtain the parameters of the online self-optimized current controller of the current control loop according to the self-optimized bandwidth of the current control loop at the current moment.

[0128] Specifically, the input of the stator current deviation module 201 is the given amplitude of the motor stator current (i sd * , i sq * ) and the amplitude of the motor stator current obtained by the signal acquisition module 3 (i sd , i sq ). Its output is the difference in the amplitude of the motor stator current (△i sd , △i sq ). The input of the stator current deviation differential module 202 is the difference in the amplitude of the motor stator current (△i sd , △i sq ). Its output is the differential of the difference in the amplitude of the motor stator current The input of the optimal adjustment rule module 203 is the difference in the amplitude of the motor stator current at the current moment (△i sd (k), △i sq (k)), the difference in the amplitude of the motor stator current and the optimal bandwidth ω of the current control loop at the previous moment bd (k - 1), ω bq (k - 1). The output is the self-optimized bandwidth ω of the current control loop at the current moment bd (k), ω bq (k). The input of the controller parameter tuning module 204 is the self-optimized bandwidth ω of the current control loop at the current moment bd (k), ω bq (k). The output is the online self-optimized controller parameter k of the current control loop at the current moment pd (k), k id (k).

[0129] It should be noted that each functional module in the embodiments of the present invention can be integrated into a processing module, or each unit can exist physically alone, or two or more units can be integrated into one module. The above integrated module can be implemented in the form of hardware or in the form of a software functional module.

[0130] In one embodiment, an online self-optimizing current regulator module is proposed, and this module includes:

[0131] A device for online self-optimizing the parameters of a current regulator, which is used to obtain the parameters of the online self-optimizing current controller of the current control loop at the current moment;

[0132] A PI controller module 205, which is used to calculate a resonant voltage signal according to the parameters of the online self-optimizing controller of the current control loop at the current moment and the difference in the amplitude of the motor stator current;

[0133] An active damping parameter self-adjusting module 206, which is used to obtain a resonant suppression signal according to the capacitor current and the parameters of the online self-optimizing controller of the current control loop at the current moment;

[0134] An inverter output voltage command module 207, which is used to calculate the inverter output voltage through the resonant voltage signal and the resonant suppression signal.

[0135] It should be noted that the device for online self-optimizing the parameters of the current regulator is described with reference to the foregoing same or similar parts, and will not be elaborated here.

[0136] Furthermore, the parameters k pd (k), k id (k), k pq (k), k iq (k) of the online self-optimizing controller of the current control loop at the current moment and the difference in the amplitude of the motor stator current (△i sd , △i sq ) output a resonant voltage signal (u idr (k), u iqr (k)), and its calculation process is as follows:

[0137]

[0138] Among them, k pd (k), k pq (k), k id (k), k iq (k) represent the parameters of the online self-optimizing current controller of the current control loop at the kth moment, Δi sd (k), Δi sq (k) represent the difference in the amplitude of the motor stator current at the kth moment, and s represents the Laplace operator.

[0139] Further, the obtained LC filter capacitor currents (i cd , i cq ) and the current control loop online self-optimizing controller parameters k pd (k), k pq (k) whose output is the resonance suppression signal (u ids (k), u iqs (k)), and the calculation process is as follows:

[0140]

[0141] Among them, T s represents the sampling period, R s represents the motor stator resistance, L f represents the LC filter inductance value, C f represents the LC filter capacitance value, L d represents the direct-axis inductance value of the permanent magnet synchronous motor, L q represents the quadrature-axis inductance value of the permanent magnet synchronous motor, ω resd represents the direct-axis resonance frequency, ω resq represents the quadrature-axis resonance frequency, cos(ω resd T s ) represents the cosine value of the direct-axis resonance component, sin(ω resd T s ) represents the sine value of the direct-axis resonance component, cos(ω resq T s ) represents the cosine value of the quadrature-axis resonance component, sin(ω resq T s ) represents the sine value of the quadrature-axis resonance component.

[0142] Referring to Figure 2 shown, an embodiment of the present invention provides an LC filter type permanent magnet synchronous motor control method, and the specific steps of the method include:

[0143] Step S100: Obtain the inverter-side current, motor stator current, and motor electrical angular velocity information at the kth moment in the three-phase stationary rotating coordinate system, and calculate the motor stator current and capacitor current in the two-phase rotating coordinate system.

[0144] Further, the inverter-side currents i ia , i ib , i ic , the motor stator currents i sa , i sb , i sc , and the motor electrical angular velocity information ω e can be obtained from the inverter-side current i in the three-phase stationary coordinate system at the kth moment.ia , i ib , the stator current i of the motor sa , i sb , the electrical angle information θ of the motor rotor e , it is calculated that:

[0145]

[0146] The stator current i of the motor in the two-phase rotating coordinate system sd , i sq , the capacitor current i cd , i cq can be obtained from the inverter-side current i in the three-phase stationary coordinate system at time k ia , i ib , i ic , the stator current i of the motor sa , i sb , i sc , the electrical angle information θ of the motor rotor e , it is calculated that:

[0147]

[0148] Step S200: Based on the electrical angular velocity of the motor and the given speed information of the motor at time k, obtain the given amplitude of the stator current of the motor.

[0149] Furthermore, the given amplitude i of the stator current of the motor sd * , i sq * can be obtained from the electrical angular velocity information ω of the motor collected at time k e , the given speed N of the motor r * , it is calculated that:

[0150]

[0151] where k p , k i respectively represent the proportional coefficient and integral coefficient of the speed loop PI controller, and n p represents the number of pole pairs of the permanent magnet synchronous motor, and s represents the Laplace operator.

[0152] Step S300: Adopt an online liberalized current regulator module to calculate the output voltage of the inverter in the two-phase rotating coordinate system at time k.

[0153] In the embodiment of the present application, adopting an online liberalized current regulator module to calculate the output voltage of the inverter in the two-phase rotating coordinate system at time k includes:

[0154] Step S310: Obtain the parameters of the online self-optimizing current controller of the current control loop at the current moment through the device for online self-optimizing the current regulator parameters;

[0155] Step S320: Calculate the resonant voltage signal based on the parameters of the online self-optimizing controller of the current control loop at the current moment and the difference in the amplitude of the motor stator current;

[0156] Step S330: Obtain the resonant suppression signal based on the capacitor current and the parameters of the online self-optimizing controller of the current control loop at the current moment;

[0157] Step S340: Calculate the output voltage of the inverter in the two-phase rotating coordinate system at time k through the resonant voltage signal and the resonant suppression signal.

[0158] Further, for the resonant voltage signal (u idr (k), u iqr (k)) and the resonant suppression signal (u ids (k), u iqs (k)), its output is the inverter output voltage command (u id * (k), u iq * (k)), and its calculation process is as follows:

[0159]

[0160] Step S400: Obtain the inverter switching state at time k based on the output voltage of the inverter in the two-phase rotating coordinate system at time k and apply it to the switching devices of the three-phase voltage source inverter.

[0161] It should be noted that the method for online self-optimizing the current controller parameters is described with reference to the same or similar parts mentioned above and will not be elaborated here.

[0162] Continue to refer to Figure 3 As shown, in one embodiment, a control device for an LC-filtered permanent magnet synchronous motor is proposed. The device includes:

[0163] A signal acquisition module 3, configured to obtain the inverter-side current, motor stator current, and motor electrical angular velocity information at time k in the three-phase stationary rotating coordinate system;

[0164] A current transformation module 4, configured to calculate the motor stator current and capacitor current in the two-phase rotating coordinate system;

[0165] A motor stator current command generation module 1, configured to obtain the given amplitude of the motor stator current through the motor electrical angular velocity and the given motor speed information at time k;

[0166] The online liberalized current regulation module 2 is used to calculate the output voltage of the inverter in the two-phase rotating coordinate system at time k;

[0167] The three-phase voltage source inverter module 5 is used to obtain the inverter switching state at time k according to the output voltage of the inverter in the two-phase rotating coordinate system at time k and apply it to the three-phase voltage source inverter switching devices.

[0168] Furthermore, the input of the signal acquisition module 3 is the two-phase inverter-side current (i ia , i ib ), the two-phase motor stator current (i sa , i sb ), and the motor rotor position parameter (θ e ), and its output is the three-phase inverter-side current (i iabc ), the three-phase motor stator current (i sabc ), the motor rotor position (θ e ), and the motor electrical angular velocity ω e . The calculation process is as follows:

[0169]

[0170] The current transformation module 4 inputs the three-phase inverter-side current (i iabc ), the three-phase motor stator current (i sabc ), and the motor rotor position (θ e ), and its output is the motor stator current in the two-phase rotating coordinate system (i sd , i sq ), the three-phase LC filter capacitor current (i cd , i cq ). The calculation process is as follows:

[0171]

[0172] The online self-optimizing current regulator module 2 includes a stator current deviation module 201, a stator current deviation differential module 202, an optimal regulation rule module 203, and a controller parameter tuning module 204.

[0173] It should be noted that the device for online self-optimizing current controller parameters is described with reference to the same or similar parts mentioned above and will not be elaborated here.

[0174] Each functional module in the embodiments of the present invention can be integrated into a processing module, or can be physically present as individual units, or two or more units can be integrated into one module. The above integrated module can be implemented in the form of hardware or in the form of a software functional module.

[0175] Continue to refer toFigure 3 As shown, in one embodiment, an LC filter type permanent magnet synchronous motor control system is proposed. The system includes an LC filter type permanent magnet synchronous motor control device and an LC filter module 6.

[0176] The current conversion module 4 outputs currents (i sd , i sq ) and (i cd , i cq ) which are input into the LC filter type permanent magnet synchronous motor control device to generate drive signals for the inverter power switch devices and act on the three-phase inverter module 5. The three-phase inverter module 5 is connected to the permanent magnet synchronous motor 7 through the LC filter 6, enabling the permanent magnet synchronous motor 7 to achieve LC filter type permanent magnet synchronous motor control based on an online self-optimizing current regulator.

[0177] It should be noted that the LC filter type permanent magnet synchronous motor control device is described with reference to the same or similar parts mentioned above and will not be elaborated here.

[0178] Figure 4 A comparison chart of experimental results of a four-pole permanent magnet synchronous motor with a rated power of 15 kW and a rated speed of 1500 r / min using the traditional method and the method proposed in the present invention is given. The chart shows the d-axis and q-axis currents and phase current waveforms of the permanent magnet synchronous motor when the motor suddenly adds a 27 N·m load and suddenly reduces a 27 N·m load (where a corresponds to the experimental waveform of the traditional method and b corresponds to the experimental waveform of the method proposed in the present invention).

[0179] Figure 5 A comparison chart of experimental results of a four-pole permanent magnet synchronous motor with a rated power of 15 kW and a rated speed of 1500 r / min using the traditional method and the method proposed in the present invention is given. The chart shows the d-axis and q-axis currents and phase current waveforms of the permanent magnet synchronous motor when the motor runs from no load at 750 r / min to 1200 r / min and then decelerates to 750 r / min (where a corresponds to the experimental waveform of the traditional method and b corresponds to the experimental waveform of the method proposed in the present invention).

[0180] Figure 4 And Figure 5 It shows that the LC filter type permanent magnet synchronous motor control method based on an online self-optimizing current regulator described in the present invention can ensure the stability of the motor drive system under different working conditions compared with the traditional method, and greatly improve the static and dynamic performance of the motor stator current, realizing the performance improvement of the motor drive system.

[0181] The above are only the preferred embodiments of the present invention and do not impose any form of limitation on the present invention. Although the present invention has been disclosed above in the preferred embodiments, it is not intended to limit the present invention. Any person skilled in the art of this patent, without departing from the scope of the technical solution of the present invention, can make some changes or modifications using the technical content prompted above to obtain equivalent embodiments of equivalent changes. However, as long as it does not depart from the content of the technical solution of the present invention, any simple modification, equivalent change, and modification made to the above embodiments according to the technical essence of the invention still fall within the scope of the present invention's solution.

Claims

1. A method for online self-optimizing the parameters of a current regulator, characterized in that: The method includes: Calculating the difference between the given amplitude of the motor stator current and the amplitude of the motor stator current, and performing differential calculation; Formulating an adjustment rule based on the difference, differential of the motor stator current amplitude, and the optimal bandwidth of the current control loop at the previous moment, and generating an adjustment rule table; Obtaining the self-optimized bandwidth of the current control loop at the current moment by looking up the adjustment rule table; Obtaining the parameters of the online self-optimized current controller of the current control loop according to the self-optimized bandwidth of the current control loop at the current moment.

2. The method for online self-optimizing the parameters of a current controller according to claim 1, characterized in that: The adjustment rule formulation is as follows: When |Δi sd (k)| > E high and When E low < |Δi sd (k)| ≤ E high and When 0 < |Δi sd (k)| ≤ E low and When 0 < |Δi sd (k)| ≤ E low and When |Δi sq (k)| > E high and When E low < |Δi sq (k)| ≤ E high and When 0 < |Δi sq (k)| ≤ E low and When 0 < |Δi sq (k)| ≤ E low and At this time, the optimal bandwidth calculation rule of the current control loop corresponding to the kth moment is as follows: ω bd ψ(k) = ω bd (k - 1) ω bq ψ(k) = ω bq (k - 1) When E low <|Δi sd (k)|≤E high and When 0 <|Δi sd (k)|≤E low and When E low <|Δi sq (k)|≤E high and When 0 <|Δi sq (k)|≤E low and At this time, the calculation rule for the optimal bandwidth of the current control loop corresponding to the k-th moment is as follows: ω bd ψ(k) = 0.998ω bd (k - 1) ω bq ψ(k) = 0.998ω bq (k - 1) When |Δi sd (k)| > E high and When E low < |Δi sd (k)| ≤ E high and When |Δi sq (k)| > E high and When E low < |Δi sq (k)| ≤ E high and When The calculation rule for the optimal bandwidth of the current control loop at the k-th moment is as follows: ω bd (k) = 1.002ω bd (k - 1) ω bq (k) = 1.002ω bq (k - 1) When |Δi sd (k)| > E high and When |Δi sq (k)| > E high and At this time, the calculation rule for the optimal bandwidth of the current control loop corresponding to the k-th moment is as follows: ω bd ψ(k) = 1.004ω bd (k - 1) ω bq ψ(k) = 1.004ω bq (k - 1) Among them, E high represents the upper limit of the absolute value of the difference △i sd of the stator current amplitude of the deviation value motor, and △i sq ; E low represents the lower limit of the absolute value of the difference △i sd of the stator current amplitude of the deviation value motor, and △i sq ; E chigh represents the absolute value upper limit of the differential of the stator current amplitude difference of the motor ; E clow represents the absolute value lower limit of the differential of the stator current amplitude difference of the motor ; ω bd (k) and ω bd (k - 1), ω bq (k) and ω bq (k - 1) are the optimal bandwidths of the d-axis and q-axis current control loops at time k and time k - 1, respectively.

3. A method for online self-optimizing the parameters of a current controller, as claimed in claim 2, wherein: The online self-optimizing controller parameters include k pd (k), k id (k), k pq (k), k iq (k), and its expression is: Among them, T d represents the time constant, T s represents the sampling period, R s represents the motor stator resistance, L f represents the inductance value of the LC filter, C f represents the capacitance value of the LC filter, L d represents the d-axis inductance value of the permanent magnet synchronous motor, L q represents the q-axis inductance value of the permanent magnet synchronous motor.

4. An apparatus for online self-optimizing the parameters of a current regulator, characterized in that: the apparatus includes: A stator current deviation module, configured to calculate the difference in the amplitude of the motor stator current according to the given amplitude of the motor stator current and the amplitude of the motor stator current; A stator current deviation differential module, configured to perform differential calculation on the difference in the amplitude of the motor stator current; An optimal adjustment rule module, configured to formulate an adjustment rule based on the difference, differential of the motor stator current amplitude, and the optimal bandwidth of the current control loop at the previous moment, and generate an adjustment rule table; A controller parameter tuning module, configured to obtain the self-optimized bandwidth of the current control loop at the current moment by looking up the adjustment rule table, and obtain the parameters of the online self-optimized current controller of the current control loop according to the self-optimized bandwidth of the current control loop at the current moment.

5. An online liberalized current regulator module, characterized in that: The module includes: The apparatus for online self-optimizing the parameters of a current regulator according to claim 4, configured to obtain the parameters of the online self-optimized current controller of the current control loop at the current moment; A PI controller module, configured to calculate a resonant voltage signal according to the parameters of the online self-optimized controller of the current control loop at the current moment and the difference in the amplitude of the motor stator current; An active damping parameter self-adjustment module, configured to obtain a resonant suppression signal according to the capacitor current and the parameters of the online self-optimized controller of the current control loop at the current moment; An inverter output voltage command module, configured to calculate the inverter output voltage in the two-phase rotating coordinate system at the k-th moment through the resonant voltage signal and the resonant suppression signal.

6. The LC filter type permanent magnet synchronous motor control method according to claim 5, characterized in that: The vibration voltage signal u idr (k), u iqr (k), and its expression is: where k pd (k), k pq (k), k id (k), k iq (k) represents the parameter of the online self-optimizing current controller of the current control loop at time k, Δi sd (k), Δi sq (k) represents the difference in the amplitude of the motor stator current at time k, and s represents the Laplace operator.

7. The LC filter type permanent magnet synchronous motor control method according to claim 6, characterized in that: The resonant suppression signal u ids (k), u iqs (k), and its expression is: Among them, T s represents the sampling period, R s represents the stator resistance of the motor, L f represents the inductance value of the LC filter, C f represents the capacitance value of the LC filter, L d represents the d-axis inductance value of the permanent magnet synchronous motor, L q represents the q-axis inductance value of the permanent magnet synchronous motor, ω resd represents the d-axis resonance frequency, ω resq represents the q-axis resonance frequency, cos(ω resd T s ) represents the cosine value of the d-axis resonance component, sin(ω resd T s ) represents the sine value of the d-axis resonance component, cos(ω resq T s ) represents the cosine value of the q-axis resonance component, sin(ω resq T s ) represents the sine value of the q-axis resonance component.

8. A control method for a permanent magnet synchronous motor with LC filtering, characterized in that: The method includes: Obtaining the current on the inverter side, the motor stator current, and the motor electrical angular velocity information at the k-th moment in the three-phase stationary rotating coordinate system, and calculating the motor stator current and the capacitor current in the two-phase rotating coordinate system; Obtaining the given amplitude of the motor stator current through the motor electrical angular velocity and the given motor speed information at the k-th moment; Using the online liberalized current regulator module according to claim 5 to calculate the inverter output voltage in the two-phase rotating coordinate system at the k-th moment; Obtaining the inverter switching state at the k-th moment according to the inverter output voltage in the two-phase rotating coordinate system at the k-th moment and applying it to the switching devices of the three-phase voltage source inverter.

9. An LC filter type permanent magnet synchronous motor control device, characterized in that: The apparatus includes: A signal acquisition module, configured to obtain the current on the inverter side, the motor stator current, and the motor electrical angular velocity information at the k-th moment in the three-phase stationary rotating coordinate system; A current transformation module, configured to calculate the motor stator current and the capacitor current in the two-phase rotating coordinate system; A motor stator current command generation module, configured to obtain the given amplitude of the motor stator current through the motor electrical angular velocity and the given motor speed information at the k-th moment; The online liberalized current regulation module described in claim 5 is used to calculate the output voltage of the inverter in the two-phase rotating coordinate system at time k; The three-phase voltage source inverter module is used to obtain the inverter switching state at time k according to the output voltage of the inverter in the two-phase rotating coordinate system at time k and apply it to the three-phase voltage source inverter switching devices.

10. An LC filter type permanent magnet synchronous motor control system, characterized in that, The system includes the LC filter type permanent magnet synchronous motor control device and the LC filter module described in claim 9.

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