Wide-input single-stage Boost-LLC converter and control method

Abstract

The invention discloses a wide-input single-stage Boost-LLC converter and a control method thereof, and relates to the field of power factor correction AC-DC converters. According to the wide-input single-stage Boost-LLC converter, a pulse frequency and pulse width hybrid modulation mode is introduced on the basis of basic pulse frequency modulation control, the voltage stress of a switching device can be reduced, and the input voltage range can be expanded; by introducing quasi-resonance control, the problem that the output of the single-stage AC-DC converter is seriously influenced by double power frequency ripples is solved, and the condition that the output voltage of the secondary side is unstable when the switching frequency is too low is suppressed by adopting a CLC filtering mode at the secondary side. The single-stage Boost-LLC converter has the beneficial effects that through a hybrid modulation mode, the single-stage Boost-LLC converter can realize wide-range input under the condition that the voltage stress requirement of a switching device is not high, double power frequency ripples on the output side are effectively reduced, the stability of the output voltage is ensured, only one-stage power conversion exists in a system, and the system is simple in structure and low in cost. The device has the advantages of small size, high efficiency and low cost.

Description

Technical Field

[0001] The present invention relates to the field of power factor correction AC-DC converters, and is applicable to the occasion of converting alternating current into low-voltage direct current to drive electronic devices, especially more applicable when dealing with a wide-range input power supply. Background Art

[0002] Compared with the traditional two-stage AC-DC converter, the integrated single-stage AC-DC converter has only one-stage structure, only has one power conversion, and shares a part of the switching tubes through single-stage integration, and has the advantages of low cost, small volume, high efficiency, and simple control.

[0003] The Boost PFC converter is widely used in the front stage due to its simple topology, small EMI, easy control of the input current, and simple drive. The LLC resonant converter is adopted in the subsequent DC-DC converter, and its characteristics of zero-voltage turn-on of the switching tube and zero-current turn-off of the output diode within a wide load range can be utilized to achieve a high conversion efficiency.

[0004] However, the gain variation range of the resonant cavity of the LLC resonant converter operating in the traditional pulse frequency modulation is too small to cope with the wide-range input situation, and in the case of a relatively high input voltage, the intermediate DC bus voltage of the Boost-LLC converter is relatively high, which limits the selection of the switching tube.

[0005] Therefore, on the basis of the pulse frequency modulation scheme, the introduction of pulse width modulation with the DC bus voltage as the feedback quantity can greatly slow down the rising speed of the DC bus voltage with the increase of the input line voltage and achieve an expansion of the gain range. However, the double power frequency ripple at the output of the single-stage Boost-LLC converter is significantly larger than that of the traditional two-stage AC-DC converter, and when the input voltage is relatively low, the operating frequency of the converter is far from the resonant frequency, and the secondary side current discontinuous situation is serious, resulting in a large ripple current of the capacitor in the case of single-capacitor filtering. Due to the existence of the internal resistance of the capacitor, the secondary side output result is not ideal. By introducing a quasi-resonant controller into the control loop, the double power frequency ripple interference can be greatly reduced, and the CLC output filter is used on the secondary side, which can significantly improve the output voltage and make the filter inductor not affect the resonant cavity on the primary side. Summary of the Invention

[0006] The purpose of the present invention is to provide a wide-input single-stage Boost-LLC converter and its hybrid control method.

[0007] The technical solution to achieve the purpose of the present invention is as follows:

[0008] The first step: Installation of the wide-input single-stage Boost-LLC converter

[0009] 1.1 The wide-input single-stage Boost-LLC converter mainly consists of a power transmission unit, a signal acquisition unit, a control unit, and an external environment unit. Among them, the power transmission unit includes a rectifier bridge D1(1), an input capacitor C1(2), anti-backflow diodes D2 and D3(3), chopping inductors L1 and L2(4), a full-bridge module(5), a DC bus capacitor C bus (6), a resonant inductor L r (7), a resonant capacitor C r (8), a high-frequency transformer T(9), full-wave rectifier diodes D4 and D5(10), a high-frequency ripple absorption capacitor C2(11), an output filter inductor L3(12), and an output filter capacitor C2(13); the signal acquisition unit includes an input voltage detector(14), a DC bus voltage detector(15), an output voltage detector(16), input inductor current detectors a and b(17), and a resonant inductor current detector(18); the control unit includes an embedded microprocessor(19); the external environment unit includes an AC input source(20) and a load(21);

[0010] 1.2 According to the structural relationship in step 1.1, install and arrange the wide-input single-stage Boost-LLC converter:

[0011] The full-bridge module(5) is composed of fully controlled devices S1~S4; the AC input source(20) is connected to the two input interfaces of the rectifier bridge D1(1), and the positive and negative poles of the input capacitor C1(2) are respectively connected to the positive and negative poles of the rectifier bridge D1(1); the anodes of the anti-backflow diodes D2 and D3(3) are connected to the positive pole of the input capacitor C1(2), and the cathodes are respectively connected to one end of the chopping inductors L1 and L2(4); the other ends of the chopping inductors L1 and L2(4) are connected to the midpoints of the two bridge arms of the full-bridge module(5); the AC side of the full-bridge module(5) is connected to the primary side of the high-frequency transformer T(9) through the resonant inductor L r (7) in series with the resonant capacitor C r (8), the drains of the fully controlled devices S1 and S3 are connected to the positive pole of the DC bus capacitor C bus (6), and the sources of the fully controlled devices S2 and S4 are connected to the negative pole of the DC bus capacitor C bus (6) and the negative pole of the input capacitor C1(2).

[0012] The first terminal of the secondary side of the high-frequency transformer T(9) is connected to the anodes of the full-wave rectifier diodes D4 and D5(10), and the cathodes are connected to the positive electrode of the high-frequency ripple absorption capacitor C2(11). The negative electrode is connected to the end of the first terminal of the secondary side of the high-frequency transformer T(9), the start of the second terminal, the negative electrode of the output filter capacitor C2(13), and the negative electrode of the load(21); One end of the output filter inductor L3(12) is connected to the positive electrode of the high-frequency ripple absorption capacitor C2(11), and the other end is connected to the positive electrode of the output filter capacitor C2(13) and the positive electrode of the load(21).

[0013] Step 2: Drive signal control of the wide-input single-stage Boost-LLC converter

[0014] The embedded microprocessor (19) applies control signals to the switching devices of the full-bridge module. The drive signal of S1 is PWM1; the drive signals of S2 are all PWM2; the drive signals of S3 are all PWM3; the drive signals of S4 are all PWM4.

[0015] PWM1 and PWM2 lead PWM3 and PWM4 by half a cycle. That is, the switching transistors S1 and S2 form the leading leg, and the drive signals are complementary; the switching transistors S3 and S4 form the lagging leg, and the drive signals are complementary. In the case of low input voltage, the duty cycle of the PWM signal of the switching transistor is always 0.5, and the control only changes the switching frequency. When the input voltage is higher than a certain value, the control changes both the switching frequency and the duty cycle at the same time.

[0016] Step 3: Collect the system output voltage V through the signal sampling unit out and the DC bus voltage V bus、 the average value of the resonant inductor current I lr、 the input inductor currents I1 and I2, set the turns ratio of the primary and secondary sides of the high-frequency transformer T(9) to N, and define the reference voltage V ref and the voltage deviation Δv;

[0017] Step 4: The voltage reference value V ref is compared and subtracted from the voltage feedback V out to obtain the voltage deviation Δv. The voltage deviation Δv is input into the embedded microprocessor (19) and calculated through formula (1):

[0018]

[0019] where K up is the voltage loop proportional constant, and K ui is the voltage loop integral constant;

[0020] To suppress the double-frequency power frequency ripple on the output side, the signal of the double-frequency power frequency at the output end is extracted through a quasi-resonant controller, and after amplification processing, it is superimposed on the output of the voltage loop as the current reference value Iref ;

[0021] Feed the current reference value and the average resonant current I lr into Equation (2), and the switching frequency f under double closed-loop control can be calculated as follows s1 :

[0022]

[0023] Since there is a large second-harmonic power frequency ripple in the output voltage of the single-pole AC-DC converter, on the basis of the original PI controller, the voltage deviation Δv is fed into the quasi-resonant controller, and the differential pressure signal caused by the interference brought by the second-harmonic power frequency ripple is amplified and superimposed on the switching frequency output by the original PI controller, so as to obtain the final switching frequency f s 。

[0024] According to the LLC resonant cavity gain curve, it is necessary to ensure that the switching frequency in the control is near the monotonic gain interval, that is, it is necessary to ensure that the switching frequency f s is greater than f smin , and at the same time, since the loss increases when the switching frequency is too high, it is necessary to set the switching frequency f s to be less than f smax 。

[0025] Step 5: When the input voltage is high, the intermediate DC bus voltage is too high, resulting in too high voltage stress requirements for the switching devices. Therefore, duty cycle feedforward control is introduced. The DC bus voltage V bus is fed into the embedded microprocessor (19), and the duty cycle D can be obtained through Equation (3):

[0026]

[0027] where V bus_t is the set bus voltage value for introducing duty cycle feedforward, and x is the proportional feedforward coefficient;

[0028] Step 6: Input the previously obtained switching frequency f s and duty cycle D into the microcontroller (19). According to the driving signal relationship between the switching devices given in Step 2, the PWM driving signals of the duty cycle and phase of the four switching devices in the full-bridge module are generated by the internal timer of the microcontroller (19). According to the actual adjustment effect, repeat Steps 3, 4, and 5 until the output voltage V2 is stabilized to the voltage reference value V ref 。 Description of the Drawings

[0029] Figure 1 is the topological structure diagram of the wide-input single-stage Boost-LLC converter of the present invention.

[0030] Figure 2This is the modal diagram of the wide-input single-stage Boost-LLC converter of the present invention when it is under-resonant during frequency modulation, over-resonant during frequency modulation, and over-resonant during the hybrid modulation of frequency modulation and pulse width modulation.

[0031] Figure 3 This is the implementation flowchart of the wide-input single-stage Boost-LLC converter of the present invention. Specific implementation manners

[0032] The present invention will be further described below with reference to the accompanying drawings.

[0033] Figure 1 This is the topological structure diagram of the wide-input single-stage Boost-LLC converter of the present invention, and it can be used in combination with Figure 3 the implementation flowchart of the wide-input single-stage Boost-LLC converter shown in the figure. By using a single-stage structure, the purpose of reducing switching devices and increasing power density is achieved. By adopting a hybrid modulation method, the purpose of increasing the input range and reducing ripples is realized. By introducing CLC output filtering, the purpose of reducing the output capacitor ripple and stabilizing the output voltage is achieved, and the filtering inductor is prevented from affecting the primary resonant cavity.

[0034] The first step: Installation of the wide-input single-stage Boost-LLC converter

[0035] 1.1 The wide-input single-stage Boost-LLC converter mainly consists of a power transmission unit, a signal acquisition unit, a control unit, and an external environment unit. Among them, the power transmission unit includes a rectifier bridge D1(1), an input capacitor C1(2), anti-backflow diodes D2 and D3(3), chopping inductors L1 and L2(4), a full-bridge module(5), a DC bus capacitor C bus (6), a resonant inductor L r (7), a resonant capacitor C r (8), a high-frequency transformer T(9), full-wave rectifier diodes D4 and D5(10), a high-frequency ripple absorption capacitor C2(11), an output filtering inductor L3(12), an output filtering capacitor C2(13); the signal acquisition unit includes an input voltage detector(14), a DC bus voltage detector(15), an output voltage detector(16), an input inductor current detector a(17), an input inductor current detector b, and a resonant inductor current detector(18); the control unit includes an embedded microprocessor(19); the external environment unit includes an AC input source(20) and a load(21);

[0036] 1.2 According to the structural relationship in step 1.1, install and arrange the wide-input single-stage Boost-LLC converter:

[0037] The full-bridge module (5) is composed of fully-controlled devices S1 to S4; the AC input source (20) is connected to two input interfaces of the rectifier bridge D1 (1), and the positive and negative electrodes of the input capacitor C1 (2) are respectively connected to the positive and negative electrodes of the rectifier bridge D1 (1); the anodes of the anti-backflow diodes D2 and D3 (3) are connected to the positive electrode of the input capacitor C1 (2), and the cathodes are respectively connected to one ends of the chopping inductors L1 and L2 (4); the other ends of the chopping inductors L1 and L2 (4) are connected to the midpoints of two bridge arms of the full-bridge module (5); the AC side of the full-bridge module (5) is connected to the primary side of the high-frequency transformer T (9) through the resonant inductor L r (7) in series with the resonant capacitor C r (8) and then connected to the primary side of the high-frequency transformer T (9). The drains of the fully-controlled devices S1 and S3 are connected to the positive electrode of the DC bus capacitor C bus (6), and the sources of the fully-controlled devices S2 and S4 are connected to the negative electrode of the DC bus capacitor C bus (6) and the negative electrode of the input capacitor C1 (2).

[0038] The first terminal at the primary side of the high-frequency transformer T (9) and the second terminal at the secondary side are respectively connected to the anodes of the full-wave rectifier diodes D4 and D5 (10), the cathodes are connected to the positive electrode of the high-frequency ripple absorption capacitor C2 (11), and its negative electrode is connected to the end of the first terminal, the start of the second terminal, the negative electrode of the output filter capacitor C2 (13) and the negative electrode of the load (21) at the secondary side of the high-frequency transformer T (9); one end of the output filter inductor L3 (12) is connected to the positive electrode of the high-frequency ripple absorption capacitor C2 (11), and the other end is connected to the positive electrode of the output filter capacitor C2 (13) and the positive electrode of the load (21).

[0039] Step 2: Drive signal control of the wide-input single-stage Boost-LLC converter

[0040] Control signals are applied to the switching devices of the full-bridge module through the embedded microprocessor (19). The drive signal of S1 is PWM1; the drive signal of S2 is also PWM2; the drive signal of S3 is also PWM3; the drive signal of S4 is also PWM4.

[0041] PWM1 and PWM2 lead PWM3 and PWM4 by half a cycle. That is, the switching tubes S1 and S2 form the leading bridge arm, and the drive signals are complementary; the switching tubes S3 and S4 form the lagging bridge arm, and the drive signals are complementary. In the case of low-voltage input, the duty cycle of the PWM signal of the switching tube is always 0.5, and the control only changes the switching frequency. When the input voltage is higher than a certain value, the control changes both the switching frequency and the duty cycle at the same time.

[0042] Step 3: Collect the system output voltage V through the signal sampling unit out 、DC bus voltage V bus、 Average value I of the resonant inductor current lr、Input inductor currents I1 and I2, set the turns ratio of the primary and secondary sides of the high-frequency transformer T(9) to N, and define the reference voltage V ref , and the voltage deviation Δv;

[0043] Step 4: The voltage reference value V ref is compared with the voltage feedback V out , and the voltage deviation Δv is obtained by subtraction. The voltage deviation Δv is input into the embedded microprocessor (19) and calculated through formula (1):

[0044]

[0045] where K up is the voltage loop proportional constant, and K ui is the voltage loop integral constant;

[0046] To suppress the double-line frequency ripple on the output side, the signal of the double-line frequency at the output end is extracted by the quasi-resonant controller. After amplification processing, it is superimposed on the output of the voltage loop as the current reference value I ref ;

[0047] The current reference value and the average value of the resonant current I lr are sent into formula (2), and the switching frequency f under the double closed-loop control can be calculated s1 :

[0048]

[0049] Since there is a large double-line frequency ripple in the output voltage of the single-pole AC-DC converter, on the basis of the original PI controller, the voltage deviation Δv is sent into the quasi-resonant controller, and the differential pressure signal caused by the interference of the double-line frequency ripple is amplified and superimposed on the switching frequency output by the original PI controller, so as to obtain the final switching frequency f s .

[0050] According to the LLC resonant cavity gain curve, it is necessary to ensure that the switching frequency in the control is near the monotonic gain interval, that is, it is necessary to ensure that the switching frequency f s is greater than f smin . At the same time, since the loss increases when the switching frequency is too high, it is necessary to set the switching frequency f s to be less than f smax .

[0051] Step 5: When the input voltage is high, the intermediate DC bus voltage is too high, resulting in too high voltage stress requirements for the switching devices. Therefore, duty cycle feedforward control is introduced. The DC bus voltage V bus is sent into the embedded microprocessor (19), and the duty cycle D can be obtained through formula (3):

[0052]

[0053] Among which V bus_t is the bus voltage value for setting the duty cycle feedforward, and x is the proportional feedforward coefficient.

[0054] Step 6: Input the previously obtained switching frequency f s and the duty cycle D into the microcontroller (19). According to the driving signal relationship between the switching devices given in Step 2, use the internal timer of the microcontroller (19) to generate the PWM driving signals of the duty cycle and phase for the four switching devices in the full-bridge module. According to the actual adjustment effect, repeat Step 3, Step 4, and Step 5 until the output voltage V2 is stabilized at the voltage reference value V ref .

Claims

1. A wide-input single-stage Boost-LLC converter and its control method, applicable to the occasion of converting alternating current into low-voltage direct current to drive electronic devices, especially more applicable when dealing with a wide-range input power supply. It includes the following steps: The first step: Installation of the wide-input single-stage Boost-LLC converter 1.1 The wide-input single-stage Boost-LLC converter mainly consists of a power transmission unit, a signal acquisition unit, a control unit, and an external environment unit. Among them, The power transmission unit includes a rectifier bridge D1(1), an input capacitor C1(2), anti-backflow diodes D2 and D3(3), chopping inductors L1 and L2(4), a full-bridge module(5), a DC bus capacitor C bus (6), a resonant inductor L r (7), a resonant capacitor C r (8), a high-frequency transformer T(9), full-wave rectifier diodes D4 and D5(10), a high-frequency ripple absorption capacitor C2(11), an output filter inductor L3(12), an output filter capacitor C2(13); the signal acquisition unit includes an input voltage detector(14), a DC bus voltage detector(15), an output voltage detector(16), an input inductor current detector a(17), an input inductor current detector b, a resonant inductor current detector(18); the control unit includes an embedded microprocessor(19); the external environment unit includes an AC input source(20), a load(21); 1.2 According to the structural relationship in step 1.1, install and arrange the wide-input single-stage Boost-LLC converter: The full-bridge module (5) is composed of fully-controlled devices S1 to S4; the AC input source (20) is connected to two input interfaces of the rectifier bridge D1 (1), and the positive and negative electrodes of the input capacitor C1 (2) are respectively connected to the positive and negative electrodes of the rectifier bridge D1 (1); the anodes of the anti-backflow diodes D2 and D3 (3) are connected to the positive electrode of the input capacitor C1 (2), and the cathodes are respectively connected to one ends of the chopping inductors L1 and L2 (4); the other ends of the chopping inductors L1 and L2 (4) are connected to the midpoints of two bridge arms of the full-bridge module (5); the AC side of the full-bridge module (5) is connected to the primary side of the high-frequency transformer T (9) through the resonance inductor L r (7) in series with the resonance capacitor C r (8) and then connected to the primary side of the high-frequency transformer T (9). The drains of the fully-controlled devices S1 and S3 are connected to the positive electrode of the DC bus capacitor C bus (6), and the sources of the fully-controlled devices S2 and S4 are connected to the negative electrode of the DC bus capacitor C bus (6) and the negative electrode of the input capacitor C1 (2); The first terminal of the secondary side of the high-frequency transformer T(9) is connected to the anodes of the full-wave rectifier diodes D4 and D5(10) respectively, and the cathodes are connected to the positive pole of the high-frequency ripple absorption capacitor C2(11). Its negative pole is connected to the end of the first terminal of the secondary side of the high-frequency transformer T(9), the beginning of the second terminal, the negative pole of the output filter capacitor C2(13), and the negative pole of the load(21); One end of the output filter inductor L3(12) is connected to the positive pole of the high-frequency ripple absorption capacitor C2(11), and the other end is connected to the positive pole of the output filter capacitor C2(13) and the positive pole of the load(21); Step 2: Control of the drive signals of the wide-input single-stage Boost-LLC converter Apply control signals to the switching devices of the full-bridge module through the embedded microprocessor(19). The drive signal of S1 is PWM1; the drive signals of S2 are the same as PWM2; the drive signals of S3 are the same as PWM3; the drive signals of S4 are the same as PWM4; PWM1 and PWM2 lead PWM3 and PWM4 by half a cycle. That is, the switching tubes S1 and S2 form the leading leg, and the drive signals are complementary; the switching tubes S3 and S4 form the lagging leg, and the drive signals are complementary. In the case of low-voltage input, the duty cycle of the PWM signal of the switching tube is always 0.5, and the control only changes the switching frequency. When the input voltage is higher than a certain value, the control changes both the switching frequency and the duty cycle; Step 3: Collect the output voltage V of the system through the signal sampling unit out , the DC bus voltage V bus、 , the average value I of the resonant inductor current lr、 , the input inductor currents I1 and I2, set the turns ratio of the primary and secondary sides of the high-frequency transformer T(9) to be N, and define the reference voltage V ref , and the voltage deviation Δv; Step 4: Voltage reference value V ref is compared with voltage feedback V out to obtain a voltage deviation Δv by subtraction. The voltage deviation Δv is input into the embedded microprocessor (19) and calculated by formula (1) as follows: Where, K up is the proportional constant of the voltage loop, and K ui is the integral constant of the voltage loop; To suppress the double power frequency ripple on the output side, a signal with double power frequency at the output end is extracted by a quasi-resonant controller, and after amplification processing, it is superimposed on the output of the voltage loop as the current reference value I ref ; The current reference value and the average resonant current I lr are fed into Equation (2), and the switching frequency f under double closed-loop control can be calculated s1 as follows: Since there is a large second-order power frequency ripple in the output voltage of the single-pole AC-DC converter, on the basis of the original PI controller, the voltage deviation Δv is sent into the quasi-resonant controller, and the pressure difference signal caused by the interference brought by the second-order power frequency ripple is amplified and superimposed on the switching frequency output by the original PI controller, so as to obtain the final switching frequency f s ; According to the LLC resonant cavity gain curve, it is necessary to ensure that the switching frequency in the control is near the monotonic gain interval, that is, it is necessary to ensure that the switching frequency f s is greater than f smin . At the same time, since the loss increases when the switching frequency is too high, and the LLC resonant cavity gain does not change significantly with frequency, the switching frequency f s is set to be less than f smax ; Step 5: When the input voltage is high, the intermediate DC bus voltage is too high, resulting in too high voltage stress requirements for the switching devices. Therefore, duty cycle feedforward control is introduced; the DC bus voltage V bus is sent into the embedded microprocessor (19), and the duty cycle D can be obtained through Equation (3): Among which, V bus_t is the bus voltage value for setting the duty cycle feedforward, and x is the proportional feedforward coefficient; Step 6: Input the switching frequency f s and the duty cycle D obtained previously into the microcontroller (19). According to the driving signal relationship between the switching devices given in Step 2, use the internal timer of the microcontroller (19) to generate the PWM driving signals of the duty cycle and phase for the four switching devices in the full-bridge module. According to the actual adjustment effect, repeat Step 3, Step 4, and Step 5 until the output voltage V2 is stabilized to the voltage reference value V ref .

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