Dead-beat current prediction control method for dual three-phase electro-magnetic linear synchronous motor

Abstract

The invention relates to a dead-beat current prediction control method for a dual three-phase electro-magnetic linear synchronous motor, and the method comprises the steps: obtaining a stator current sampling value and an electrical angle sampling value of the dual three-phase electro-magnetic linear synchronous motor, and respectively calculating stator currents in a d-q coordinate system and an x-y coordinate system through coordinate system transformation; calculating a stator current one-step predicted value and a stator current disturbance observed value through a current disturbance sliding mode observer; a control voltage vector is calculated through a dead-beat current prediction controller. Compared with the prior art, compared with a traditional PI vector control method, the method has the advantages that the dynamic response speed of a system can be optimized, the current control precision can be improved, current harmonics can be reduced, and motor thrust fluctuation can be inhibited; secondly, a current disturbance sliding-mode observer is designed, disturbance existing in a current control system can be observed, compensation can be conducted on a control voltage vector, and the robustness of the dead-beat current prediction control method is improved.

Description

Technical Field

[0001] The present invention relates to the technical field of synchronous motor control, and in particular to a deadbeat current predictive control method for a dual three-phase electrically excited linear synchronous motor. Background Art

[0002] The long stator linear synchronous motor is an important support and power source for a conventional conductive high-speed maglev train, and its high-performance precise control is the basis for realizing the safe and stable operation of the train. As a multi-phase motor, the dual three-phase electrically excited linear synchronous motor has a smaller single-phase capacity, a higher voltage utilization rate, and a stronger fault-tolerant operation ability compared with the three-phase electrically excited linear synchronous motor. Therefore, it has higher application value for the traction system of high-speed maglev trains. There are few existing methods developed for the traction control of dual three-phase electrically excited linear synchronous motors. When the traditional three-phase linear synchronous motor PI vector control is applied to the traction control of dual three-phase electrically excited linear synchronous motors, the traction force fluctuates greatly, which easily affects the safe and stable operation of high-speed maglev trains. Therefore, it is necessary to develop a higher-performance current control method for dual three-phase electrically excited linear synchronous motors.

[0003] In recent years, current predictive control methods have been widely applied to the control of different motors such as asynchronous motors, permanent magnet synchronous motors, and permanent magnet linear motors due to their characteristics of higher bandwidth and lower current harmonics. The deadbeat predictive control designs a predictive controller based on the motor mathematical model. By sampling the current and position of each phase of the motor and calculating the voltage control vector required in the future, it can achieve precise control of the current and thrust.

[0004] Therefore, a control voltage vector is obtained by using deadbeat predictive control. For example, the invention with the publication number CN117411376A discloses a model predictive current control method for a dual three-phase open winding permanent magnet motor. It adopts the model predictive current control method. By detecting the stator current and position angle, coordinate transformation and discretization are performed, the given current is set to zero, and the deadbeat current prediction principle and value function evaluation are used to optimize the inverter output voltage vector to achieve the suppression of the third, fifth, and seventh harmonics.

[0005] However, the deadbeat current predictive control method is limited by the accuracy of the motor model parameters and is relatively sensitive to current disturbances, resulting in lower robustness of the predictive control strategy. Summary of the Invention

[0006] The purpose of the present invention is to overcome the above-mentioned defects that the deadbeat current predictive control method is limited by the accuracy of the motor model parameters and is relatively sensitive to current disturbances in the prior art, and to provide a deadbeat current predictive control method for a dual three-phase electrically excited linear synchronous motor.

[0007] The purpose of the present invention can be achieved by the following technical solutions:

[0008] A deadbeat current predictive control method for a dual-three-phase electrically excited linear synchronous motor, comprising the following steps:

[0009] Obtain the stator current sampling value and the electrical angle sampling value of the dual-three-phase electrically excited linear synchronous motor, and calculate the stator currents in the d-q coordinate system and the x-y coordinate system respectively through coordinate transformation;

[0010] Calculate the electrical angular velocity sampling value according to the electrical angle sampling value, and combine the stator currents in the d-q coordinate system and the x-y coordinate system and the control voltage vector at the previous moment, and calculate the one-step predicted value of the stator current and the stator current disturbance observation value through a current disturbance sliding mode observer, and the current disturbance sliding mode observer is designed based on the mathematical model of the dual-three-phase electrically excited linear synchronous motor and the discrete exponential reaching law sliding mode control method;

[0011] Use the one-step predicted value of the stator current, the stator current disturbance observation value and the corresponding stator current reference value to calculate the control voltage vector through a deadbeat current predictive controller.

[0012] Further, the calculation process of the stator currents in the d-q coordinate system and the x-y coordinate system includes:

[0013] Taking the α-β coordinate system as an intermediate coordinate system, convert the stator current sampling value in the ABC coordinate system to the α-β coordinate system, and then convert the stator current sampling value in the α-β coordinate system to the stator current in the d-q coordinate system;

[0014] Convert the stator current sampling value in the ABC coordinate system to the stator current in the x-y coordinate system.

[0015] Further, the calculation expression for converting the stator current sampling value in the ABC coordinate system to the α-β coordinate system is:

[0016]

[0017] In the formula, k represents the kth moment, and k + 1 represents the (k + 1)th moment; i A (k), i B (k), i C (k), i D (k), i E (k), i F (k) respectively represent the stator current sampling values of the inverter arm of phase A, phase B, phase C, phase D, phase E and phase F at the kth moment, and i α (k), i β (k) respectively represent the stator currents of the α and β axes at the kth moment;

[0018] The calculation expression for converting the sampled value of the stator current in the α-β coordinate system to the stator current in the d-q coordinate system is as follows:

[0019] i d (k) = i α (k)cosθ + i β (k)sinθ

[0020] i q (k) = -i α (k)sinθ + i β (k)cosθ

[0021] Where θ represents the sampled value of the electrical angle of the dual-three-phase electrically excited linear synchronous motor, and i d (k), i q (k) represent the stator currents of the d and q axes at the k-th moment.

[0022] Furthermore, the calculation expression for converting the sampled value of the stator current in the ABC coordinate system to the stator current in the x-y coordinate system is as follows:

[0023]

[0024] Where k represents the k-th moment, and k + 1 represents the (k + 1)-th moment; i A (k), i B (k), i C (k), i D (k), i E (k), i F (k) respectively represent the sampled values of the stator currents of the inverter arm of phase A, B, C, D, E, and F at the k-th moment, and i x (k), i y (k) represent the stator currents of the x and y axes at the k-th moment.

[0025] Furthermore, the model expression of the current disturbance sliding mode observer is as follows:

[0026]

[0027]

[0028] Where k represents the k-th moment, and k + 1 represents the (k + 1)-th moment; i d (k), i q (k), i x (k), i y (k) respectively represent the stator currents of the d, q, x, and y axes at the k-th moment; respectively represent the predicted values of the stator currents of the d, q, x, and y axes at the k-th moment; ed (k), e q (k), e x (k), e y (k) represents the stator current errors of the d, q, x, and y axes at the k-th moment; respectively represent the predicted values of the stator currents of the d, q, x, and y axes at the (k + 1)-th moment; respectively represent the observed values of the stator current disturbances of the d, q, x, and y axes at the k-th moment; respectively represent the observed values of the stator current disturbances of the d, q, x, and y axes at the (k + 1)-th moment; respectively represent the calculated values of the voltage control vectors of the d, q, x, and y axes at the k-th moment; ω e represents the electrical angular velocity sampling of the motor; T s represents the sampling period, R s represents the stator resistance, L d , L q respectively represent the inductances of the d and q axes, L xy represents the stator leakage inductance, M sm represents the self-inductance of the rotor winding, i fd represents the rotor excitation current; U dsm , U qsm , U xsm , U ysm are respectively the sliding mode control functions of the d, q, x, and y axes; k1, k2, k3, and k4 represent the sliding mode observer parameters.

[0029] Furthermore, the sliding mode control functions U of the d, q, x, and y axes dsm , U qsm , U xsm , U ysm have the following expressions:

[0030] U dsm =-R s e d (k)+L d k1|e d (k)| 1 / 2 tanh(e d (k))

[0031] U qsm =-R s e q (k)+L q k1|e q (k)| 1 / 2 tanh(e q (k))

[0032] U xsm = -R s e x (k) + L xy k2|e x (k)| 1 / 2 tanh(e x (k))

[0033] U ysm = -R s e y (k) + L xy k2|e y (k)| 1 / 2 tanh(e y (k))

[0034] Wherein, tanh is the hyperbolic tangent function.

[0035] Furthermore, the expression of the hyperbolic tangent function is:

[0036]

[0037] Wherein, s represents the function variable.

[0038] Furthermore, the parameters k1, k2, k3, k4 of the sliding mode observer satisfy the following conditions:

[0039]

[0040] Wherein, e d , e q , e x , e y respectively represent the errors of the stator currents on the d, q, x, and y axes at any moment.

[0041] Furthermore, the calculation expression of the deadbeat current predictive controller is:

[0042]

[0043] Wherein, k represents the kth moment, and k + 1 represents the (k + 1)th moment; respectively represent the reference values of the stator currents on the d, q, x, and y axes; respectively represent the predicted values of the stator currents on the d, q, x, and y axes at the (k + 1)th moment; respectively represent the disturbance observation values of the stator currents on the d, q, x, and y axes at the (k + 1)th moment; ω e represents the sampling of the electrical angular velocity of the motor; T s represents the sampling period, R s represents the stator resistance, L d , Lq represent the inductances of the d-axis and q-axis respectively, L xy represent the stator leakage inductance, M sm represent the self-inductance of the rotor winding, i fd represent the rotor excitation current; represent the calculated values of the voltage control vectors for the d, q, x, and y axes at the (k + 1)-th moment respectively.

[0044] Further, the method further includes controlling the operation of the dual three-phase electro-magnetic linear synchronous motor based on the control voltage vector calculated by the deadbeat current predictive controller.

[0045] Compared with the prior art, the present invention has the following advantages:

[0046] (1) Considering that the deadbeat current predictive control method is limited by the accuracy of the motor model parameters and is sensitive to current disturbances, the present invention proposes to design a current disturbance sliding mode observer for the current loop. The current disturbance sliding mode observer is designed based on the mathematical model of the dual three-phase electro-magnetic linear synchronous motor and the discrete exponential reaching law sliding mode control method. The sliding mode control function is added to calculate each stator current prediction value and stator current disturbance observation value, and the sliding mode control function of each axis is calculated according to the stator current error. The stability of the sliding mode observer system is maintained through the limitation of the sliding mode observer parameters; thereby realizing the suppression of current harmonics and thrust fluctuations of the linear motor to improve the robustness of the predictive control strategy.

[0047] (2) Compared with the traditional PI vector control method, the present invention can optimize the system dynamic response speed, improve the current control accuracy, reduce current harmonics and suppress motor thrust fluctuations; secondly, the present invention designs a current disturbance sliding mode observer, which can observe the disturbances existing in the current control system and compensate the control voltage vector to improve the robustness of the deadbeat current predictive control method. BRIEF DESCRIPTION OF THE DRAWINGS

[0048] Figure 1 is a schematic flow chart of a deadbeat current predictive control method for a dual three-phase electro-magnetic linear synchronous motor provided in an embodiment of the present invention;

[0049] Figure 2 is a principle block diagram of a deadbeat current predictive control method for a dual three-phase electro-magnetic linear synchronous motor provided in an embodiment of the present invention;

[0050] Figure 3 is a comparison diagram of the rotor speed waveforms of the method of the present invention and the traditional technical method. Among them, Figure 3 (a) is the rotor speed waveform of the algorithm proposed by the present invention, Figure 3 (b) is the rotor speed waveform of the traditional PI vector control algorithm;

[0051] Figure 4 This is the comparison chart of electromagnetic thrust waveforms between the method of the present invention and the traditional technology method. Among them, Figure 4 (a) in it is the electromagnetic thrust waveform of the algorithm proposed by the present invention, Figure 4 (b) in it is the electromagnetic thrust waveform of the traditional PI vector control algorithm;

[0052] Figure 5 This is the comparison chart of stator current waveforms per phase between the method of the present invention and the traditional technology method. Among them, Figure 5 (a) in it is the stator current waveform per phase of the algorithm proposed by the present invention, Figure 5 (b) in it is the stator current waveform per phase of the traditional PI vector control algorithm;

[0053] Figure 6 This is the comparison chart of stator dq-axis current waveforms between the method of the present invention and the traditional technology method. Among them, Figure 6 (a) in it is the stator dq-axis current waveform of the algorithm proposed by the present invention, Figure 6 (b) in it is the stator dq-axis current waveform of the traditional PI vector control algorithm;

[0054] Figure 7 This is the comparison chart of FFT analysis of thrust fluctuations between the method of the present invention and the traditional technology method. Among them, Figure 7 (a) in it is the FFT analysis of thrust fluctuations of the algorithm proposed by the present invention, Figure 7 (b) in it is the FFT analysis of thrust fluctuations of the traditional PI vector control algorithm. Detailed implementation manners

[0055] To make the objectives, technical solutions and advantages of the embodiments of the present invention clearer, the technical solutions in the embodiments of the present invention will be clearly and completely described below with reference to the accompanying drawings in the embodiments of the present invention. Obviously, the described embodiments are some, but not all, of the embodiments of the present invention. Usually, the components of the embodiments of the present invention described and shown in the drawings here can be arranged and designed in various different configurations.

[0056] Therefore, the following detailed description of the embodiments of the present invention provided in the drawings is not intended to limit the scope of the claimed present invention, but merely represents selected embodiments of the present invention. All other embodiments obtained by those of ordinary skill in the art based on the embodiments of the present invention without creative efforts belong to the scope of protection of the present invention.

[0057] It should be noted that: similar reference numerals and letters denote similar items in the following drawings. Therefore, once an item is defined in one drawing, it does not need to be further defined and explained in subsequent drawings.

[0058] Embodiment 1

[0059] As Figure 1 shown, this embodiment provides a deadbeat current predictive control method for a dual three-phase electrically excited linear synchronous motor, including the following steps:

[0060] S1: Obtain the stator current sampling value and the electrical angle sampling value of the dual three-phase electrically excited linear synchronous motor, and calculate the stator current in the d-q coordinate system and the x-y coordinate system respectively through coordinate system transformation;

[0061] S2: Calculate the electrical angular velocity sampling value according to the electrical angle sampling value, and combine the stator current of the dual three-phase electrically excited linear synchronous motor in the d-q coordinate system and the x-y coordinate system and the control voltage vector at the previous moment, and calculate the one-step predicted value of the stator current and the stator current disturbance observed value through the current disturbance sliding mode observer;

[0062] S3: Use the one-step predicted value of the stator current, the stator current disturbance observed value and the corresponding stator current reference value to calculate the control voltage vector through the deadbeat current predictive controller.

[0063] Step S1 specifically includes the following steps:

[0064] The calculation of the stator current in the d-q coordinate system and the x-y coordinate system is obtained through the vector space decoupling transformation from the ABC coordinate system to the d-q coordinate system and the x-y coordinate system with the α-β coordinate system as the intermediate coordinate system.

[0065] The transformation from the ABC coordinate system to the α-β coordinate system is expressed as:

[0066]

[0067] where k represents the kth moment, and k + 1 represents the (k + 1)th moment; i A (k), i B (k), i C (k), i D (k), i E (k), i F (k) respectively represent the stator current sampling values of the inverter arm of phase A, phase B, phase C, phase D, phase E and phase F at the kth moment, and i α (k), i β (k) respectively represent the stator currents of the α and β axes at the kth moment.

[0068] The transformation from the α-β coordinate system to the d-q coordinate system is the Park transformation, and the transformation formula is:

[0069]

[0070] Among them, θ represents the sampled value of the electrical angle of the dual-three-phase electrically excited linear synchronous motor, and i d (k), i q (k) represent the stator currents of the d and q axes at the k-th moment.

[0071] The transformation from the ABC coordinate system to the x-y coordinate system is:

[0072]

[0073] Among them, i x (k), i y (k) represent the stator currents of the x and y axes at the k-th moment.

[0074] Step S2 specifically includes the following steps:

[0075] The current disturbance sliding mode observer is designed based on the mathematical model of the dual-three-phase electrically excited linear synchronous motor and the discrete exponential reaching law sliding mode control method. The algorithm model of the current disturbance sliding mode observer is as follows:

[0076]

[0077] Among them, k represents the k-th moment, and k + 1 represents the (k + 1)-th moment; i d (k), i q (k), i x (k), i y (k) respectively represent the stator currents of the d, q, x, and y axes at the k-th moment; respectively represent the predicted values of the stator currents of the d, q, x, and y axes at the k-th moment; e d (k), e q (k), e x (k), e y (k) respectively represent the stator current errors of the d, q, x, and y axes at the k-th moment; respectively represent the predicted values of the stator currents of the d, q, x, and y axes at the (k + 1)-th moment; respectively represent the observed values of the stator current disturbances of the d, q, x, and y axes at the k-th moment; respectively represent the observed values of the stator current disturbances of the d, q, x, and y axes at the (k + 1)-th moment; respectively represent the calculated values of the voltage control vectors of the d, q, x, and y axes at the k-th moment; ω e represents the sampled electrical angular velocity of the motor; T s represents the sampling period, R s represents the stator resistance, L d 、L q respectively represent the inductances of the d and q axes, L xyRepresents the stator leakage inductance, M sm Represents the self - inductance of the rotor winding, i fd Represents the rotor excitation current; U dsm 、U qsm 、U xsm 、U ysm Are the sliding - mode control functions of the d, q, x, y axes respectively; k1, k2, k3, k4 represent the parameters of the sliding - mode observer.

[0078] The aforementioned U dsm 、U qsm 、U xsm 、U ysm Are the sliding - mode control functions of the d, q, x, y axes respectively, expressed as:

[0079]

[0080] Where, tanh is the hyperbolic tangent function.

[0081] k represents the k - th moment, k + 1 represents the (k + 1) - th moment; k1, k2, k3, k4 represent the parameters of the sliding - mode observer. To keep the sliding - mode observer system stable, k1, k2, k3, k4 can be selected as parameters satisfying the following conditions:

[0082]

[0083] k3>0, k4>0

[0084] Where, e d 、e q 、e x 、e y Represent the errors of the stator currents of the d, q, x, y axes at any moment respectively.

[0085] The hyperbolic tangent function is expressed as:

[0086]

[0087] Where, s represents the function variable.

[0088] Step S3 specifically includes the following steps:

[0089] Obtain the current reference value, the one - step predicted value of the stator current obtained by the current disturbance sliding - mode observer, and the stator current disturbance observed value, and input them into the dead - beat current prediction controller, which is expressed as:

[0090]

[0091]

[0092] Among them, k represents the k-th moment, and k + 1 represents the (k + 1)-th moment; respectively represent the reference values of the stator currents on the d, q, x, and y axes; respectively represent the calculated values of the voltage control vectors on the d, q, x, and y axes at the (k + 1)-th moment.

[0093] Experimental verification:

[0094] Based on a dual three-phase electric-excited linear synchronous motor with a bus voltage of 5000V and a rated current of 1800A in this embodiment, the effectiveness of the deadbeat current predictive control method for the high-speed maglev dual three-phase electric-excited linear synchronous motor proposed by the present invention is verified.

[0095] Taking the condition that the mover speed accelerates from 0 m / s to 100 m / s and the load is 60 tons as an example, with the inverter switching frequency of 1 KHz, the control of the traditional PI vector control is compared with the deadbeat current predictive control method based on the sliding mode observer proposed by the present invention;

[0096] The comparison diagrams of the mover speed waveforms are as shown in Figure 3 (a) and (b) therein; the comparison diagrams of the electromagnetic thrust waveforms are as shown in Figure 4 (a) and (b) therein; the comparison diagrams of the waveforms of each phase stator current are as shown in Figure 5 (a) and (b) therein; the comparison diagrams of the stator dq-axis currents are as shown in Figure 6 (a) and (b) therein; the comparison diagrams of the FFT analysis of the thrust fluctuation are as shown in Figure 7 (a) and (b) therein.

[0097] It can be seen from the figures that under the control of the two methods, the high-speed maglev dual three-phase electric-excited linear synchronous motor can smoothly accelerate from 0 m / s to 100 m / s. The deadbeat current predictive control method based on the sliding mode observer proposed by the present invention can effectively control the stator currents of each phase and the electromagnetic thrust, with faster dynamic response, lower current harmonics, and significantly reduced thrust fluctuation THD.

[0098] The preferred specific embodiments of the present invention have been described in detail above. It should be understood that those of ordinary skill in the art can make many modifications and variations based on the concept of the present invention without creative labor. Therefore, all technical solutions that can be obtained by those skilled in the art in the technical field of the present invention through logical analysis, reasoning, or limited experiments based on the concept of the present invention on the basis of the prior art should be within the protection scope determined by the claims.

Claims

1. A deadbeat current predictive control method for a dual-three-phase electrically excited linear synchronous motor, characterized in that It includes the following steps: Obtain the stator current sampling value and electrical angle sampling value of the dual-three-phase electrically excited linear synchronous motor, and calculate the stator currents in the d-q coordinate system and x-y coordinate system respectively through coordinate system transformation; Calculate the electrical angular velocity sampling value according to the electrical angle sampling value, and combine the stator currents in the d-q coordinate system and x-y coordinate system and the control voltage vector at the previous moment, and calculate the one-step predicted value of the stator current and the observed value of the stator current disturbance through a current disturbance sliding mode observer, and the current disturbance sliding mode observer is designed based on the mathematical model of the dual-three-phase electrically excited linear synchronous motor and the discrete exponential reaching law sliding mode control method; Use the one-step predicted value of the stator current, the observed value of the stator current disturbance and the corresponding stator current reference value to calculate the control voltage vector through a deadbeat current predictive controller.

2. A deadbeat current predictive control method for a dual-three-phase electrically excited linear synchronous motor according to claim 1, characterized in that The calculation process of the stator currents in the d-q coordinate system and x-y coordinate system includes: Taking the α-β coordinate system as an intermediate coordinate system, convert the stator current sampling value in the ABC coordinate system to the α-β coordinate system, and then convert the stator current sampling value in the α-β coordinate system to the stator current in the d-q coordinate system; Convert the stator current sampling value in the ABC coordinate system to the stator current in the x-y coordinate system.

3. A deadbeat current predictive control method for a dual-three-phase electrically excited linear synchronous motor according to claim 2, characterized in that, The calculation expression for converting the stator current sampling value in the ABC coordinate system to the α-β coordinate system is: where k represents the k-th moment, and k + 1 represents the (k + 1)-th moment; i A (k), i B (k), i C (k), i D (k), i E (k), i F (k) respectively represent the sampled stator current values of the inverter arm of phase A, B, C, D, E, and F at the k-th moment, i α (k), i β (k) respectively represent the stator currents of the α and β axes at the k-th moment; The calculation expression for converting the stator current sampling value in the α-β coordinate system to the stator current in the d-q coordinate system is: i d (k) = i α (k) cosθ + i β (k) sinθ i q (k) = -i α (k)sinθ + i β (k)cosθ Where θ represents the sampled electrical angle value of the dual-three-phase electrically excited linear synchronous motor, and i d (k), i q (k) represent the stator currents of the d and q axes at the k-th moment.

4. A deadbeat current predictive control method for a dual-three-phase electrically excited linear synchronous motor according to claim 2, characterized in that The calculation expression for converting the stator current sampling value in the ABC coordinate system to the stator current in the x-y coordinate system is: where k represents the k-th moment, and k + 1 represents the (k + 1)-th moment; i A (k), i B (k), i C (k), i D (k), i E (k), i F (k) respectively represent the sampled stator current values of the inverter's phase A, B, C, D, E, and F bridge arms at the k-th moment, and i x (k), i y (k) represents the stator current on the x and y axes at the k-th moment.

5. A deadbeat current predictive control method for a dual-three-phase electrically excited linear synchronous motor according to claim 1, characterized in that The model expression of the current disturbance sliding mode observer is: where k represents the k-th moment, and k + 1 represents the (k + 1)-th moment; i d (k), i q (k), i x (k), i y (k) respectively represent the stator currents of the d, q, x, and y axes at the k-th moment; respectively represent the predicted values of the stator currents of the d, q, x, and y axes at the k-th moment; e d (k), e q (k), e x (k), e y (k) respectively represent the stator current errors of the d, q, x, and y axes at the k-th moment; respectively represent the predicted values of the stator currents of the d, q, x, and y axes at the (k + 1)-th moment; respectively represent the observed values of the stator current disturbances of the d, q, x, and y axes at the k-th moment; respectively represent the observed values of the stator current disturbances of the d, q, x, and y axes at the (k + 1)-th moment; respectively represent the calculated values of the voltage control vectors of the d, q, x, and y axes at the k-th moment; ω e represents the sampling of the electrical angular velocity of the motor; T s represents the sampling period, R s represents the stator resistance, L d , L q respectively represent the inductances of the d, q axes, L xy represents the stator leakage inductance, M sm represents the self - inductance of the rotor winding, i fd represents the rotor excitation current; U dsm , U qsm , U xsm , U ysm are respectively the sliding - mode control functions of the d, q, x, and y axes; k1, k2, k3, k4 represent the parameters of the sliding - mode observer.

6. A deadbeat current predictive control method for a dual-three-phase electrically excited linear synchronous motor according to claim 5, characterized in that The sliding mode control functions U of the d, q, x, and y axes dsm , U qsm , U xsm , U ysm are expressed as follows: U dsm = -R s e d (k) + L d k1|e d (k)| 1 / 2 tanh(e d (k)) U qsm = -R s e q (k) + L q k1|e q (k)| 1 / 2 tanh(e q (k)) U xsm = -R s e x (k) + L xy k2|e x (k)| 1 / 2 tanh(e x (k)) U ysm = -R s e y (k) + L xy k2|e y (k)| 1 / 2 tanh(e y (k)) In the formula, tanh is the hyperbolic tangent function.

7. A deadbeat current predictive control method for a dual-three-phase electrically excited linear synchronous motor according to claim 6, characterized in that, The expression of the hyperbolic tangent function is: In the formula, s represents the function variable.

8. A deadbeat current predictive control method for a dual-three-phase electrically excited linear synchronous motor according to claim 5, characterized in that, The parameters k1, k2, k3, k4 of the sliding mode observer satisfy the following conditions: k3>0, k4>0 where, e d , e q , e x , e y respectively represent the errors of the stator currents on the d, q, x, and y axes at any moment.

9. A deadbeat current predictive control method for a dual-three-phase electrically excited linear synchronous motor according to claim 1, characterized in that The calculation expression of the deadbeat current predictive controller is: Where k represents the k-th moment, and k + 1 represents the (k + 1)-th moment; respectively represent the reference values of the stator currents on the d, q, x, and y axes; respectively represent the predicted values of the stator currents on the d, q, x, and y axes at the (k + 1)-th moment; respectively represent the observed values of the stator current disturbances on the d, q, x, and y axes at the (k + 1)-th moment; ω e represents the electrical angular velocity sampling of the motor; T s represents the sampling period, R s represents the stator resistance, L d , L q respectively represent the inductances of the d and q axes, L xy represents the stator leakage inductance, M sm represents the self-inductance of the rotor winding, i fd represents the rotor excitation current; respectively represent the calculated values of the voltage control vectors for the d, q, x, and y axes at the (k + 1)-th moment.

10. A deadbeat current predictive control method for a dual-three-phase electrically excited linear synchronous motor according to claim 1, characterized in that The method further includes controlling the operation of the dual-three-phase electrically excited linear synchronous motor based on the control voltage vector calculated by the deadbeat current predictive controller.

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