FSBB-LLC-based discrete frequency conversion control method

Abstract

The invention relates to the technical field of direct current converters, in particular to a discrete frequency conversion control method based on FSBB-LLC. The FSBB-LLC comprises a Buck bridge arm and a Boost bridge arm of a four-switch Buck-Boost converter, a leading bridge arm and a rear bridge arm of an LLC converter, and a resonant circuit, the Boost bridge arm and the leading bridge arm are multiplexed, and the discrete frequency conversion control method comprises the following steps: a1, determining a switching frequency for a specific output voltage, and calculating a bus voltage Vb and a duty ratio d1 of the Buck bridge arm; a2, setting a plurality of discrete working frequencies near the resonant frequency; a3, calculating a phase shift duty ratio dF according to the given output power and the voltage level; and a4, judging whether the phase-shifting angle exceeds the maximum phase-shifting angle limit or not, if so, returning to execute the step a1, and if not, outputting the control quantity. The wide input and wide output capability of the circuit is guaranteed, and the soft switching capability of the circuit is expanded.

Description

Technical Field

[0001] The present invention relates to the technical field of DC converters, and particularly to a discrete frequency conversion control method based on FSBB-LLC. Background Art

[0002] The wide-gain DC-DC converter is a key power management device in the field of power electronics, which shows excellent adaptability and flexibility in dealing with diverse application scenarios. This converter can maintain efficient and stable output within an extremely wide input voltage range, making it perform well in various power systems, especially those with large voltage fluctuations. With the continuous progress of new energy technologies, distributed energy systems have become increasingly popular, and microgrid systems such as solar photovoltaic systems, wind power generation systems, and electric vehicle charging stations have been comprehensively applied. Due to the complex voltage levels in microgrid systems, there is an urgent need for wide-gain converters to undertake the work of power conversion and achieve seamless connection of various loads and power sources.

[0003] The LLC converter is a high-frequency switching power supply topology widely used in medium and high-power applications, which is favored for its high efficiency and good electromagnetic compatibility. However, under light load conditions, the LLC converter may encounter the problem of gain runaway, which will have a series of adverse effects on the performance and stability of the system. The LLC usually adopts frequency conversion control. When the operating frequency of the LLC deviates too much from the resonant frequency, it is very easy to have gain runaway under light load, resulting in a decrease in the control stability of the device and thus affecting the control accuracy of the output voltage, leading to too high output voltage ripple and causing a series of adverse effects. In addition, gain runaway will cause large fluctuations in the control signal of the device, making the switching devices inside the converter work irregularly, resulting in large thermal stress on the components. Long-term operation will cause the components to overheat, increasing the risk of system failure and shortening the service life of the device.

[0004] Compared with other DC-DC converters, the four-switch Buck-Boost converter has many advantages such as a wide gain range, low device stress, and simple control algorithms. It is a DC-DC converter widely used in the field of power electronics. Currently, it is mainly applied to application scenarios with wide input and output, high power, etc., such as DC microgrids. Moreover, the structure of the four-switch Buck-Boost converter is relatively simple, consisting of only one power inductor and two basic half-bridge structures, which makes the converter have excellent control stability. Since the half-bridge structure is also the basic unit of high-power circuits such as LLC and dual-active bridge circuits, it is easier for the four-switch Buck-Boost converter to be combined with other topologies. Currently, there are already a large number of applications of multiplexing and cascading. For example, the cascading application of the four-switch Buck-Boost converter with the dual-active bridge circuit and LLC not only broadens the wide-gain characteristics of the four-switch Buck-Boost converter but also improves the soft-switching ability of the four-switch Buck-Boost.

[0005] However, the existing DC converters have the following technical drawbacks:

[0006] (1) The control complexity is too high;

[0007] Patent CN202410444874.X designed a hybrid DC converter. The converter integrates a switched-capacitor converter, an LLC resonant converter, and a Buck converter, and its output is the parallel connection of the three converters. This invention realizes high-power output and high voltage drop through parallel connection, but the circuit consists of three converters, which makes its control complexity too high, and the final circuit stability also drops greatly.

[0008] Patent CN202410238861.7 designed a hybrid DC converter that integrates a resonant converter and a Buck converter. The circuit connects the outputs of the resonant converter and the Buck converter in parallel, reducing the output stress of both converters. However, its circuit is too complex, and the control complexity and difficulty are extremely high.

[0009] (2) The gain range is narrow;

[0010] Patent CN202410711356.X is based on the four-switch Buck-Boost and LC resonant circuits to build a DC-DC converter. The circuit connects the LC resonant circuit in series with the inductor of the four-switch Buck-Boost circuit, which can realize the soft switching of all switches. However, the interaction between the resonant circuit and the four-switch Buck-Boost is too large, resulting in a narrow output range of the circuit.

[0011] Patent CN202410220946.2 designed a bidirectional buck-boost DC converter, whose circuit is simple and consists of only four switches, two inductors and three capacitors. The control method is also relatively simple, but its gain range is narrow and it is not suitable for wide input and wide output application scenarios.

[0012] Patent CN202410027369.5 uses three-level technology to improve the basic Buck circuit. The circuit can achieve a relatively high buck ratio and the control algorithm is relatively simple, but the gain range of the circuit is still narrow and it cannot adapt to a relatively wide input voltage range. Summary of the Invention

[0013] The present invention provides a discrete variable frequency control method based on FSBB-LLC, aiming to solve the problems of too high control complexity and narrow gain range of existing DC converters.

[0014] The present invention provides a discrete variable frequency control method based on FSBB-LLC. The FSBB-LLC includes a Buck arm and a Boost arm of a four-switch Buck-Boost converter, a leading arm and a trailing arm of an LLC converter, and a resonant circuit. The Buck arm is composed of switch tubes Q1 and Q2, the leading arm is composed of switch tubes S1, S2, S3, S4 and the resonant circuit, the trailing arm is composed of switch tubes S5, S6, S7, S8, and the Boost arm shares switch tubes S1 and S2 with the leading arm. The resonant circuit includes capacitor C r , inductor L m , inductor L r , and the Buck arm and the Boost arm are connected by inductor L1. The discrete variable frequency control method includes the steps:

[0015] a1. Determine the switching frequency for a specific output voltage, and calculate the bus voltage V b and the duty cycle d1 of the Buck arm;

[0016] a2. Set multiple discrete operating frequencies near the resonant frequency;

[0017] a3. Calculate the phase-shifted duty cycle d F according to the given output power and voltage level;

[0018] a4. Determine whether the phase-shifted angle exceeds the maximum phase-shifted angle limit. If it exceeds, return to step a1 to execute. If it does not exceed, output the control quantity.

[0019] As a further improvement of the present invention, the gain of the multiplexed cascade circuit of the FSBB-LLC is:

[0020]

[0021] Wherein:

[0022]

[0023] In the formula, M: the circuit gain of the multiplexing cascade scheme; M1, M2: the gains of two cascade stages; d1: the duty cycle of the switching tube on the Buck leg; k: the coupling coefficient of the resonant network; L m : magnetically coupled inductor; L r : resonant inductor; Q: the quality factor of the resonant network; C r : resonant capacitor; R eq : equivalent resistance; n: the turns ratio of the transformer; R L : load resistance; f n : resonant frequency; f r : the resonant frequency of the resonant network, N: transformer turns ratio.

[0024] As a further improvement of the present invention, when 0 < d F < d1, the FSBB-LLC is in the optimal operating mode.

[0025] As a further improvement of the present invention, when the FSBB-LLC is in the optimal operating mode, set the parameters and include the following modes:

[0026] Mode 0 [t0~t1]: At time t0, the switching tube Q2 is turned off, and the switching tube Q1 realizes soft switching during the dead time. At this time, the voltage across the inductor L m is negatively clamped,

[0027]

[0028] u Cr (t) = i Lr (t0)Z r sin(ω r (t - t0)) + (V b + nV o ) + [(V b + nV o ) - V Cr (t0)]cos(ω r (t - t0))

[0029]

[0030] Mode 1 [t1~t2]: At time t1, the clamping voltage across L m disappears. During this stage, the capacitor C r , inductor L m , inductor L r all participate in resonance,

[0031]

[0032] Mode 2 [t2~t3]: At time t2, switches S2 and S3 are turned off, and switches S1 and S4 achieve soft switching. During this stage, the inductor L m has its voltage clamped in the positive direction at both ends,

[0033]

[0034] u Cr (t) = i Lr (t2)Z r sin(ω r (t - t2)) + (V b -nV o ) + [(V b -nV o ) - V Cr (t2)]cos(ω r (t - t2))

[0035]

[0036] Mode 3 [t3~t4]: Switch Q1 is turned off, and switch Q2 achieves soft switching,

[0037]

[0038] Mode 4 [t4~t5]: At time t4, the clamping at both ends of the inductor L m disappears. During this stage, capacitors C r , inductor L m , inductor L r all participate in resonance,

[0039]

[0040] Mode 5 [t5~t6]: At time t5, switches S1 and S4 are turned off, and switches S2 and S3 achieve soft switching. The voltage at both ends of the inductor L m is clamped in the negative direction,

[0041] i L1 (t) = i L1 (t5)

[0042]

[0043] u Cr (t) = i Lr (t5)Z r sin(ω r (t - t5)) + (V b +nV o ) + [(Vb +nV o )-V Cr (t5)]cos(ω r (t - t5))

[0044]

[0045] Wherein, V in : input voltage; V o : output voltage; t0 to t6: time points of switch change, see Figure 2 ; ω r : resonance frequency; V b : voltage across capacitor C b ; L m : magnetically coupled inductor; L r : resonance inductor; C r : resonance capacitor; k: coupling coefficient of resonance network; the remaining voltages and currents are represented by the corresponding subscripts for the currents or voltages of the capacitors or inductors.

[0046] As a further improvement of the present invention, when the FSBB-LLC is in the optimal operating mode, the duty cycle d1 is fixed under the condition that the bus voltage and frequency are determined, and the only variable degree of freedom is the phase-shifted duty cycle d F ; According to power conservation, there is:

[0047]

[0048] Wherein, P in : input power; T: time of one cycle; D1: duty cycle of Q1; S a 、S b : integral area of the current during the conduction period of switch Q1;

[0049] The minimum phase-shifted duty cycle for realizing soft switching is obtained as:

[0050]

[0051] Wherein, I ZVS : minimum current for soft switching of the current switch Q1.

[0052] As a further improvement of the present invention, the condition for the phase-shifted duty cycle d F to have a solution is restricted to:

[0053]

[0054] As a further improvement of the present invention, in step S2, when the output voltage is relatively high, the bus voltage is relatively high, and a low operating frequency is adopted. The switching frequency is adjusted to a low frequency point, and the gain of the LLC converter is greater than 1, and the circuit is under-resonant. When the output voltage is relatively low, the bus voltage is relatively low, and a high operating frequency is adopted. The switching frequency is adjusted to a relatively high frequency point, and the gain of the LLC converter is less than 1, and the circuit is over-resonant.

[0055] As a further improvement of the present invention, step a1 specifically includes: setting the resonant parameters of the LLC according to the acceptable bus voltage range and output voltage range, segmenting the output voltage, taking the resonance point as the center, dividing discrete frequency points, and calculating the bus capacitor voltage value so that the bus voltage is positively correlated with the output voltage.

[0056] As a further improvement of the present invention, in step a2, the process of setting multiple discrete operating frequencies is specifically as follows: near the resonant frequency, a gain interval is selected such that the switching frequency is positively correlated with the magnitude of the output voltage to set 3 to 5 discrete operating frequencies.

[0057] The beneficial effects of the present invention are as follows: The present invention is based on a multiplexed cascaded circuit of a four-switch Buck-Boost and an LLC, and improves the circuit control scheme. Compared with the traditional fixed bus + variable frequency control scheme, the present invention guarantees the wide input and wide output capabilities of the circuit, broadens the soft-switching capabilities of the circuit, and also reduces the inductor current ripple of the four-switch Buck-Boost circuit to a certain extent, which is beneficial to reducing the circuit loss and improving the overall efficiency of the converter. Description of the Drawings

[0058] Figure 1 is the circuit topology evolution diagram of FSBB-LLC in the present invention;

[0059] Figure 2 is the waveform diagram of four operating modes of FSBB-LLC in the present invention;

[0060] Figure 3 is the circuit mode diagram of the optimal operating mode of FSBB-LLC in the present invention;

[0061] Figure 4 is the inductor current waveform of the optimal operating mode of FSBB-LLC in the present invention;

[0062] Figure 5 is the inductor current change diagram when the phase shift increases in the present invention;

[0063] Figure 6 is the change trend diagram of the output voltage, frequency, and LLC gain in the present invention;

[0064] Figure 7It is the control flow chart of the discrete variable frequency control method of the present invention;

[0065] Figure 8 It is the FHA gain curve in the present invention. Specific embodiments

[0066] In order to make the objectives, technical solutions and advantages of the present invention clearer, the present invention will be further described in detail below with reference to the accompanying drawings and embodiments.

[0067] In a discrete variable frequency control method based on FSBB-LLC of the present invention, as Figure 1 shown, the topology of FSBB-LLC evolves as follows: The four-switch Buck-Boost converter is composed of two half-bridge circuits, namely the Buck leg and the Boost leg. The full-bridge LLC converter also has two half-bridge circuits, namely the leading leg and the trailing leg. Since the four-switch Buck-Boost converter and the full-bridge LLC have consistent basic units, the Boost leg of the four-switch Buck-Boost is multiplexed with the leading leg of the LLC to save the number of active switches.

[0068] The finally formed FSBB-LLC includes the Buck leg and the Boost leg of the four-switch Buck-Boost converter, the leading leg and the trailing leg of the LLC converter, and a resonant circuit. The Buck leg is composed of switch tubes Q1 and Q2, the leading leg is composed of switch tubes S1, S2, S3, S4 and the resonant circuit, the trailing leg is composed of switch tubes S5, S6, S7, S8, the Boost leg and the leading leg share switch tubes S1 and S2, and the resonant circuit includes capacitor C r , inductor L m , inductor L r , and the Buck leg and the Boost leg are connected by inductor L1.

[0069] Since in the control algorithm of LLC, the duty cycle of the leg is usually fixed at 0.5, which makes the duty cycle of the Boost leg of the four-switch Buck-Boost circuit fixed. Assuming the duty cycle of the switch tubes on the Buck leg is d1, the circuit gain of the multiplexed cascade scheme has changed compared with the simple cascade

[0070]

[0071] Among them:

[0072]

[0073] Among them, M: the circuit gain of the multiplexing cascade scheme; M1, M2: the gains of two cascade stages; d1: the duty cycle of the switching tube on the Buck arm; k: the coupling coefficient of the resonant network; L m : magnetically coupled inductor; L r : resonant inductor; Q: the quality factor of the resonant network; C r : resonant capacitor; R eq : equivalent resistance; n: the turns ratio of the transformer; R L : load resistance; f n : resonant frequency; f r : the resonant frequency of the resonant network; N: transformer turns ratio. These parameters together determine the performance of the LLC resonant converter, including gain, efficiency, and stability, etc.

[0074] The selection process of the optimal operating mode of FSBB-LLC is as follows:

[0075] Since in the control process, the duty cycle of the Boost arm of the four-switch Buck-Boost is fixed at 0.5, the four-switch Buck-Boost circuit has only two control degrees of freedom: the duty cycle d1 of the Buck arm and the phase difference duty cycle d between the two arms F . Therefore, taking the LLC operating in under-resonance as an example, the circuit will have four operating modes, as Figure 2 shown. Figure 2 The time nodes t0 to t6 in are based on the time points for switching the switches, and when the switches are switched is determined by the duty cycle.

[0076] Among the four operating modes, from the perspective of the difficulty of realizing soft switching, in Mode I (0 < d F < d1) and IV (d1 + 0.5 < d F ≤ 1), when the switch Q1 is turned on, the inductor current is still rising (falling), which brings great difficulty to the parameter calculation of soft switching. Moreover, when the switch S2 is turned on, the inductor current is positive, making the soft-switching ability of the switch S2 highly dependent on the subsequent LLC, which also weakens the soft-switching characteristics of the subsequent LLC itself.

[0077] From the perspective of power transmission capacity, the charging slope of the inductor current in Mode I is too low, and its power transmission capacity is too poor compared with other modes.

[0078] From the perspective of circulating current, in Mode II (d1 < d F ≤ 0.5), when the inductor current is relatively large, it enters the commutation stage, which will increase the circuit loss and also limit the output ability of the circuit. Therefore, from many perspectives, the optimal operating mode is Mode III (0.5 < d F ≤ d1 + 0.5), and its circuit mode diagram is as Figure 3(a to f), for simplicity of analysis, the following relevant parameters are set:

[0079]

[0080] Mode 0 [t0 to t1] corresponds to Figure 3 (a): At time t0, switch Q2 is turned off, and switch Q1 achieves soft switching during the dead time. At this time, the voltage across L m is negatively clamped,

[0081]

[0082] where, V in : input voltage; V o : output voltage; Z r : parameter set in formula (2); t0 to t6: time points of switch changes, see Figure 2 ; ω r : resonance frequency; V b : voltage across capacitor C b For the rest of the voltage and current, they represent the current or voltage of the corresponding capacitor or inductor according to the subscript. The same applies to the following formulas.

[0083] Mode 1 [t1 to t2] corresponds to Figure 3 (b): At time t1, the clamping voltage across L m disappears. During this stage, capacitor C r , inductor L m , inductor L r all participate in resonance,

[0084]

[0085] Mode 2 [t2 to t3] corresponds to Figure 3 (c): At time t2, switches S2 and S3 are turned off, and switches S1 and S4 achieve soft switching. During this stage, the voltage across inductor L m is positively clamped,

[0086]

[0087] Mode 3 [t3 to t4] corresponds to Figure 3 (d): Switch Q1 is turned off, and switch Q2 achieves soft switching,

[0088]

[0089] Mode 4 [t4 to t5] corresponds to Figure 3 (e): At time t4, the clamping across inductor L m disappears. During this stage, capacitor C r , inductor L m , inductor L rAll participate in resonance,

[0090]

[0091] Mode 5 [t5~t6] corresponds to Figure 3 (f): At time t5, the switching transistors S1 and S4 are turned off, and the switching transistors S2 and S3 achieve soft switching. The voltage across the inductor L m is negatively clamped at both ends,

[0092]

[0093] Regarding the realization of soft switching in the circuit, Reference 1 can divide the circuit into two parts: a four-switch Buck-Boost and an LLC. Among them, the LLC circuit can achieve soft switching within the working range, so only the soft switching scheme of the four-switch Buck-Boost needs to be designed.

[0094] As shown in Figure 4 is the inductor current waveform. When the bus voltage and frequency are determined, the duty cycle d1 is fixed. Therefore, the only variable degree of freedom is the phase-shifted duty cycle d F . According to the power conservation, there is:

[0095]

[0096] Among them, P in : Input power; T: The time of one cycle; D1: The duty cycle of Q1; S a 、S b : The integral area of the current during the conduction period of the switch Q1.

[0097] The minimum phase-shifted duty cycle to achieve soft switching can be obtained as:

[0098]

[0099] Among them, I ZVS : The minimum current for the current soft switching of the switch Q1.

[0100] However, the phase-shifted angle cannot be increased indefinitely. As shown in Figure 5 is the diagram of the change in inductor current with the increase of phase shift. Since the four-switch Buck-Boost transfers energy to the subsequent stage only when the switching transistor S2 is turned on, the charge increment that the phase shift can bring is limited. Therefore, there must be a maximum power limit. At the same time, in order to modulate the inductor current waveform into the required shape, the phase-shifted duty cycle cannot exceed the duty cycle d1 of the switching transistor S1. Therefore, the condition limit for d F to have a solution is:

[0101]

[0102] Since the LLC circuit is outside the resonant point frequency, the gain will be falsely high during light load operation. Therefore, the control scheme adopts a discrete variable frequency scheme. Near the resonant frequency, according to Figure 8 the FHA gain curve shown, select the gain interval that can make the switching frequency positively correlated with the magnitude of the output voltage to set 3 to 5 discrete operating frequencies. The number of discrete frequency points is selected and adjusted according to the actual situation to avoid the discrete frequency points being too compact or sparse. Through this scheme, the change trend of the bus voltage is consistent with the change trend of the output voltage. When the output voltage is high, the bus voltage is also relatively high, and a low operating frequency is adopted at this time. The low operating frequency corresponds to a high output voltage. At the low operating frequency, the gain of the LLC converter part is greater than 1, and the circuit is under-resonant. In actual operation, the switching frequency needs to be adjusted to a lower frequency point. It can be found from reference formula (10) that the ability of the circuit to achieve soft switching is enhanced. When the output voltage is low, the bus voltage is also relatively low, and a high operating frequency is adopted at this time. The high operating frequency corresponds to a low output voltage. At the high operating frequency, the gain of the LLC converter part is less than 1, and the circuit is over-resonant. In actual operation, the switching frequency needs to be adjusted to a higher frequency point. Under the condition of meeting the circuit soft-switching conditions, the ripple of the inductor current of the four-switch Buck-Boost circuit can be greatly reduced to reduce the loss. The change trend diagrams of the output voltage with parameters such as the operating frequency, LLC gain, and bus voltage are as shown in Figure 6 shown.

[0103] The control focus of the circuit includes three parts. The final control flow chart is as shown in Figure 7 shown. First, determine the switching frequency for a specific output voltage, and then calculate the bus voltage and the duty cycle d1 of the Buck bridge arm. According to the acceptable bus voltage range and output voltage range, design the resonant parameters of the LLC (transformer, resonant capacitor, inductor, etc.). Then, segment the output voltage, with the resonant point as the center, divide discrete frequency points, and calculate the bus capacitor voltage value to ensure that the bus capacitor voltage is positively correlated with the output voltage. After that, calculate the phase-shifted duty cycle d F . Finally, judge whether the phase-shifted angle exceeds the maximum phase-shifted angle limit and output the control quantity.

[0104] The present invention is based on a multiplexed cascade circuit of a four-switch Buck-Boost and an LLC, and improves the control scheme of the circuit. Compared with the traditional fixed bus + variable frequency control scheme, the present invention ensures the wide input and wide output capabilities of the circuit, broadens the soft-switching ability of the circuit, and also reduces the inductor current ripple of the four-switch Buck-Boost circuit to a certain extent, which is beneficial to reducing the loss of the circuit and improving the overall efficiency of the converter.

[0105] The above content is a further detailed description of the present invention in combination with specific preferred embodiments. It cannot be determined that the specific implementation of the present invention is only limited to these descriptions. For those of ordinary skill in the technical field to which the present invention pertains, without departing from the concept of the present invention, several simple deductions or substitutions can still be made, and all should be regarded as belonging to the protection scope of the present invention.

Claims

1. A discrete variable frequency control method based on FSBB-LLC, characterized in that, The FSBB-LLC includes the Buck arm and the Boost arm of a four-switch Buck-Boost converter, the leading arm and the trailing arm of an LLC converter, and a resonant circuit. The Buck arm is composed of switch Q1 and switch Q2. The leading arm is composed of switches S1, S2, S3, S4, and the resonant circuit. The trailing arm is composed of switches S5, S6, S7, S8. The Boost arm shares switches S1 and S2 with the leading arm. The resonant circuit includes capacitor C r , inductor L m , inductor L r . The Buck arm and the Boost arm are connected by inductor L1. The discrete variable-frequency control method includes the steps: a1. Determine the switching frequency for a specific output voltage and calculate the bus voltage V b and the duty cycle d1 of the Buck leg; a2. Set multiple discrete operating frequencies near the resonant frequency; a3. Calculate the phase-shifting duty cycle d based on the given output power and voltage level F ; a4. Determine whether the phase shift angle exceeds the maximum phase shift angle limit. If it exceeds, return to step a1 for execution. If it does not exceed, output the control quantity.

2. The discrete variable frequency control method based on FSBB-LLC according to claim 1, characterized in that The gain of the multiplexed cascaded circuit of the FSBB-LLC is: Where: Where, M: Circuit gain of the multiplexing cascade scheme; M1, M2: Gains of two cascade stages; d1: Duty cycle of the switching transistor on the Buck leg; k: Coupling coefficient of the resonant network; L m : Magnetically coupled inductor; L r : Resonant inductor; Q: Quality factor of the resonant network; C r : Resonant capacitor; R eq : Equivalent resistance; n: Turns ratio of the transformer; R L : Load resistance; f n : Resonant frequency; f r : Resonant frequency of the resonant network, N: Transformer turns ratio.

3. The discrete variable frequency control method based on FSBB-LLC according to claim 1, wherein When 0 < d F <d1, the FSBB-LLC is in the optimal operating mode.

4. The discrete variable frequency control method based on FSBB-LLC according to claim 3, wherein When the FSBB-LLC is in the optimal working mode, set the parameters and include the following modes: Mode 0 [t0~t1]: At time t0, switch Q2 is turned off, and switch Q1 achieves soft switching within the dead time. At this time, the voltage across inductor L m is negatively clamped at both ends. u Cr (t) = i Lr (t0)Z r sin(ω r (t - t0)) + (V b + nV o ) + [(V b + nV o ) - V Cr (t0)]cos(ω r (t - t0)) Mode 1 [t1~t2]: At time t1, L m The clamping voltages at both ends disappear. During this stage, capacitor C r , inductor L m , inductor L r all participate in resonance. Mode 2 [t2 - t3]: At time t2, switches S2 and S3 are turned off, and switches S1 and S4 achieve soft switching. During this stage, the voltage across inductor L m is positively clamped at both ends. u Cr (t) = i Lr (t2)Z r sin(ω r (t - t2)) + (V b -nV o ) + [(V b -nV o ) - V Cr (t2)]cos(ω r (t - t2)) Mode 3 [t3~t4]: Switch Q1 is turned off, and switch Q2 realizes soft switching. Mode 4 [t4~t5]: At time t4, the clamping at both ends of inductor L m disappears. During this stage, capacitor C r , inductor L m , inductor L r all participate in resonance. Mode 5 [t5 - t6]: At time t5, switches S1 and S4 are turned off, and switches S2 and S3 achieve soft switching. The voltage across inductor L m is negatively clamped at both ends. i L1 i(t) = L1 (t5) u Cr (t) = i Lr (t5)Z r sin(ω r (t - t5)) + (V b + nV o ) + [(V b + nV o ) - V Cr (t5)]cos(ω r (t - t5)) Where, V in : input voltage; V o : output voltage; t0 to t6: time points of switch changes, see Figure 2; ω r : resonance frequency; V b : voltage across capacitor C b ; L m : magnetically coupled inductor; L r : resonance inductor; C r : resonance capacitor; k: coupling coefficient of the resonance network; the remaining voltages and currents are represented according to the subscripts as the currents or voltages of the corresponding capacitors or inductors.

5. The discrete variable frequency control method based on FSBB-LLC according to claim 3, wherein When the FSBB-LLC is in the optimal working mode, the duty cycle d1 is fixed under the condition that the bus voltage and frequency are determined, and the only variable degree of freedom is the phase-shifted duty cycle d F ; According to the power conservation, we have: Among them, P in : input power; T: time of one cycle; D1: duty cycle of Q1; S a , S b : integral area of current during the conduction period of switch Q1; The minimum phase shift duty cycle for realizing soft switching is obtained as: Among them, I ZVS : The minimum current of the soft switch of the current switch Q1.

6. The discrete variable frequency control method based on FSBB-LLC according to claim 5, wherein Phase-shifted duty cycle d F The condition for having a solution is restricted to:

7. The discrete variable frequency control method based on FSBB-LLC according to claim 1, characterized in that In step S2, when the output voltage is relatively high, the bus voltage is relatively high. A low operating frequency is adopted, and the switching frequency is adjusted to a low frequency point. The gain of the LLC converter is greater than 1, and the circuit is under-resonant. When the output voltage is relatively low, the bus voltage is relatively low. A high operating frequency is adopted, and the switching frequency is adjusted to a relatively high frequency point. The gain of the LLC converter is less than 1, and the circuit is over-resonant.

8. The discrete variable frequency control method based on FSBB-LLC according to claim 1, characterized in that, Step a1 specifically includes: According to the acceptable bus voltage range and output voltage range, set the resonant parameters of the LLC, segment the output voltage, divide discrete frequency points centered on the resonant point, and calculate the bus capacitor voltage value to make the bus voltage positively correlated with the output voltage.

9. The discrete variable frequency control method based on FSBB-LLC according to claim 1, characterized in that In step a2, the process of setting multiple discrete operating frequencies is specifically: Near the resonant frequency, select a gain interval where the switching frequency is positively correlated with the magnitude of the output voltage to set 3 to 5 discrete operating frequencies.

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