Current resonant DC / DC converter

Abstract

A single-phase current resonant DC / DC converter (1) is provided with a main circuit unit (10) provided with a transformer circuit (Tr), a primary-side switching circuit (11), a primary-side resonant circuit (12), and a secondary-side rectifying circuit (13), and a control unit (20) that performs frequency modulation control, output suppression conversion control for reducing the output, and output rise conversion control for increasing the output. The output suppression conversion control is performed on the basis of a first control amount calculated using the difference between the first frequency and the drive frequency, and the output rise conversion control is performed on the basis of a second control amount calculated using the difference between the second frequency and the drive frequency.

Description

Technical Field

[0001] The present invention relates to a current resonance type DC / DC converter. The current resonance type DC / DC converter includes a single-phase current resonance type DC / DC converter and a three-phase current resonance type DC / DC converter. Background Art

[0002] In recent years, LLC and CLLC (or CLLLC) current resonant DC / DC converters have attracted considerable attention. These converters offer advantages such as compactness, high efficiency, a reduced number of components, and the potential for cost reduction. Consequently, they are used in a variety of applications. For example, they are used as power sources for fast chargers that charge electric vehicle batteries and for bidirectional V2H (Vehicle to Home) systems that charge and discharge electric vehicle batteries.

[0003] LLC and CLLC current resonant DC / DC converters use frequency modulation to control output, but this results in a narrow output voltage range. Frequency modulation requires increasing the drive frequency to reduce the output voltage, but increasing the drive frequency increases losses. Furthermore, even increasing the drive frequency can prevent the output from being reduced to zero. Furthermore, frequency modulation cannot fully boost the input voltage when it is low, preventing the desired output voltage from being achieved.

[0004] For example, the CHAdeMO standard for electric vehicle charging requires support for a wide output voltage range from 150 V to 450 V. Therefore, existing current resonant DC / DC converters used in fast chargers perform intermittent control (burst control) when reducing output voltage in addition to frequency modulation control. However, intermittent control increases output current ripple.

[0005] Patent Document 1 proposes a method for controlling a wide range of output voltages by switching between frequency modulation control and phase shift control. This method reduces the output by increasing the drive frequency to its maximum through frequency modulation control, and then initiates phase shift control to further reduce the output. However, this method requires separate frequency modulation control and phase shift control. Since the control amount of the frequency modulation control is unrelated to the control amount of the phase shift control, this leads to issues of increased control complexity and software structure.

[0006] Patent Document 2 proposes a scheme that switches between frequency modulation control and phase shift control, similar to Patent Document 1. However, by performing phase shift control simultaneously with frequency modulation control within a predetermined switching range, the resulting output fluctuations are suppressed. However, this method also switches between two control modes, and thus, in addition to the same issues as the method described in Patent Document 1, it also faces the difficulty of simultaneously controlling two controlled variables.

[0007] Furthermore, in V2H systems requiring bidirectional operation, if the battery voltage is low during discharge operation, the desired output voltage cannot be achieved. Therefore, a bidirectional buck-boost converter is combined with a bidirectional CLLC current resonant converter to perform buck-boost operation. This results in reduced efficiency, an increased number of components, and an increase in power supply size.

[0008] Patent Document 3 proposes a current resonant DC / DC converter that incorporates a boost circuit within an LLC resonant circuit. However, this configuration requires separate frequency modulation control and boost control. This not only complicates the software architecture due to switching between the two control modes, but also increases the number of components by an amount corresponding to the boost switching element.

[0009] Patent Document 4 proposes the following method: In a current resonant circuit with a full-bridge drive circuit, the drive circuit is controlled so that there is a phase difference between the turn-off of the switching element in the upper arm of the first leg and the turn-off of the switching element in the lower arm of the second leg, such that the phase shift increases with increasing drive frequency. In this method, when the input-output voltage ratio is large and the drive frequency is below the resonant frequency, the phase shift is fixed to a minimum value to perform a step-up operation. When the input-output voltage ratio decreases, and the drive frequency is between the resonant frequency and the maximum drive frequency, the phase shift increases as the drive frequency increases. When the drive frequency reaches the maximum drive frequency, the drive frequency is fixed and only the phase shift increases to perform a step-down operation.

[0010] However, the method described in Patent Document 4 performs phase shift control when the drive frequency is above the resonant frequency. This can sometimes cut off large resonant currents, preventing soft switching and resulting in reduced efficiency. Furthermore, when the input-to-output voltage ratio is large, boosting the voltage by setting the drive frequency below the resonant frequency also presents challenges, such as insufficient boosting depending on the input-to-output conditions and inability to support a wide range of output voltages.

[0011] Patent Document 5 proposes a control method for an LLC-based current resonant DC / DC converter. This control method includes the following steps: calculating a theoretical frequency corresponding to frequency modulation control based on an input signal; performing frequency modulation control by fixing a phase shift amount when the theoretical frequency is within a first frequency range; and performing phase shift control by fixing the drive frequency to a maximum frequency when the theoretical frequency is within a second frequency range. However, in LLC-based current resonant DC / DC converters, the optimal drive frequency varies depending on conditions such as input and output voltages and the load at the power supply destination. Therefore, calculating the theoretical frequency is practically difficult.

[0012] In recent years, three-phase current resonance DC / DC converters using the CLLC / LLC method have been adopted as DC / DC converters and bidirectional DC / DC converters due to demands for miniaturization, high efficiency, low cost, and large capacity.

[0013] In a three-phase current resonant DC / DC converter, a single-phase current resonant DC / DC converter with a half-bridge or full-bridge CLLC / LLC structure is made into a three-phase drive, which is characterized by the ability to achieve high power and low ripple output based on the self-balancing function. However, the three-phase current resonant DC / DC converter has the same problem of narrow output voltage range as the single-phase current resonant DC / DC converter with a half-bridge or full-bridge CLLC / LLC structure. In particular, in the case of the CLLC method for bidirectional power transmission, for step-down operation, by carefully designing control such as burst (intermittent) control and phase shift control, it is possible to reduce power without additional circuits. However, for step-up operation, additional circuits are required, which poses many problems.

[0014] Patent Document 6 discloses a CLLC-type three-phase current resonant DC / DC converter in which a boost circuit is connected in series. However, when a boost circuit is connected in series to a three-phase current resonant DC / DC converter, the multiple switching elements and boost coils that constitute the boost circuit increase the number of components, leading to higher costs, an increase in the size of the entire device (or system), and a decrease in power conversion efficiency due to switching losses in the switching elements and iron and copper losses in the boost coils.

[0015] Patent Document 7 discloses a voltage-current type DC / DC converter with a single-phase full-bridge structure. This converter incorporates an LC resonant circuit in series with the secondary winding of a transformer. The converter performs boost operation by short-circuiting the secondary switching circuit. In the single-phase voltage-current type DC / DC converter described in Patent Document 7, the duration of the secondary switching circuit short-circuiting (the on-duty cycle of the short-circuiting switch) is controlled based on the transformer's winding ratio and output voltage, enabling boost operation without requiring a boost circuit. However, this control (boost control) cannot be directly applied to a current resonant type DC / DC converter. Current resonant type DC / DC converters require frequency modulation control, which controls the driving frequency of the switching elements (because the output varies with the driving frequency). For example, consideration must be given to controlling the driving frequency during boost control. Furthermore, in a three-phase current resonant type DC / DC converter, two resonant current paths exist on the driving side. Even when the switching elements of the primary switching circuit are turned on, the resonant current does not flow continuously, making control more complex. Therefore, when performing boost control using a three-phase current resonance type DC / DC converter, for example, at what timing and through which path the secondary-side switching circuit is short-circuited becomes a problem.

[0016] Patent Document 8 discloses a single-phase full-bridge DC / DC converter that performs a boost operation by short-circuiting the primary-side switching circuit when transferring power from the secondary side to the primary side. Because the DC / DC converter described in Patent Document 8 has a voltage-type circuit structure on the secondary side, the aforementioned control (boost control) cannot be directly applied to a current resonant DC / DC converter, as with the voltage-current type DC / DC converter described in Patent Document 7. In particular, in the case of a three-phase current resonant DC / DC converter, as described above, the timing and path for performing the boost operation become problematic.

[0017] Patent Document 9 discloses a converter that detects the critical current flowing through the secondary winding of a transformer to turn on the switching elements of a primary-side switching circuit (a half-bridge structure). In sync with this turn-on timing, the switching elements of a secondary-side switching circuit (a bridge rectifier circuit) are turned on for a predetermined time, thereby short-circuiting the secondary-side switching circuit. In the converter described in Patent Document 9, the secondary-side switching circuit performs a boost operation during the predetermined time through this control, thereby increasing the output voltage. However, when this control is applied to a current resonant DC / DC converter, the switching cycle of the primary-side switching circuit is shortened and the driving frequency of the primary-side switching circuit is increased. As a result, in a current resonant DC / DC converter, the output voltage increase caused by the boost operation and the output voltage decrease caused by the increased driving frequency occur simultaneously. Therefore, even if the secondary-side switching circuit is short-circuited for the same period of time, the output voltage increase dominates at relatively low driving frequencies close to the resonant frequency, resulting in a higher output. However, at relatively high driving frequencies far from the resonant frequency, the output voltage decrease dominates, resulting in a lower output. As described above, when the control described in Patent Document 9 is applied to a current resonant DC / DC converter, the control characteristics change as the drive frequency changes. Furthermore, when applied to a three-phase current resonant DC / DC converter, in addition to the aforementioned issues, there are also the questions of when and through which path the secondary-side switching circuit should be short-circuited.

[0018] Prior art literature

[0019] Patent Literature

[0020] Patent Document 1: Japanese Patent No. 6898511

[0021] Patent Document 2: International Publication No. 2022 / 153723

[0022] Patent Document 3: U.S. Patent Application Publication No. 2015 / 0162840

[0023] Patent Document 4: Japanese Patent Application Laid-Open No. 2017-99182

[0024] Patent Document 5: Chinese Patent No. 114785150

[0025] Patent Document 6: Chinese Patent No. 114157159

[0026] Patent Document 7: Japanese Patent Application Laid-Open No. 2005-224012

[0027] Patent Document 8: Japanese Patent Application Laid-Open No. 2014-180167

[0028] Patent Document 9: Japanese Patent Application Laid-Open No. 2021-112003 Summary of the Invention

[0029] Problems to be solved by the invention

[0030] The present invention has been made in view of the above circumstances, and an object of the present invention is to provide a current resonance type DC / DC converter that can support a wide range of output voltages without adding components and circuits or complicating the control software structure.

[0031] Technical solutions to solve problems

[0032] In order to solve the above problems, the single-phase current resonance type DC / DC converter of the present invention includes: a main circuit unit and a control unit,

[0033] The main circuit unit comprises:

[0034] Transformer circuit;

[0035] a primary-side switching circuit, disposed on the primary side of the transformer circuit, comprising at least one primary-side branch, wherein the primary-side branch comprises a group of primary-side switching elements connected in series;

[0036] a primary-side resonant circuit, disposed between the primary-side branch and the transformer circuit, comprising a resonant coil and a resonant capacitor; and

[0037] The secondary-side rectifier circuit is provided on the secondary side of the transformer circuit and includes at least one secondary-side branch. The secondary-side branch includes a set of rectifier units connected in series. The rectifier units include diodes or secondary-side switching elements.

[0038] The control unit controls the primary-side switching element and / or the secondary-side switching element.

[0039] The single-phase current resonance type DC / DC converter is characterized in that:

[0040] The control unit performs:

[0041] Frequency modulation control, when the driving frequency of the primary-side switching element is between a first frequency and a second frequency lower than the first frequency, controlling the output of the main circuit portion using the driving frequency as a control variable; and

[0042] Control of at least one of output suppression conversion control and output increase conversion control, wherein the output suppression conversion control is a control in which, when the driving frequency exceeds the first frequency, the output is reduced by using a first control amount calculated using the difference between the driving frequency and the first frequency as the control amount, and the output increase conversion control is a control in which, when the driving frequency is lower than the second frequency, the output is increased by using a second control amount calculated using the difference between the second frequency and the driving frequency as the control amount.

[0043] In this configuration, frequency modulation control, output suppression switching control based on the first controlled variable, and output increase switching control based on the second controlled variable all employ frequency control in which the drive frequency is included in the controlled variable. Specifically, this configuration eliminates the need to switch to another control mode for control, nor does it require countermeasures for output fluctuations associated with switching control modes; control can be performed solely based on frequency control. Consequently, this configuration supports a wide range of output voltages without adding additional components or circuits, or complicating the control software architecture.

[0044] In the single-phase current resonance type DC / DC converter,

[0045] The output suppression conversion control can be configured to include:

[0046] Frequency PWM conversion control, based on the first control amount, controlling the pulse width of the PWM signal for the primary side switching element or the secondary side switching element, or

[0047] Frequency intermittent conversion control, based on the first control amount, controls the standby period during which the primary side switching element is not turned on or off, or

[0048] The frequency phase shift conversion control controls the phase shift amount between the primary-side branches based on the first control amount.

[0049] In the single-phase current resonance type DC / DC converter, for example,

[0050] The output rising conversion control is a frequency boosting conversion control in which, based on the second control amount, the secondary-side rectifier circuit is short-circuited by controlling the conduction period of the secondary-side switching element of the secondary-side rectifier circuit while the resonant current flows in the primary-side resonant circuit.

[0051] The single-phase current resonance type DC / DC converter can be configured as follows:

[0052] The first control amount is calculated by calculating the difference between the driving frequency and the first frequency and a predetermined first gain.

[0053] The control unit during the output suppression conversion control makes the value of the first frequency and / or the first gain variable according to input / output conditions related to the input voltage, input / output voltage ratio, or output power of the main circuit unit.

[0054] Furthermore, the single-phase current resonance type DC / DC converter can be configured as follows:

[0055] The second control amount is calculated by calculating the difference between the second frequency and the driving frequency and a predetermined second gain.

[0056] The control unit during the output step-up conversion control makes the value of the second frequency and / or the second gain variable according to input / output conditions related to the input voltage, input-output voltage ratio, or output power of the main circuit unit.

[0057] The single-phase current resonance type DC / DC converter can be configured as follows:

[0058] The control unit during the output suppression conversion control performs control so that the first control amount does not exceed a predetermined first maximum control amount.

[0059] The control unit during the output increase conversion control performs control so that the second control amount does not exceed a predetermined second maximum control amount.

[0060] The single-phase current resonance type DC / DC converter can be configured as follows:

[0061] The main circuit unit includes a resonant coil and a resonant capacitor.

[0062] The primary-side switching circuit includes two primary-side branches consisting of a first branch and a second branch connected in parallel.

[0063] The secondary-side rectifier circuit includes two secondary-side branches consisting of a third branch and a fourth branch connected in parallel, and the rectifier unit of each branch is composed of the secondary-side switching element, or only the rectifier units constituting the upper and lower branches of the third branch or the fourth branch are composed of the secondary-side switching element, or only the rectifier units constituting the lower branch of the third branch and the upper branch or the lower branch of the fourth branch are composed of the secondary-side switching element, and a reverse-connected diode is connected in parallel on the current path of the secondary-side switching element.

[0064] The single-phase current resonance type DC / DC converter can be configured as follows:

[0065] The control unit performs frequency phase shift conversion control as the output suppression conversion control,

[0066] During the frequency phase shift conversion control, the control unit determines the driving frequency by comparing the output value of the output with the target value, and uses the value obtained by multiplying the difference between the driving frequency and the first frequency by a predetermined first gain as the phase shift amount, so that there is a phase difference corresponding to the phase shift amount between the first branch and the second branch.

[0067] The single-phase current resonance type DC / DC converter can be configured as follows:

[0068] The control unit performs frequency boost conversion control as the output increase conversion control,

[0069] During the frequency boost conversion control, the control unit determines the driving frequency by comparing the output value of the output with a target value, and uses the value obtained by multiplying the difference between the second frequency and the driving frequency by a predetermined second gain as the boost amount, so that the secondary-side switching element has a conduction period corresponding to the boost amount.

[0070] The single-phase current resonance type DC / DC converter can be configured as follows:

[0071] The main circuit unit includes a resonant coil and a resonant capacitor.

[0072] The primary-side switching circuit includes two primary-side branches consisting of a first branch and a second branch connected in parallel.

[0073] The secondary-side rectifier circuit includes two secondary-side branches consisting of a third branch and a fourth branch connected in parallel, and the rectifier unit of each branch is composed of the secondary-side switching element.

[0074] The control unit performs frequency PWM conversion control as the output suppression conversion control,

[0075] During the frequency PWM conversion control, the control unit determines the driving frequency by comparing the output value of the output with the target value, and uses the difference between the driving frequency and the first frequency to calculate the phase difference as the first control quantity, so that the conduction timing of the secondary side switching element is staggered relative to the conduction timing of the primary side switching element by an amount corresponding to the phase difference.

[0076] The three-phase current resonance type DC / DC converter of the present invention comprises:

[0077] a transformer unit including a first transformer circuit, a second transformer circuit, and a third transformer circuit, each transformer circuit including a primary-side coil and a secondary-side coil;

[0078] a primary-side switching circuit comprising a first branch, a second branch, and a third branch connected in parallel, each branch comprising an upper branch and a lower branch connected in series, each branch comprising a switching element, a reverse-connected diode, and a portion of a resonant capacitor connected in parallel;

[0079] a primary-side resonant circuit comprising a first resonant circuit, a second resonant circuit, and a third resonant circuit, wherein the first resonant circuit is connected to the first branch and the primary-side coil of the first transformer circuit, the second resonant circuit is connected to the second branch and the primary-side coil of the second transformer circuit, and the third resonant circuit is connected to the third branch and the primary-side coil of the third transformer circuit, each resonant circuit comprising a resonant coil and a resonant capacitor;

[0080] a secondary-side switching circuit comprising a fourth branch, a fifth branch, and a sixth branch connected in parallel, each branch comprising a switching element, a reverse-connected diode, and a portion of a resonant capacitor connected in parallel; and

[0081] a control unit that controls the primary-side switching circuit and the secondary-side switching circuit,

[0082] The three-phase current resonance type DC / DC converter performs forward power transmission from the primary side switching circuit to the secondary side switching circuit, and is characterized in that:

[0083] When the driving frequency of the primary-side switching circuit is higher than a first frequency, the control unit performs frequency modulation control to control the driving frequency according to the output of the secondary-side switching circuit.

[0084] The control unit performs boost control for generating a boost period in which the secondary-side switching circuit is short-circuited when the driving frequency is equal to or lower than the first frequency.

[0085] During the boost control, the control unit turns on any of the switching elements in the secondary-side switching circuit to generate the boost period during a period in which the resonant current flows through the first resonant circuit and the second resonant circuit, a period in which the resonant current flows through the second resonant circuit and the third resonant circuit, and a period in which the resonant current flows through the third resonant circuit and the first resonant circuit.

[0086] In this configuration, boost control is performed when the drive frequency is below a predetermined first frequency. This allows a large resonant current to flow through the resonant circuit during the boost period, quickly accumulating a large amount of energy in the resonant coil. This energy is then released as load current, enabling a high output voltage to be achieved even with low input voltages. This configuration supports a wide range of output voltages without requiring additional components or circuitry for boosting the voltage.

[0087] The three-phase current resonance type DC / DC converter can be configured as follows:

[0088] In the transformer unit,

[0089] The secondary side coil of the first transformer circuit is connected to the fourth branch.

[0090] The secondary side coil of the second transformer circuit is connected to the fifth branch.

[0091] The secondary side coil of the third transformer circuit is connected to the sixth branch.

[0092] In the secondary-side switching circuit, each of the fourth branch, the fifth branch, and the sixth branch includes an upper branch and a lower branch connected in series, and each branch includes the switching element, the reverse-connected diode, and the partial resonant capacitor connected in parallel.

[0093] The control unit during the boost control:

[0094] During a period in which a resonant current flows through the first resonant circuit and the second resonant circuit, the switching element of the lower arm of the fourth branch or the switching element of the upper arm of the fifth branch is turned on to generate the boost period.

[0095] During a period in which a resonant current flows through the second resonant circuit and the third resonant circuit, the switching element of the lower arm of the fifth branch or the switching element of the upper arm of the sixth branch is turned on to generate the boost period.

[0096] During a period in which the resonant current flows in the third resonant circuit and the first resonant circuit, the switching element of the lower arm of the sixth branch or the switching element of the upper arm of the fourth branch is turned on to generate the boost period.

[0097] The three-phase current resonance type DC / DC converter can be configured as follows:

[0098] During the boost control, the control unit calculates a boost amount by multiplying a difference between the driving frequency and the first frequency by a frequency boost conversion gain, and determines a length of the boost period based on the boost amount.

[0099] The three-phase current resonance type DC / DC converter can be configured as follows:

[0100] During the boost control, the control unit calculates a boost amount by multiplying a difference between an output value of the secondary-side switching circuit and a predetermined target value by an output boost conversion gain, and determines a length of the boost period based on the boost amount.

[0101] The three-phase current resonance type DC / DC converter can be configured as follows:

[0102] The control unit during the boost control starts the boost period at the timing when the period during which the resonant current flows starts.

[0103] The three-phase current resonance type DC / DC converter can be configured as follows:

[0104] In the secondary-side switching circuit, each of the fourth branch, the fifth branch, and the sixth branch includes an upper branch and a lower branch connected in series, and each branch includes the switching element, the reverse-connected diode, and the partial resonant capacitor connected in parallel.

[0105] The control unit controls the secondary-side switching circuit to turn on any of the switching elements in each branch to generate a synchronous rectification period.

[0106] The control unit in the boost control and the synchronous rectification control starts the boost period at the start of the period in which the resonant current flows, and starts the synchronous rectification period at the end of the boost period.

[0107] The three-phase current resonance type DC / DC converter can be configured as follows:

[0108] The control unit makes the maximum value of the boost period variable according to the input-output voltage ratio so that the boost period becomes longer when the input-output voltage ratio of the input voltage of the primary-side switching circuit to the output voltage of the secondary-side switching circuit is higher.

[0109] The three-phase current resonance type DC / DC converter can be configured as follows:

[0110] In the secondary side switching circuit,

[0111] Each of the fourth branch, the fifth branch, and the sixth branch includes an upper branch and a lower branch connected in series.

[0112] One of the lower branch of the fourth branch and the upper branch of the fifth branch is composed of the switching element, the reverse-connected diode, and the partial resonant capacitor connected in parallel, and the other is composed of only a diode.

[0113] One of the lower branch of the fifth branch and the upper branch of the sixth branch is composed of the switching element, the reverse-connected diode, and the partial resonant capacitor connected in parallel, and the other is composed of only a diode.

[0114] One of the lower branch of the sixth branch and the upper branch of the fourth branch is composed of the switching element, the reverse-connected diode, and the partial resonant capacitor connected in parallel, and the other is composed of only a diode.

[0115] The control unit during the boost control:

[0116] During a period in which a resonant current flows through the first resonant circuit and the second resonant circuit, the switching element of the lower arm of the fourth branch or the switching element of the upper arm of the fifth branch is turned on to generate the boost period.

[0117] During a period in which a resonant current flows through the second resonant circuit and the third resonant circuit, the switching element of the lower arm of the fifth branch or the switching element of the upper arm of the sixth branch is turned on to generate the boost period.

[0118] During a period in which the resonant current flows in the third resonant circuit and the first resonant circuit, the switching element of the lower arm of the sixth branch or the switching element of the upper arm of the fourth branch is turned on to generate the boost period.

[0119] The three-phase current resonance type DC / DC converter can be configured as follows:

[0120] The secondary-side resonant circuit further comprises a fourth resonant circuit, a fifth resonant circuit, and a sixth resonant circuit, wherein the fourth resonant circuit is connected to the fourth branch and the secondary-side coil of the first transformer circuit, the fifth resonant circuit is connected to the fifth branch and the secondary-side coil of the second transformer circuit, and the sixth resonant circuit is connected to the sixth branch and the secondary-side coil of the third transformer circuit, each resonant circuit comprising a resonant coil and a resonant capacitor.

[0121] In the secondary-side switching circuit, each of the fourth branch, the fifth branch, and the sixth branch includes an upper branch and a lower branch connected in series, and each branch includes the switching element, the reverse-connected diode, and the partial resonant capacitor connected in parallel.

[0122] When performing reverse power transmission from the secondary-side switching circuit to the primary-side switching circuit, the control unit:

[0123] When the driving frequency of the secondary-side switching circuit is higher than the second frequency, frequency modulation control is performed to control the driving frequency according to the output of the primary-side switching circuit.

[0124] When the driving frequency is equal to or lower than the second frequency, a boost control is performed to generate a boost period in which the primary-side switching circuit is short-circuited.

[0125] During the boost control, the control unit turns on any of the switching elements in the primary-side switching circuit to generate the boost period during a period in which the resonant current flows through the fourth resonant circuit and the fifth resonant circuit, a period in which the resonant current flows through the fifth resonant circuit and the sixth resonant circuit, and a period in which the resonant current flows through the sixth resonant circuit and the fourth resonant circuit.

[0126] Effects of the Invention

[0127] According to the present invention, it is possible to provide a single-phase and three-phase current resonance type DC / DC converter that can support a wide range of output voltages without adding components and circuits or complicating the control software configuration. BRIEF DESCRIPTION OF THE DRAWINGS

[0128] Figure 1 1 is a diagram showing a single-phase current resonance type DC / DC converter according to the first embodiment.

[0129] Figure 2 Graphs showing the relationship between the drive frequency, the phase shift amount, and the boost amount during the discharge operation of the single-phase current resonance type DC / DC converter according to the first embodiment.

[0130] Figure 3 This is a control flow chart of the single-phase current resonance type DC / DC converter according to the first embodiment.

[0131] Figure 4 This is a control block diagram of the single-phase current resonance DC / DC converter according to the first embodiment.

[0132] Figure 5 This is a driving waveform diagram of the switching element during frequency modulation control of the single-phase current resonance type DC / DC converter according to the first embodiment.

[0133] Figure 6 A diagram showing the current flow of the single-phase current resonance type DC / DC converter according to the first embodiment, (A) shows Figure 5 Figure 1 shows the flow during mode 1, (B) shows the flow during mode 1 Figure 5 Diagram of the flow during Mode 3.

[0134] Figure 7This is a driving waveform diagram of the switching element during frequency phase shift conversion control of the single-phase current resonance type DC / DC converter according to the first embodiment.

[0135] Figure 8 A diagram showing the current flow of the single-phase current resonance type DC / DC converter according to the first embodiment, (A) shows Figure 7 Figure 2 shows the flow during mode 2, (B) shows the flow during mode 2 Figure 7 Diagram of the flow during Mode 4.

[0136] Figure 9 This is a driving waveform diagram of the switching element during frequency step-up conversion control of the single-phase current resonance type DC / DC converter according to the first embodiment.

[0137] Figure 10 A diagram showing the current flow of the single-phase current resonance type DC / DC converter according to the first embodiment, (A) shows Figure 9 Figure 1B shows the flow during mode 1B. (B) Figure 9 Diagram of the flow during mode 3B.

[0138] Figure 11 1 is a diagram showing a single-phase current resonance type DC / DC converter according to a first modification.

[0139] Figure 12 1 is a diagram showing a single-phase current resonance type DC / DC converter according to a second modification.

[0140] Figure 13 This is a driving waveform diagram of the switching element during frequency step-up conversion control of the single-phase current resonance type DC / DC converter according to the second modification.

[0141] Figure 14 is a diagram showing the current flow of a single-phase current resonance type DC / DC converter according to the second modification, (A) is a diagram showing Figure 13 Figure 1B shows the flow during mode 1B. (B) Figure 13 Diagram of the flow during mode 3B.

[0142] Figure 15 1 is a diagram showing a single-phase current resonance type DC / DC converter according to a third modification.

[0143] Figure 16 This is a driving waveform diagram of the switching element during frequency PWM conversion control of the single-phase current resonance type DC / DC converter according to the third modification.

[0144] Figure 17 is a diagram showing the current flow of a single-phase current resonance type DC / DC converter according to the third modification, (A) is a diagram showing Figure 16Figure 1 shows the flow during mode 1', (B) Figure 16 Diagram of the flow during Mode 2'.

[0145] Figure 18 is a diagram showing the current flow of a single-phase current resonance type DC / DC converter according to the third modification, (A) is a diagram showing Figure 16 Figure 3 'flow during mode, (B) shows Figure 16 Diagram of the flow during Mode 4'.

[0146] Figure 19 This is an example of a circuit diagram of a three-phase current resonance type DC / DC converter according to the second embodiment.

[0147] Figure 20 This is a diagram showing the relationship between the driving frequency and the boost amount of the three-phase current resonance type DC / DC converter according to the second embodiment.

[0148] Figure 21 This is a diagram showing a frequency step-up conversion control block of a three-phase current resonance type DC / DC converter according to the second embodiment.

[0149] Figure 22 This is an example of a control flow chart of the three-phase current resonance type DC / DC converter according to the second embodiment.

[0150] Figure 23 This is an example of a diagram showing control timing during diode rectification control of the three-phase current resonance type DC / DC converter according to the second embodiment.

[0151] Figure 24 The current path diagram of the three-phase current resonance type DC / DC converter of the second embodiment, (A) is Figure 23 The graph during mode 1 of Figure 23 Figure 2 during Mode 2.

[0152] Figure 25 The current path diagram of the three-phase current resonance type DC / DC converter of the second embodiment, (A) is Figure 23 The diagram during mode 3 of Figure 23 Figure 4 during Mode 4.

[0153] Figure 26 The current path diagram of the three-phase current resonance type DC / DC converter of the second embodiment, (A) is Figure 23 The graph during mode 5 of Figure 23 Figure 6 during mode 6.

[0154] Figure 27This is an example of a diagram showing control timings during diode rectification control and boost control of the three-phase current resonance type DC / DC converter according to the second embodiment.

[0155] Figure 28 The current path diagram of the three-phase current resonance type DC / DC converter of the second embodiment is shown in FIG. Figure 27 The graph during mode 1-1 is (B) Figure 27 Figure 1 during Mode 1.

[0156] Figure 29 The current path diagram of the three-phase current resonance type DC / DC converter of the second embodiment, (A) is Figure 27 The graph during Mode 3-1 is (B) Figure 27 Figure 3 during Mode 3.

[0157] Figure 30 The current path diagram of the three-phase current resonance type DC / DC converter of the second embodiment, (A) is Figure 27 The graph during mode 5-1 is (B) Figure 27 Figure 5 during Mode 5.

[0158] Figure 31 This is an example of a diagram showing control timing during synchronous rectification control of the three-phase current resonance type DC / DC converter according to the second embodiment.

[0159] Figure 32 The current path diagram of the three-phase current resonance type DC / DC converter of the second embodiment, (A) is Figure 31 The graph during mode 1 of Figure 31 Figure 2 during Mode 2.

[0160] Figure 33 The current path diagram of the three-phase current resonance type DC / DC converter of the second embodiment, (A) is Figure 31 The diagram during mode 3 of Figure 31 Figure 4 during Mode 4.

[0161] Figure 34 The current path diagram of the three-phase current resonance type DC / DC converter of the second embodiment is shown in FIG. Figure 31 The graph during mode 5 of Figure 31 Figure 6 during Mode 6.

[0162] Figure 35 This is an example of a control timing diagram during synchronous rectification control and boost control of the three-phase current resonance type DC / DC converter according to the second embodiment.

[0163] Figure 36The current path diagram of the three-phase current resonance type DC / DC converter of the second embodiment is shown in FIG. Figure 35 The graph during mode 1-1 is (B) Figure 35 Figure 1 during Mode 1.

[0164] Figure 37 The current path diagram of the three-phase current resonance type DC / DC converter of the second embodiment is shown in FIG. Figure 35 The graph during Mode 3-1 is (B) Figure 35 Figure 3 during Mode 3.

[0165] Figure 38 The current path diagram of the three-phase current resonance type DC / DC converter of the second embodiment is shown in FIG. Figure 35 The graph during mode 5-1 is (B) Figure 35 Figure 5 during Mode 5.

[0166] Figure 39 This is an example of a circuit diagram of a three-phase current resonance type DC / DC converter according to a first modification.

[0167] Figure 40 1 is a diagram showing an output voltage step-up conversion control block of a three-phase current resonance type DC / DC converter according to a first modification.

[0168] Figure 41 This is an example of a control flow chart of a three-phase current resonance DC / DC converter according to the first modification.

[0169] Figure 42 This is an example of a circuit diagram of a three-phase current resonance DC / DC converter according to a second modification.

[0170] Reference numerals

[0171] 1. 1A to 1C: Single-phase current resonant DC / DC converter; 10. 10A to 10C: Main circuit; 11: Primary-side switching circuit; 12: Primary-side resonant circuit; 13. 13A to 13C: Secondary-side rectifier circuit; 14: Secondary-side resonant circuit; 20. 20A to 20C: Control unit; 30: Frequency phase shift conversion control block; 31: First operation unit; 32: First multiplication unit; 33: First clamping unit; 40: Frequency boost conversion control block; 41: Second operation unit; 42: Second multiplication unit; 43: Second clamping unit; 100A Figures 100C to 100C: three-phase current resonant DC / DC converter; 110: transformer unit; 111: primary-side switching circuit; 112: primary-side resonant circuit; 113, 113C: secondary-side switching circuit; 114: secondary-side resonant circuit; 115A to 115C: control unit; 120: output voltage boost conversion control block; 120': frequency boost conversion control block; 121: first adding unit; 121': second adding unit; 122: first multiplying unit; 122': second multiplying unit; 123: first clamping unit; 123': second clamping unit. DETAILED DESCRIPTION

[0172] Hereinafter, embodiments of single-phase and three-phase current resonance type DC / DC converters according to the present invention will be described with reference to the accompanying drawings.

[0173] [Embodiment of Single-Phase Current Resonance DC / DC Converter (First Embodiment)]

[0174] Figure 1 A current resonant DC / DC converter 1 with a bidirectional CLLC single-phase full-bridge structure (hereinafter referred to as a "single-phase current resonant DC / DC converter") according to a first embodiment of the present invention is shown. The single-phase current resonant DC / DC converter 1 is composed of a main circuit unit 10 and a control unit 20. The main circuit unit 10 includes a transformer circuit Tr, a primary-side switching circuit 11, a primary-side resonant circuit 12, a secondary-side rectifier circuit 13, a secondary-side resonant circuit 14, and terminals T1 to T4.

[0175] The transformer circuit Tr is composed of one or more high-frequency isolation transformers. Each high-frequency isolation transformer includes a primary-side coil and a secondary-side coil. The primary-side coil is connected to the primary-side switching circuit 11 via the primary-side resonant circuit 12, while the secondary-side coil is connected to the secondary-side rectifier circuit 13 via the secondary-side resonant circuit 14.

[0176] The primary-side switching circuit 11 is a full-bridge circuit comprised of switching elements Q1 to Q4 (equivalent to the "primary-side switching elements" in this disclosure). Specifically, the primary-side switching circuit 11 includes a first branch and a second branch (equivalent to the "primary-side branch" in this disclosure) connected in parallel. In the first branch, a pair of switching elements Q1 and Q2 connected in series form the upper and lower branches. In the second branch, a pair of switching elements Q3 and Q4 connected in series form the upper and lower branches. While each of the switching elements Q1 to Q4 consists of a single switching element, they may also consist of two or more switching elements connected in parallel to distribute the current.

[0177] As switching elements Q1 to Q4, for example, power semiconductors such as IGBTs (insulated gate bipolar transistors) and MOSFETs (metal oxide semiconductor field effect transistors) using SiC (silicon carbide) or GaN (gallium nitride) can be used. In addition, as a drive circuit of the LLC method, it is preferred that reverse-connected diodes (hereinafter referred to as diodes) D1 to D4 for regeneration and partial resonant capacitors (hereinafter referred to as capacitors) C1 to C4 for soft switching are connected in parallel in the current paths of the switching elements Q1 to Q4. The diodes D1 to D4 can be built-in diodes of the switching elements Q1 to Q4 or external diodes. The capacitors C1 to C4 can be parasitic capacitances of the switching element Q1, external capacitors, or a combination thereof.

[0178] The primary-side resonant circuit 12 includes a resonant coil Lr1 and a resonant capacitor Cr1. Together with the excitation coil and primary-side coil of the transformer circuit Tr, the resonant coil Lr1 and resonant capacitor Cr1 form an LLC resonant circuit. The excitation coil of the transformer circuit Tr is omitted from illustration, assuming it is included within the transformer circuit Tr. The resonant coil Lr1 may consist solely of leakage magnetic flux from the transformer circuit Tr or may consist of leakage magnetic flux and a separate coil.

[0179] Secondary-side rectifier circuit 13 is a full-bridge circuit comprised of switching elements Q5-Q8 (equivalent to the "secondary-side switching elements" in this disclosure). Specifically, secondary-side rectifier circuit 13 includes a third branch and a fourth branch (equivalent to the "secondary-side branches" in this disclosure) connected in parallel. In the third branch, a pair of switching elements Q5 and Q6 connected in series form the upper and lower branches. In the fourth branch, a pair of switching elements Q7 and Q8 connected in series form the upper and lower branches. While each of switching elements Q5-Q8 consists of a single switching element, they can also be constructed with two or more switching elements connected in parallel to distribute the current.

[0180] As switching elements Q5-Q8, for example, the same power semiconductors as those used for switching elements Q1-Q4 can be used. Furthermore, reverse-connected diodes (hereinafter referred to as diodes) D5-D8 and partial resonant capacitors (hereinafter referred to as capacitors) C5-C8 for rectification are connected in parallel in the current paths of switching elements Q5-Q8. Furthermore, diodes D5-D8 can be external diodes or internal diodes of switching elements Q5-Q8, and capacitors C5-C8 can be parasitic capacitances of switching elements Q5-Q8 or can be absent when the rectification circuit is operating. Thus, the secondary-side rectifier circuit 13 has the same structure as the primary-side switching circuit 11, and the switching elements Q5-Q8, diodes D5-D8, and capacitors C5-C8 have the same structure as the primary-side switching elements Q1-Q4, diodes D1-D4, and capacitors C1-C4.

[0181] The secondary-side resonant circuit 14 includes a resonant coil Lr2 and a resonant capacitor Cr2. Together with the excitation coil and secondary-side coil of the transformer circuit Tr, the resonant coil Lr2 and resonant capacitor Cr2 form an LLC resonant circuit. Specifically, the secondary-side resonant circuit 14 has the same structure as the primary-side resonant circuit 12, with the main circuit unit 10 being a symmetrical circuit with the transformer circuit Tr interposed therebetween.

[0182] The control unit 20 includes a control circuit for controlling the switching elements Q1-Q4 of the primary-side switching circuit 11 and the switching elements Q5-Q8 of the secondary-side rectifier circuit 13. The control circuit of the control unit 20 is composed of a control processor, memory, and peripheral circuits. For example, a microcontroller or DSP can be used. The control unit 20 also includes a detection circuit composed of sensors and A / D converters for detecting voltage, current, and other parameters required for control. However, the description and illustration of the detection circuit are omitted.

[0183] The control unit 20 monitors the input (input current, input voltage, or input power) of the main circuit unit 10 and controls the opening and closing (on and off) of the switching elements Q1 to Q8 so that the output (output current, output voltage, or output power) of the main circuit unit 10 becomes a desired value.

[0184] Specifically, the control unit 20 performs frequency modulation control, frequency phase shift conversion control (equivalent to the "output suppression conversion control" of the present invention), and frequency boost conversion control (equivalent to the "output increase conversion control" of the present invention). These controls are all frequency controls. Frequency phase shift conversion control differs from conventional phase shift control in that the phase shift amount calculated using the drive frequency (switching frequency) is used as the control variable. Frequency boost conversion control differs from conventional boost control in that the boost amount calculated using the drive frequency (switching frequency) is used as the control variable.

[0185] In the following, it is assumed that a rechargeable battery (e.g., an electric vehicle battery) is connected to terminals T1 and T2, and that the DC terminals of a DC / AC inverter are connected to terminals T3 and T4. Furthermore, it is assumed that the main circuit unit 10 performs a discharge operation to generate a bus voltage (output voltage) V2 for the DC / AC inverter based on the battery voltage (input voltage) V1.

[0186] During discharge operation, the control unit 20 monitors the input voltage V1 while controlling the driver-side switching elements Q1-Q4 and the rectifier-side switching elements Q5-Q8 to achieve the desired output voltage V2. For example, during frequency modulation control, the control unit 20 switches the switching elements Q1-Q4 on and off at an appropriate drive frequency and performs synchronous rectification or diode rectification on the switching elements Q5-Q8. For simplicity, the following assumes that the switching elements Q5-Q8 are OFF (diode rectification control is performed).

[0187] Figure 2 The following diagram shows the relationship between the driving frequency f of switching elements Q1-Q4 during discharge operation, the phase shift amount θ of the frequency phase shift conversion control, and the boost amount ψ of the frequency boost conversion control. The phase shift amount θ corresponds to the "first control amount" of the present invention, and the boost amount ψ corresponds to the "second control amount" of the present invention. Here, the phase shift amount θ is the phase difference between the first branch (switching elements Q1 and Q2) and the second branch (switching elements Q3 and Q4) of the primary-side switching circuit 11. The boost amount ψ is the value expressed in [°], which represents the period during which any of the switching elements Q5-Q8 of the secondary-side rectifier circuit 13 is turned on and short-circuited while the resonant current flows through the primary-side resonant circuit 12.

[0188] The driving frequency f is included in the predetermined second frequency fψs ( Figure 2 90 [kHz]) to the first frequency fθs (in Figure 2 When the output is within the range of 140 [kHz], the control unit 20 performs frequency modulation control to control the output of the main circuit unit 10 using the drive frequency f as the control amount. During frequency modulation control, the control unit 20 decreases the drive frequency f when increasing the output and increases the drive frequency f when decreasing the output. Figure 2 In the example, the high resonant frequency fr determined by the resonant coil Lr1 and the resonant capacitor Cr1 is set to 125 kHz, and the low resonant frequency f0 including the exciting coil of the transformer circuit Tr is set to 30 kHz. That is, the frequencies are set to satisfy f0 < fψs < fr < fθs in operation.

[0189] When the drive frequency f increases as the output decreases and the drive frequency f exceeds the first frequency fθs, or 140 kHz, the control unit 20 begins frequency phase shift conversion control. The control unit 20 sets the phase shift amount θ, or the control variable for frequency phase shift conversion control, to the value obtained by multiplying the difference between the current drive frequency f and the first frequency fθs by the first gain Kθ. The first gain Kθ is a predetermined frequency phase shift conversion gain. During frequency phase shift conversion control, the control unit 20 decreases the phase shift amount θ when increasing the output and increases the phase shift amount θ when decreasing the output.

[0190] When the phase shift θ increases as the output further decreases and the drive frequency f exceeds the maximum phase shift drive frequency fθh, i.e., 225 [kHz], the control unit 20 fixes the phase shift θ at the maximum phase shift θh (equivalent to the "first maximum control amount" of the present invention), i.e., 170°. The phase shift θ [°] is the angle relative to one cycle T (T = 360°) of the drive frequency f, and ranges from 0° to a maximum of 180°.

[0191] On the other hand, when the drive frequency f decreases as the output increases and the drive frequency f falls below the second frequency fψs, i.e., 90 [kHz], the control unit 20 starts the frequency boost conversion control. The control unit 20 sets the control amount of the frequency boost conversion control, i.e., the boost amount ψ, to a value obtained by multiplying the difference between the second frequency fψs and the drive frequency f at that time by the second gain Kψ. The second gain Kψ is a predetermined frequency boost conversion gain, which is obtained when the input voltage V1, i.e., the battery voltage V1x, is V11 (at Figure 2 When V11=300[V]) is Kψ(V11), when the battery voltage V1x is V12 (in Figure 2 When V12=150 [V]), Kψ(V12) is used (where Kψ(V12)>Kψ(V11)). The control unit 20 during frequency boost conversion control increases the boost amount ψ when increasing the output and decreases the boost amount ψ when decreasing the output.

[0192] When the boost amount ψ increases as the output further increases and the drive frequency f is lower than the maximum boost drive frequency fψh, i.e., 67 [kHz], the control unit 20 fixes the boost amount ψ to the maximum boost amount ψh (equivalent to the "second maximum control amount" of the present invention). Figure 2 , the maximum boost amount ψh(V11) when the battery voltage V1x is V11 is 15°, and the maximum boost amount ψh(V12) when the battery voltage V1x is V12 is 45°.

[0193] During frequency boost conversion control, switching elements Q5 and Q6 are turned on to perform boost drive. However, during boost drive, the resonant current short-circuits in the secondary-side rectifier circuit 13, briefly accumulating a large amount of energy in the resonant coils Lr1 and Lr2. Consequently, changes in the boost pulse width (boost amount ψ) used to turn on switching elements Q5 and Q6 cause significant fluctuations in output. Consequently, control can become unstable depending on the battery voltage V1x. Regarding this issue, in this embodiment, the second gain Kψ and the maximum boost amount ψh are made variable based on the battery voltage V1x. This allows the slope of the boost amount ψ to be adjusted in conjunction with the battery voltage V1x, enabling a large boost ratio to be achieved in accordance with input and output conditions while preventing control instability.

[0194] For example, the control unit 20 may pre-store predetermined input / output conditions (input voltage V1, input / output voltage ratio V1 / V2, or output power) and the second gain Kψ and maximum boost amount ψh associated with these conditions in a memory. The detection circuit acquires input / output information and, based on the detected input / output information, reads the second gain Kψ and maximum boost amount ψh from the memory to initiate frequency-based boost conversion control. By making the second gain Kψ and maximum boost amount ψh variable according to the input / output conditions, an optimal boost ratio can be achieved even when the input voltage V1 is low, and the range of output voltage V2 that can be outputted can be appropriately expanded. Furthermore, similarly to the second gain Kψ and maximum boost amount ψh, the second frequency fψs can be made variable according to the input / output conditions.

[0195] Alternatively, the first gain Kθ, maximum phase shift θh, and first frequency fθs may be made variable according to input / output conditions. For example, the control unit 20 may pre-store predetermined input / output conditions and the first gain Kθ, maximum phase shift θh, and first frequency fθs associated with the conditions in a memory, acquire input / output information using a detection circuit, and read the first gain Kθ, maximum phase shift θh, and first frequency fθs from the memory based on the detected input / output information to initiate frequency / phase shift conversion control.

[0196] Figure 3 A control flow chart related to the transition control between frequency modulation control, frequency phase shift conversion control, and frequency step-up conversion control executed by the control circuit of the control unit 20 is shown. Furthermore, it is assumed that the control unit 20, before performing the transition control, performs frequency modulation control, switches the switching elements Q1 to Q4 at an appropriate drive frequency f, and turns off the switching elements Q5 to Q8 to enable the secondary-side rectifier circuit 13 to perform diode rectification.

[0197] When the control unit 20 starts the transfer control (S1), the detection circuit acquires input and output information (S2). The input and output information includes at least one output value of the input voltage V1, input current, output voltage V2, and output current.

[0198] The control unit 20 that has acquired the input and output information reads the control parameters from the memory (S3). The control parameters include the first frequency fθs, the first gain Kθ, the maximum phase shift θh, the second frequency fψs, the second gain Kψ, and the maximum boost ψh. Figure 2 As shown, the second gain Kψ and the maximum boost amount ψh take different values depending on the input voltage V1. Furthermore, the fixed parameters among the control parameters may be read only once when the transition control is started.

[0199] Next, the control unit 20 compares the output value (output current value, output voltage value, or output power value) included in the input / output information with the target value (target output current value, target output voltage value, or target output power value) and determines the driving frequency f of the switching elements Q1-Q4 so that the output value approaches the target value (S4). For example, if the input / output information includes the output value of output voltage V2 and the target value is the target output voltage value, the control unit 20 compares the output value of output voltage V2 with the target output voltage value and determines the driving frequency f so that the output value of output voltage V2 approaches the target output voltage value.

[0200] When the output value is less than the target value, the control unit 20 decreases the drive frequency f to increase the output. On the other hand, when the output value is greater than the target value, the control unit 20 increases the drive frequency f to decrease the output. The target value may be, for example, a value preset in the single-phase current resonant DC / DC converter 1, a value determined based on input and output conditions, or a value input from an external device.

[0201] The control unit 20 determines whether the drive frequency f determined in step S4 is greater than the first frequency fθs read in step S3 (S5). If the drive frequency f is not greater than the first frequency fθs (No in S5), the control unit 20 determines whether the drive frequency f determined in step S4 is less than the second frequency fψs read in step S3 (S6).

[0202] If the drive frequency f is not less than the second frequency fψs (No in S6), the control unit 20 performs frequency modulation control. During frequency modulation control, the control unit 20 sets the phase difference between the switching elements Q1 and Q2 of the first branch and the switching elements Q3 and Q4 of the second branch to zero, switches the switching elements Q1 to Q4 on and off at the drive frequency f determined in step S4, and turns off the switching elements Q5 and Q6 of the third branch and the switching elements Q7 and Q8 of the fourth branch to enable the secondary-side rectifier circuit 13 to perform diode rectification (S7).

[0203] In step S5, if the drive frequency f is greater than the first frequency fθs (YES in S5), the control unit 20 performs frequency phase shift conversion control. During frequency phase shift conversion control, the control unit 20 multiplies the difference between the drive frequency f determined in step S4 and the first frequency fθs by the first gain Kθ read in step S3 to calculate the phase shift amount θ (S8).

[0204] The control unit 20, which has calculated the phase shift amount θ, determines whether the phase shift amount θ is greater than the maximum phase shift amount θh read in step S3 (S9). If the phase shift amount θ is greater than the maximum phase shift amount θh (YES in S9), the control unit 20 fixes the phase shift amount θ at the maximum phase shift amount θh by changing the phase shift amount θ calculated in step S8 to the maximum phase shift amount θh (S10), and then moves on to the next step S11. If the phase shift amount θ is less than the maximum phase shift amount θh (NO in S9), the control unit 20 maintains the phase shift amount θ at the phase shift amount θ calculated in step S8 and moves on to step S11.

[0205] The control unit 20 then proceeds to step S11, causing the primary-side switching circuit 11 to perform phase shift driving. Specifically, the control unit 20 shifts the phase of the second branch relative to the phase of the first branch by an amount corresponding to the phase shift amount θ, switches the switching elements Q1 to Q4 on and off at the drive frequency f determined in step S4, and turns off the switching elements Q5 and Q6 of the third branch and the switching elements Q7 and Q8 of the fourth branch, causing the secondary-side rectifier circuit 13 to perform diode rectification (S11).

[0206] In step S6, if the drive frequency f is less than the second frequency fψs (YES in S6), the control unit 20 performs frequency boost conversion control. During frequency boost conversion control, the control unit 20 multiplies the difference between the second frequency fψs and the drive frequency f determined in step S4 by the second gain Kψ read in step S3 to calculate the boost amount ψ (S12).

[0207] The control unit 20, which has calculated the boost amount ψ, determines whether the boost amount ψ is greater than the maximum boost amount ψh read in step S3 (S13). If the boost amount ψ is greater than the maximum boost amount ψh (YES in S13), the control unit 20 fixes the boost amount ψ at the maximum boost amount ψh by changing the boost amount ψ calculated in step S12 to the maximum boost amount ψh (S14), and then moves on to the next step S15. If the boost amount ψ is less than the maximum boost amount ψh (NO in S13), the control unit 20 maintains the boost amount ψ at the boost amount ψ calculated in step S12 and moves on to step S15.

[0208] The control unit 20 then proceeds to step S15, causing the secondary-side rectifier circuit 13 to perform a boost drive. Specifically, the control unit 20 sets the phase difference between the first and second branches to zero, switches the switching elements Q1 to Q4 on and off at the drive frequency f determined in step S4, turns on the switching elements Q5 and Q6 of the third branch by an amount corresponding to the boost amount ψ in synchronization with the first branch, and turns off the switching elements Q7 and Q8 of the fourth branch to perform diode rectification (S15).

[0209] After steps S7, S11, and S15, the control unit 20 determines whether to continue control (S16). The control unit 20 makes this determination in step S16, for example, based on whether it has received a control end command from an external device. If the control unit 20 has not received a control end command, it determines that the transfer control is to continue (Yes in S16), returns to step S2, and repeats the process from step S2 onward. If the control unit 20 has received a control end command, it determines that the transfer control is not to continue (No in S16), and ends the transfer control (S17).

[0210] Figure 4 The following is a control block diagram of frequency phase shift conversion control and frequency boost conversion control performed by the control unit 20. The phase shift amount θ[°] and the boost amount ψ[°] are angles when the period T of the drive frequency f[Hz] is 360°.

[0211] The control unit 20 includes a frequency phase shift conversion control block 30 and a frequency boost conversion control block 40. The frequency phase shift conversion control block 30 includes a first operation unit 31, a first multiplication unit 32, and a first clamping unit 33, while the frequency boost conversion control block 40 includes a second operation unit 41, a second multiplication unit 42, and a second clamping unit 43.

[0212] When reducing the output, the control unit 20 increases the drive frequency f of the switching elements Q1-Q4. When the drive frequency f exceeds the first frequency fθs, the first calculation unit 31 outputs the difference between the drive frequency f and the first frequency fθs to the first multiplication unit 32. The first multiplication unit 32 multiplies the difference by the first gain Kθ to calculate the phase shift amount θ, and outputs this phase shift amount θ to the first clamping unit 33. The first clamping unit 33 sets the upper limit of the phase shift amount θ to the maximum phase shift amount θh and outputs the phase shift amount θ [°] = (f - fθs) × Kθ ≤ θh.

[0213] The control unit 20 drives the primary-side switching circuit 11 with a phase shift based on the phase shift amount θ[°] = (f - fθs) × Kθ ≤ θh. Specifically, the control unit 20 switches the switching elements Q3 and Q4 of the second branch on and off with the phase shifted by an amount corresponding to the phase shift amount θ (an amount corresponding to the phase shift amount θ relative to the period T) relative to the on and off of the switching elements Q1 and Q2 of the first branch.

[0214] In the frequency phase shift conversion control block 30, the drive frequency f increases when the output is reduced, so the difference (f-fθs) output from the first operation unit 31 increases. When the drive frequency f reaches the maximum phase shift drive frequency fθh (see Figure 2 ), the phase shift amount θ increases uniformly with the increase of the drive frequency f. When the drive frequency f exceeds the maximum phase shift drive frequency fθh, the phase shift amount θ is fixed at the maximum phase shift amount θh. Furthermore, it is assumed that the control unit 20 controls the drive frequency f so as not to exceed the maximum phase shift drive frequency fθh.

[0215] When the control unit 20 increases the output, the driving frequency f of the switching elements Q1 to Q4 is reduced. When the driving frequency f is less than the second frequency fψs, the second operation unit 41 outputs the difference between the second frequency fψs and the driving frequency f to the second multiplication unit 42. The second multiplication unit 42 multiplies the above difference by the second gain Kψ(V1x) to calculate the boost amount ψ, and outputs the boost amount ψ to the second clamping unit 43. The second clamping unit 43 sets the upper limit of the boost amount ψ to the maximum boost amount ψh(V1x), and outputs the boost amount ψ[°]=(fψs-f)×Kψ(V1x)≤ψh(V1x). The second gain Kψ(V1x) and the maximum boost amount ψh(V1x) are based on the input voltage V1, that is, the battery voltage V1x (for example, refer to Figure 2 The amount that changes with V11, V12).

[0216] The control unit 20 drives the secondary-side rectifier circuit 13 to perform a boost operation based on the following equation: boost amount ψ[°] = (fψs - f) × Kψ(V1x) ≤ ψh(V1x). Specifically, the control unit 20 turns on the switching elements Q5 and Q6 of the third branch by an amount corresponding to the boost amount ψ (an amount corresponding to the boost amount ψ relative to period T) in synchronization with the switching elements Q1 and Q2 of the first branch. Furthermore, the control unit 20 turns off the switching elements Q7 and Q8 of the fourth branch to perform diode rectification.

[0217] In the frequency boost conversion control block 40, the drive frequency f decreases when the output is increased, so the difference (fψs-f) output from the second operation unit 41 increases. When the drive frequency f decreases and reaches the maximum boost drive frequency fψh (see Figure 2 ), the boost amount ψ increases uniformly as the drive frequency f decreases. When the drive frequency f falls below the maximum boost drive frequency fψh, the boost amount ψ is fixed at the maximum boost amount ψh (V1x). Furthermore, the turns ratio (step-up ratio) of the transformer circuit Tr is pre-designed based on input and output conditions so that the maximum boost drive frequency fψh is higher than the resonant frequency f0 and the drive frequency f does not fall below the maximum boost drive frequency fψh. The control unit 20 controls the drive frequency f to prevent it from falling below the maximum boost drive frequency fψh.

[0218] Figure 5 The following diagram shows the driving waveforms for switching elements Q1-Q8 during frequency modulation control. Switching element Q1 is turned on when the driving waveform is high and turned off when the driving waveform is low. Similarly, switching elements Q2-Q8 are turned on when the driving waveform is high and turned off when the driving waveform is low.

[0219] like Figure 5 As shown, the driving waveforms for switching elements Q1 and Q2 in the first branch and Q3 and Q4 in the second branch are in phase (with a phase difference of zero). The driving waveforms for switching elements Q5 and Q6 in the third branch and Q7 and Q8 in the fourth branch are always low.

[0220] The control unit 20 sets the phase difference between the switching elements Q1 and Q2 of the first branch and the switching elements Q3 and Q4 of the second branch to zero, drives the switching elements Q1 to Q4 with a period To and an on-duty cycle D (for example, D = 50%), and disconnects the switching elements Q5 and Q6 of the third branch and the switching elements Q7 and Q8 of the fourth branch to enable the secondary-side rectifier circuit 13 to perform diode rectification.

[0221] Furthermore, the control unit 20 causes the switching elements Q1 and Q2 constituting the first arm to alternately turn on and off with a phase difference of 180°, and causes the switching elements Q3 and Q4 constituting the second arm to alternately turn on and off with a phase difference of 180°. Furthermore, there is a predetermined dead time in the on-off timing of the switching elements Q1 and Q2, and there is also a predetermined dead time in the on-off timing of the switching elements Q3 and Q4. However, for simplicity of description, illustration of the dead time is omitted. This applies to the following figures as well.

[0222] Figure 6 (A) shows Figure 5 The current flowing in the main circuit unit 10 during mode 1, Figure 6 (B) shows Figure 5 The current of the main circuit unit 10 flows during Mode 3. Figure 6 The currents shown are the resonant current and the load current, and the illustration of the excitation current is omitted. In addition, the currents when the switching elements Q1 to Q4 are turned on and off (currents during transition) are also omitted.

[0223] exist Figure 5 During Mode 1, the switching elements Q1 and Q4 are turned on, and the switching elements Q2 and Q3 are turned off. Figure 6 As shown in (A), battery voltage V1 is applied between terminals T1 and T2, and a resonant current flows through resonant coil Lr1 and resonant capacitor Cr1 via switching elements Q1 and Q4. As a result, on the rectifier side, a load current flows between terminals T3 and T4 via diodes D5 and D8.

[0224] exist Figure 5 During Mode 3, the switching elements Q1 and Q4 are turned off, and the switching elements Q2 and Q3 are turned on. Figure 6 As shown in (B), battery voltage V1 is applied between terminals T1 and T2, and a resonant current in the opposite direction to that in Mode 1 flows through resonant coil Lr1 and resonant capacitor Cr1 via switching elements Q2 and Q3. As a result, on the rectifier side, a load current flows between terminals T3 and T4 via diodes D6 and D7.

[0225] Figure 7 The driving waveforms of the switching elements Q1 to Q8 during frequency phase shift control are shown. Figure 8 (A) shows Figure 7 The current flowing in the main circuit unit 10 during mode 2 is Figure 8 (B) shows Figure 7 The current of the main circuit unit 10 flows during Mode 4. Figure 8 The currents shown are the resonant current and the load current; the excitation current and the transient current are not shown.

[0226] like Figure 7 As shown, the phase of the driving waveform of the switching elements Q3 and Q4 of the second branch is shifted by an amount corresponding to the phase shift θ relative to the driving waveform of the switching elements Q1 and Q2 of the first branch. The driving frequency f of the switching elements Q1 to Q4 increases by an amount corresponding to the value obtained by dividing the phase shift θ by the first gain Kθ, and the period Tθ becomes shorter than Figure 5 In addition, the drive waveforms of the switching elements Q5 and Q6 of the third branch and the drive waveforms of the switching elements Q7 and Q8 of the fourth branch are always low and are turned off.

[0227] Figure 7 The states of the switching elements Q1 to Q4 during Mode 1 and Mode 3 are the same as Figure 5 The current flow of the main circuit unit 10 during this period is the same as that of mode 1 and mode 3. Figure 6 (A) and (B) are the same. However, Figure 7 The periods of Mode 1 and Mode 3 are shortened by at least the amount corresponding to the phase shift θ. Figure 5 Compared with the cases of mode 1 and mode 3, the resonant current is reduced and the load current is also reduced.

[0228] exist Figure 7 During Mode 2, the switching elements Q1 and Q3 are turned on, and the switching elements Q2 and Q4 are turned off. Figure 8 As shown in (A), the resonant current on the driving side continues to flow through the switching elements Q1 and Q3, but gradually decreases to zero. Similarly, the load current on the rectifying side continues to flow through the diodes D5 and D8, but gradually decreases to zero.

[0229] exist Figure 7 During Mode 4, the switching elements Q2 and Q4 are turned on, and the switching elements Q1 and Q3 are turned off. Figure 8 As shown in (B), the resonant current on the drive side continues to flow through the switching elements Q2 and Q4, but gradually decreases to zero. Similarly, the load current on the rectifier side continues to flow through the diodes D6 and D7, but gradually decreases to zero.

[0230] Figure 9 The driving waveforms of the switching elements Q1 to Q8 during frequency step-up conversion control are shown. Figure 10 (A) shows Figure 9 The current flowing in the main circuit unit 10 during mode 1B, Figure 10 (B) shows Figure 9 The current of the main circuit unit 10 during mode 3B flows. Figure 10 The currents shown are the resonant current and the load current; the excitation current and the transient current are not shown.

[0231] like Figure 9As shown, the driving waveforms of the switching elements Q1 and Q2 of the first branch and the driving waveforms of the switching elements Q3 and Q4 of the second branch are in the same phase (the phase difference is zero). The driving frequency f of the switching elements Q1 to Q4 is reduced by an amount corresponding to the value obtained by dividing the boost amount ψ by the second gain Kψ, and the period Tψ becomes longer than Figure 5 The cycle is To.

[0232] The drive waveform of the third-branch switching element Q6 goes high in synchronization with the drive waveform of the first-branch switching element Q1 (or the drive waveform of the second-branch switching element Q4), and then goes low after the period of boost ψ (during mode 1B). The drive waveform of the third-branch switching element Q5 goes high in synchronization with the drive waveform of the first-branch switching element Q2 (or the drive waveform of the second-branch switching element Q3), and then goes low after the period of boost ψ (during mode 3B). The drive waveforms of the fourth-branch switching elements Q7 and Q8 remain low.

[0233] Figure 9 The states of the switching elements Q1 to Q8 during Mode 1 and Mode 3 are the same as Figure 5 The current flow of the main circuit unit 10 during this period is the same as that of mode 1 and mode 3. Figure 6 (A) and (B) are the same.

[0234] exist Figure 9 During mode 1B, the switching elements Q1, Q4, and Q6 are turned on, and the switching elements Q2, Q3, Q5, Q7, and Q8 are turned off. Figure 10 As shown in (A), battery voltage V1 is applied between terminals T1 and T2, causing a resonant current to flow through resonant coil Lr1 and resonant capacitor Cr1 via switching elements Q1 and Q4. The rectifier side is short-circuited, forming a current path through switching element Q6 and diode D8, allowing current to continue flowing. Consequently, a large resonant current flows through resonant coil Lr1 and resonant capacitor Cr1, accumulating a large amount of energy. This accumulated energy flows as load current between terminals T3 and T4 in the subsequent Mode 1. This results in a voltage boost operation in the main circuit unit 10.

[0235] exist Figure 9 During mode 3B, the switching elements Q2, Q3, and Q5 are turned on, and the switching elements Q1, Q4, and Q6 to Q8 are turned off. Figure 10As shown in (B), battery voltage V1 is applied between terminals T1 and T2, and a resonant current in the opposite direction to that in mode 1B flows through resonant coil Lr1 and resonant capacitor Cr1 via switching elements Q2 and Q3. The rectifier side is short-circuited, forming a current path through switching element Q5 and diode D7, allowing current to continue flowing. Consequently, a large resonant current flows through resonant coil Lr1 and resonant capacitor Cr1, accumulating a large amount of energy. This accumulated energy flows as load current between terminals T3 and T4 in the subsequent mode 3. This results in a voltage boost operation in the main circuit unit 10.

[0236] As described above, when the drive frequency f exceeds the first frequency fθs in order to reduce output, the single-phase current resonant DC / DC converter 1 performs frequency phase shift conversion control with a phase shift amount θ corresponding to (f - fθs) × Kθ. During Modes 2 and 4 of the frequency phase shift conversion control, the battery voltage V1 is not applied to the drive side, and the main circuit unit 10 performs output suppression (voltage reduction) operation.

[0237] Furthermore, when the drive frequency f falls below the second frequency fψs in an attempt to increase output, the single-phase current resonant DC / DC converter 1 performs frequency-boosting conversion control with a boost amount ψ corresponding to (fψs - f) × Kψ. During Modes 1B and 3B of frequency-boosting conversion control, energy generated by the resonant current is accumulated and converted into load current during the subsequent Modes 1 and 3. In other words, the main circuit unit 10 performs an output-boosting operation (step-up operation).

[0238] The phase shift amount θ during frequency phase shift conversion control and the boost amount ψ during frequency boost conversion control are both controlled variables related to the drive frequency f. Therefore, the single-phase current resonant DC / DC converter 1 does not require switching to another control mode for control, nor does it require countermeasures for output fluctuations associated with switching control modes. Control can be performed solely based on frequency control. Consequently, the single-phase current resonant DC / DC converter 1 can support a wide range of output voltages without adding additional components or circuits, or complicating the control software structure.

[0239] Furthermore, in the single-phase current resonant DC / DC converter 1, the control parameters (first frequency fθs, first gain Kθ, maximum phase shift θh, second frequency fψs, second gain Kψ, maximum boost ψh) can be changed according to input and output conditions (e.g., input voltage V1, input / output voltage ratio V1 / V2, or output power). This allows switching elements Q1-Q4 to be switched within a soft-switching range during frequency phase shift conversion control, improving the power conversion efficiency of the main circuit unit 10. Furthermore, during frequency boost conversion control, the slope of the boost ψ can be adjusted in conjunction with the battery voltage V1x, preventing control instability.

[0240] While the description thus far describes the case where the main circuit unit 10 is discharging the battery connected between terminals T1 and T2, when the main circuit unit 10 is charging the battery, the secondary-side rectifier circuit 13 and secondary-side resonant circuit 14 serve as the drive-side circuit, while the primary-side switch circuit 11 and primary-side resonant circuit 12 serve as the rectifier-side circuit. During charging operation, V1 is the output voltage, and V2 is the input voltage. During charging operation, the control unit 20 controls the secondary-side rectifier circuit 13 on the drive side in the same manner as the primary-side switch circuit 11 during discharging operation, and controls the primary-side switch circuit 11 on the rectifier side in the same manner as the secondary-side rectifier circuit 13 during discharging operation.

[0241] [First Modification of Single-Phase Current Resonance DC / DC Converter]

[0242] Figure 11 A single-phase current resonance type DC / DC converter 1A according to a first modification of the first embodiment is shown. The single-phase current resonance type DC / DC converter 1A comprises a main circuit unit 10A and a control unit 20A. The main circuit unit 10A includes a transformer circuit Tr, a primary-side switching circuit 11, a primary-side resonant circuit 12, a secondary-side rectifier circuit 13A, and terminals T1 to T4.

[0243] The single-phase current resonant DC / DC converter 1A is configured to perform unidirectional power transmission from terminals T1 and T2 (V1) to terminals T3 and T4 (V2). The single-phase current resonant DC / DC converter 1A differs from the first embodiment in that it does not include the secondary-side resonant circuit 14, includes a secondary-side rectifier circuit 13A in place of the secondary-side rectifier circuit 13, and includes a control unit 20A in place of the control unit 20. All other aspects are the same.

[0244] The secondary-side rectifier circuit 13A has the same configuration as the secondary-side rectifier circuit 13 of the first embodiment, except that the fourth branch is formed only of diodes D7 and D8 connected in series.

[0245] The control unit 20A controls switching elements Q1-Q6 to enable the main circuit unit 10A to transmit power in one direction, from terminals T1 and T2 (V1) to terminals T3 and T4 (V2). In the first embodiment, the control during discharge operation was described with switching elements Q7 and Q8 off. Therefore, the control of the control unit 20A is identical to the control during discharge operation in the first embodiment. Specifically, the control unit 20A includes the components of the control unit 20 for discharge operation control and, similarly to the control unit 20, performs frequency modulation control, frequency phase shift conversion control, and frequency step-up conversion control.

[0246] [Second Modification of Single-Phase Current Resonance DC / DC Converter]

[0247] Figure 12 A single-phase current resonance type DC / DC converter 1B according to a second modified example of the first embodiment is shown. The single-phase current resonance type DC / DC converter 1B comprises a main circuit unit 10B and a control unit 20B. The main circuit unit 10B includes a transformer circuit Tr, a primary-side switching circuit 11, a primary-side resonant circuit 12, a secondary-side rectifier circuit 13B, and terminals T1 to T4.

[0248] The single-phase current resonant DC / DC converter 1B is configured to perform unidirectional power transmission from terminals T1 and T2 (V1) to terminals T3 and T4 (V2). The single-phase current resonant DC / DC converter 1B differs from the first embodiment in that it lacks the secondary-side resonant circuit 14, includes a secondary-side rectifier circuit 13B in place of the secondary-side rectifier circuit 13, and includes a control unit 20B in place of the control unit 20. Other aspects are the same.

[0249] The secondary-side rectifier circuit 13B has the same configuration as the secondary-side rectifier circuit 13 of the first embodiment, except that the upper arm of the third branch is composed only of the diode D5 and the upper arm of the fourth branch is composed only of the diode D7.

[0250] The control unit 20B controls switching elements Q1-Q4, Q6, and Q8 so that the main circuit unit 10B performs unidirectional power transmission from terminals T1 and T2 (V1) to terminals T3 and T4 (V2). The control unit 20B includes the components of the control unit 20 for controlling discharge operation. Similar to the control unit 20, it performs frequency modulation control, frequency phase shift conversion control, and frequency step-up conversion control. However, unlike the first embodiment, switching elements Q6 and Q8 are turned on during frequency step-up conversion control.

[0251] Figure 13 The driving waveforms of the switching elements Q1 to Q4, Q6, and Q8 during frequency step-up conversion control are shown. Figure 14 (A) shows Figure 13 The current flowing in the main circuit unit 10B during mode 1B, Figure 14 (B) shows Figure 13 The current of the main circuit unit 10B flows during the mode 3B. Figure 14 The currents shown are the resonant current and the load current; the excitation current and the transient current are not shown.

[0252] exist Figure 13 During mode 1B, the switching elements Q1, Q4, and Q6 are turned on, and the switching elements Q2, Q3, and Q8 are turned off. Figure 14 As shown in (A), battery voltage V1 is applied between terminals T1 and T2, causing a resonant current to flow through resonant coil Lr1 and resonant capacitor Cr1 via switching elements Q1 and Q4. The rectifier side is short-circuited, forming a current path through switching element Q6 and diode D8, allowing current to continue flowing. Consequently, a large resonant current flows through resonant coil Lr1 and resonant capacitor Cr1, accumulating a large amount of energy. This accumulated energy flows as load current between terminals T3 and T4 in the subsequent Mode 1. This results in a voltage boost operation in main circuit section 10B.

[0253] exist Figure 13 During mode 3B, the switching elements Q2, Q3, and Q8 are turned on, and the switching elements Q1, Q4, and Q6 are turned off. Figure 14 As shown in (B), battery voltage V1 is applied between terminals T1 and T2, and a resonant current in the opposite direction to that in mode 1B flows through resonant coil Lr1 and resonant capacitor Cr1 via switching elements Q2 and Q3. The rectifier side is short-circuited, forming a current path through switching element Q8 and diode D6, allowing current to continue flowing. Consequently, a large resonant current flows through resonant coil Lr1 and resonant capacitor Cr1, accumulating a large amount of energy. This accumulated energy flows as load current between terminals T3 and T4 in the subsequent mode 3. This results in a voltage boost operation in main circuit section 10B.

[0254] [Third Modification of Single-Phase Current Resonance DC / DC Converter]

[0255] Figure 15 A single-phase current resonance type DC / DC converter 1C is shown as a third modified example of the first embodiment. The single-phase current resonance type DC / DC converter 1C is composed of a main circuit unit 10C and a control unit 20C. The main circuit unit 10C includes a transformer circuit Tr, a primary-side switching circuit 11, a primary-side resonant circuit 12, a secondary-side rectifier circuit 13C, and terminals T1 to T4.

[0256] The single-phase current resonant DC / DC converter 1C is configured to perform unidirectional power transmission from terminals T1 and T2 (V1) to terminals T3 and T4 (V2). The single-phase current resonant DC / DC converter 1C differs from the first embodiment in that it does not include the secondary-side resonant circuit 14, but includes a secondary-side rectifier circuit 13C in place of the secondary-side rectifier circuit 13, and a control unit 20C in place of the control unit 20. All other aspects are the same.

[0257] The secondary-side rectifier circuit 13C has the same configuration as the secondary-side rectifier circuit 13 of the first embodiment, except that the diodes D5 to D8 are not connected to the current paths of the switching elements Q5 to Q8 .

[0258] The control unit 20C has the same configuration as the control unit 20 of the first embodiment, except that it performs frequency PWM conversion control instead of frequency phase shift conversion control. The control unit 20C performs frequency modulation control, frequency PWM conversion control, and frequency boost conversion control.

[0259] Figure 16 FIG. 1 shows the driving waveforms of the switching elements Q1 to Q8 during frequency PWM conversion control. Figure 17 (A) shows Figure 16 The current flowing in the main circuit unit 10C during Mode 1' is Figure 17 (B) shows Figure 16 The current of the main circuit unit 10C during the mode 2' flows. Figure 18 (A) shows Figure 16 The current flowing in the main circuit unit 10C during mode 3' is Figure 18 (B) shows Figure 16 The current flows in the main circuit unit 10C during the mode 4'.

[0260] like Figure 16 As shown, the driving waveforms of the switching elements Q1 and Q2 of the first branch and the driving waveforms of the switching elements Q3 and Q4 of the second branch are in the same phase (the phase difference is zero), except for the period To. Figure 5 The drive waveforms used during frequency modulation control in the first embodiment shown are common. The drive waveforms for switching elements Q5 and Q6 in the third branch and the drive waveforms for switching elements Q7 and Q8 in the fourth branch have the same phase (a phase difference of zero), but the timing of the transition from low to high (turn-on timing) lags behind the drive waveforms for switching elements Q1 to Q4 by an amount corresponding to the phase difference θ'.

[0261] Similar to the phase shift amount θ in the first embodiment, the control unit 20C during frequency PWM conversion control calculates the phase difference θ' using the driving frequency f, thereby controlling the pulse width of the driving waveform (PWM signal) of the switching elements Q5 to Q8. The driving frequency f is obtained by Figure 3 The phase difference θ' is determined by the same process as step S4. For example, the phase difference θ' is the value obtained by multiplying the difference between the driving frequency f and the first frequency (the frequency of the starting frequency PWM conversion control) by a predetermined first gain (but the value is different from the first gain Kθ). The phase difference θ' increases uniformly with the increase of the driving frequency f, so the pulse width decreases uniformly with the increase of the driving frequency f. Therefore, the period Tθ' of the switching elements Q1 to Q4 is Figure 5 Furthermore, when the driving frequency f exceeds the predetermined maximum PWM driving frequency, the phase difference θ' may also be fixed to the predetermined maximum phase difference.

[0262] During Mode 1', if Figure 17 As shown in (A), battery voltage V1 is applied between terminals T1 and T2, and a resonant current flows through the resonant coil Lr1 and resonant capacitor Cr1 via switching elements Q1 and Q4. As a result, on the rectifier side, a load current flows between terminals T3 and T4 via switching elements Q5 and Q8. During the subsequent Mode 2', in the primary-side switching circuit 11, a voltage opposite to that in Mode 1' is applied to the resonant coil Lr1, causing the excitation current to decay rapidly, and a current opposite to that in Mode 1' flows. However, in the secondary-side rectifier circuit 13C, since switching elements Q5 to Q8 are off, Figure 17 As shown in (B), no current flows on the load side.

[0263] During Mode 3', if Figure 18 As shown in (A), battery voltage V1 is applied between terminals T1 and T2, and a resonant current in the opposite direction to that of mode 1' flows through the resonant coil Lr1 and resonant capacitor Cr1 via switching elements Q2 and Q3. As a result, on the rectifier side, a load current flows between terminals T3 and T4 via switching elements Q6 and Q7. During the following mode 4', in the primary-side switching circuit 11, a voltage in the opposite direction to that of mode 3' is applied to the resonant coil Lr1, causing the excitation current to decay rapidly, and a current in the opposite direction to that of mode 3' flows. However, in the secondary-side rectifier circuit 13C, since switching elements Q5 to Q8 are off, Figure 18 As shown in (B), no current flows on the load side.

[0264] As described above, when the drive frequency f exceeds a preset first frequency in order to reduce output, the single-phase current resonant DC / DC converter 1C performs frequency PWM conversion control, controlling the pulse widths of switching elements Q5-Q8 using a phase difference θ' calculated using the drive frequency f. During Modes 2' and 4' of frequency PWM conversion control, the load current on the rectifier side is limited, and the main circuit unit 10C performs an output suppression operation (voltage reduction operation).

[0265] The phase difference θ' during frequency PWM conversion control is a controlled variable related to the drive frequency f. Therefore, the single-phase current resonant DC / DC converter 1C does not require switching to another control mode for control, nor does it require countermeasures for output fluctuations associated with switching control modes. Control can be performed solely based on frequency control. Consequently, the single-phase current resonant DC / DC converter 1C supports a wide range of output voltages without adding additional components or circuits, or complicating the control software structure.

[0266] Furthermore, when performing frequency boost conversion control, in Mode 1B of the first embodiment, while switching elements Q7 and Q8 are off, switching element Q6 is turned on, and secondary-side rectifier circuit 13 is short-circuited by conduction of diode D8. In Mode 3B of the first embodiment, while switching elements Q7 and Q8 are off, switching element Q5 is turned on, and secondary-side rectifier circuit 13 is short-circuited by conduction of diode D7, thereby performing boost operation. In contrast, in the third modified example, secondary-side rectifier circuit 13C is not provided with diodes D7 and D8. Therefore, for example, by turning on switching element Q8 in synchronization with switching element Q1 during a period corresponding to Mode 1B of the first embodiment and turning on switching element Q7 in synchronization with switching element Q3 during a period corresponding to Mode 3B of the first embodiment, similar current flows can be generated to perform boost operation.

[0267] [Other Modifications of Single-Phase Current Resonance DC / DC Converter]

[0268] While the embodiment and its modified examples of the single-phase current resonance type DC / DC converter of the present invention have been described above, the present invention is not limited to the embodiment and its modified examples.

[0269] For example, the single-phase current resonant DC / DC converter 1 of the first embodiment performs both frequency phase shift conversion control and frequency boost conversion control, but it may also perform only one of these. The same applies to the single-phase current resonant DC / DC converters 1A and 1B of the first and second modified examples. Furthermore, the single-phase current resonant DC / DC converter 1C of the third modified example performs both frequency PWM conversion control and frequency boost conversion control, but it may also perform only one of these.

[0270] As output suppression conversion control related to output suppression operation (step-down operation), controls other than frequency phase shift conversion control and frequency PWM conversion control can be performed as long as the control variable calculated using the drive frequency f is used. For example, frequency intermittent conversion control that causes the switching element to perform intermittent operation (burst operation) can be performed. In frequency intermittent conversion control, it is preferable to use the standby period during which the switching element is not turned on or off as the control variable, and calculate this standby period using the drive frequency f and a predetermined gain.

[0271] Similarly, as the output increasing conversion control related to the output increasing operation (boosting operation), as long as the control amount calculated using the driving frequency f is used, control other than the frequency increasing conversion control may be performed.

[0272] Regarding frequency phase-shift conversion control, in the above embodiment, the driving-side circuit (primary-side switching circuit 11 during discharge operation) is phase-shifted. However, this is not limited to this method, as long as phase-shift control is performed using a control variable calculated using the driving frequency f. Furthermore, regarding frequency boost conversion control, in the above embodiment, boost operation is performed by short-circuiting the secondary-side rectifier circuit. However, this is not limited to this method, as long as frequency boost conversion control is performed using a control variable calculated using the driving frequency f. For example, a boost circuit may be added to the primary or secondary-side circuit, and the additional boost circuit may be short-circuited. Furthermore, regarding frequency PWM conversion control, in the third variant, PWM control is performed using the secondary-side rectifier circuit 13C. However, PWM control may also be performed on the primary-side switching circuit 11 to control the resonant current period.

[0273] Regarding the structure of the main circuit of the present invention, the above-described embodiments and variations illustrate examples of an LLC or CLLC method with two parallel branches in a full-bridge configuration. However, this is not limiting. The configuration can be modified as appropriate, as long as the output suppression operation (step-down operation) or output boosting operation (step-up operation) is performed using the control variable calculated using the drive frequency f. In the first variation, the third branch is configured as a switching element, and the fourth branch is configured as a diode. However, the fourth branch can also be configured as a switching element, and the third branch can be configured as a diode. In the second variation, the lower branches of the third and fourth branches are configured as switching elements, and the upper branches of the third and fourth branches are configured as diodes. However, the upper branches of the third and fourth branches can also be configured as switching elements, and the lower branches of the third and fourth branches can also be configured as diodes. Furthermore, the LLC or CLLC method can be configured as a single half-bridge configuration, a three-phase LLC or three-phase CLLC method with three parallel branches, or a multi-parallel multi-phase LLC or multi-phase CLLC method with four or more parallel branches.

[0274] In the CLLC method with a half-bridge structure, for example, the pulse width of the PWM signal's on-time (the period during which the resonant current flows) is used as the control variable to perform frequency PWM conversion control on the driver-side switching element, while the on-time of the drive signal (the short-circuit time during which the load current flows), i.e., the boost amount, is used as the control variable to perform frequency boost conversion control on the rectifier-side switching element. Each control variable is calculated using the drive frequency f.

[0275] In the case of a three-phase CLLC system, for example, the pulse width of the PWM signal's on-time (the period during which the resonant currents of each phase, shifted by 120 degrees), can be used as the control variable to perform frequency PWM conversion control on the driver-side switching elements. Alternatively, the phase difference (phase shift amount) between the other two phase branches relative to the one-phase branch can be used as the control variable to perform frequency phase shift conversion control on the driver-side switching elements. Furthermore, the on-time of the drive signal (the short-circuit time during which the load currents of each phase flow), i.e., the boost amount, can be used as the control variable to perform frequency boost conversion control on the rectifier-side switching elements. Each control variable is calculated using the drive frequency f.

[0276] In the multi-phase CLLC method, the pulse width of the PWM signal's on-time (the period during which the resonant current of each phase flows) is used as the control variable to perform frequency PWM conversion control on the driver-side switching elements, while the on-time of the drive signal (the short-circuit time during which the load current of each phase flows), i.e., the boost amount, is used as the control variable to perform frequency boost conversion control on the rectifier-side switching elements. Each control variable is calculated using the drive frequency f.

[0277] [Embodiment of a Three-Phase Current Resonance DC / DC Converter (Second Embodiment)]

[0278] Figure 19 A three-phase current resonance type DC / DC converter 100A (hereinafter referred to as DC / DC converter 100A) according to a second embodiment of the present invention is shown. DC / DC converter 100A is a three-phase current resonance type DC / DC converter using a CLLC method, and performs forward power transmission from the primary-side circuit to the secondary-side circuit and reverse power transmission from the secondary-side circuit to the primary-side circuit.

[0279] The DC / DC converter 100A includes terminals T1 to T4. The DC-side terminals of a bidirectional inverter, which converts AC voltage into DC voltage, are connected to terminals T3 and T4. A load (e.g., an electric vehicle battery) is connected to terminals T1 and T2. For example, during forward power transmission from terminals T1 and T2 to terminals T3 and T4, a DC voltage V1 is input to terminals T1 and T2, and a DC voltage V2 and a DC current to be supplied to the load are output from terminals T3 and T4. When the electric vehicle battery is connected to terminals T1 and T2, the voltage of V1 varies from 150V to 450V. Furthermore, when terminals T3 and T4 are at the bus voltage, V2 is a constant voltage of approximately 380V. Therefore, as voltage V1 decreases, the input-output voltage ratio increases. Therefore, to increase the voltage gain (step-up ratio) of the LLC method, the driving frequency of the primary-side switching circuit 111 is reduced.

[0280] The DC / DC converter 100A includes a transformer unit 110; a primary-side circuit, including the primary-side switching circuit 111, the primary-side resonant circuit 112, and the capacitor Co1; a secondary-side circuit, a control unit 115A; and drive circuits (not shown) for the primary-side switching circuit 111 and the secondary-side switching circuit 113; and a detection circuit (not shown). The detection circuit detects voltage and current values required for control by the control unit 115A and is comprised of, for example, various sensors and an A / D converter.

[0281] The transformer unit 110 (corresponding to the "transformer unit" of the present invention) is constructed from a high-frequency insulation transformer and includes a first-phase transformer circuit Tr1, a second-phase transformer circuit Tr2, and a third-phase transformer circuit Tr3. Each phase transformer circuit has a primary-side coil (winding) and a secondary-side coil (winding). The primary-side coils of the transformer circuits Tr1-Tr3 are connected to a primary-side switching circuit 111 via a primary-side resonant circuit 112, while the secondary-side coils of the transformer circuits Tr1-Tr3 are connected to a secondary-side switching circuit 113 via a secondary-side resonant circuit 114. Each phase transformer circuit Tr1-Tr3 may be formed from one or more transformers, or the transformer unit 110 may be formed from a single three-phase transformer or multiple transformers having multiple phases. Furthermore, the primary-side excitation coils of the transformer circuits Tr1-Tr3 are included in the primary-side coils of each transformer circuit, while the secondary-side excitation coils of the transformer circuits Tr1-Tr3 are included in the secondary-side coils of each transformer circuit; illustration of the excitation coils is omitted.

[0282] Primary-side switching circuit 111 includes a first branch for the first phase, a second branch for the second phase, and a third branch for the third phase, connected in parallel. Each branch includes an upper branch and a lower branch connected in series. The upper branch of the first branch includes a switching element Q1, a diode D1, and a capacitor C1. Similarly, the lower branch of the first branch includes a switching element Q2, a diode D2, and a capacitor C2. The upper branch of the second branch includes a switching element Q3, a diode D3, and a capacitor C3. The lower branch of the second branch includes a switching element Q4, a diode D4, and a capacitor C4. The upper branch of the third branch includes a switching element Q5, a diode D5, and a capacitor C5. The lower branch of the third branch includes a switching element Q6, a diode D6, and a capacitor C6. Diodes D1-D6 are reverse-connected diodes for return current during forward power transmission and for rectification during reverse power transmission. Capacitors C1-C6 are partial resonant capacitors for soft switching.

[0283] As switching element Q1, for example, an IGBT (insulated gate bipolar transistor), a MOSFET (metal oxide semiconductor field effect transistor) using SiC (silicon carbide) or GaN (gallium nitride), or other power semiconductor switching element capable of switching at high frequencies can be used. The same applies to switching elements Q2 to Q6 and switching elements Q7 to Q12 described later.

[0284] Diode D1 is connected in reverse parallel to the current path of switching element Q1. Diode D1 can be a parasitic diode of switching element Q1, an external diode independent of switching element Q1, or both. The same applies to diodes D2-D6 and diodes D7-D12 described later.

[0285] Capacitor C1 is connected in parallel to the current path of switching element Q1 and diode D1. Capacitor C1 can be the parasitic capacitance of switching element Q1, an external capacitor independent of switching element Q1, or both. The same applies to capacitors C2-C6 and capacitors C7-C12 described later.

[0286] The primary-side resonant circuit 112 includes a first resonant circuit comprising a resonant coil Lr1 and a resonant capacitor Cr1; a second resonant circuit comprising a resonant coil Lr2 and a resonant capacitor Cr2; and a third resonant circuit comprising a resonant coil Lr3 and a resonant capacitor Cr3. Resonant coil Lr1 has one end connected to the connection point X1 between switching elements Q1 and Q2 in the first arm, and its other end connected to one end of resonant capacitor Cr1 via the primary-side coil of transformer circuit Tr1. Resonant coil Lr2 has one end connected to the connection point X2 between switching elements Q3 and Q4 in the second arm, and its other end connected to one end of resonant capacitor Cr2 via the primary-side coil of transformer circuit Tr2. Resonant coil Lr3 has one end connected to the connection point X3 between switching elements Q5 and Q6 in the third arm, and its other end connected to one end of resonant capacitor Cr3 via the primary-side coil of transformer circuit Tr3. The other ends of resonant capacitors Cr1 to Cr3 are connected to form a neutral point Y1. Furthermore, the "resonant coil" of the present invention may be a coil independent of the "transformer unit" of the present invention, may be formed from the leakage inductance of the transformer unit, or may be a combination of both. Specifically, the resonant coil Lr1 may utilize the primary-side leakage inductance of the transformer circuit Tr1 of the transformer unit 110, may utilize a reactor separate from the transformer circuit Tr1, or may utilize a combination of both. The same applies to the resonant coils Lr2 and Lr3.

[0287] The resonant coil Lr1 and resonant capacitor Cr1 can be swapped, or both can be placed on one or the other end of the primary coil of the transformer circuit Tr1. The same applies to the resonant coil Lr2 and resonant capacitor Cr2, and the resonant coil Lr3 and resonant capacitor Cr3. Furthermore, while the primary side of the transformer unit 110 uses a Y connection, a Δ connection is also acceptable as long as it is a balanced connection. The secondary side also uses a Y connection, but a Δ connection is also acceptable.

[0288] Capacitor Co1 is a capacitor for removing output ripple during reverse power transmission and also for removing input voltage ripple during forward power transmission. The capacitor is connected between terminals T1 and T2.

[0289] Secondary-side switching circuit 113 has the same structure as primary-side switching circuit 111, comprising a fourth branch for the first phase, a fifth branch for the second phase, and a sixth branch for the third phase, connected in parallel. Each branch comprises an upper branch and a lower branch connected in series. The upper branch of each branch includes switching elements Q7, Q9, and Q11, diodes D7, D9, and D11, and capacitors C7, C9, and C11. Similarly, the lower branch of each branch includes switching elements Q8, Q10, and Q12, diodes D8, D10, and D12, and capacitors C8, C10, and C12. Diodes D7-D12 are reverse-connected diodes for return current during reverse power transmission and for rectification during forward power transmission. Capacitors C7-C12 are partial resonant capacitors for soft switching.

[0290] The secondary-side resonant circuit 114 has the same structure as the primary-side resonant circuit 112, comprising a fourth resonant circuit comprising a resonant coil Lr4 and a resonant capacitor Cr4, a fifth resonant circuit comprising a resonant coil Lr5 and a resonant capacitor Cr5, and a sixth resonant circuit comprising a resonant coil Lr6 and a resonant capacitor Cr6. One end of the resonant coil Lr4 is connected to the connection point X4 between the switching elements Q7 and Q8 of the fourth arm, and the other end is connected to one end of the resonant capacitor Cr4 via the secondary coil of the transformer circuit Tr1. One end of the resonant coil Lr5 is connected to the connection point X5 between the switching elements Q9 and Q10 of the fifth arm, and the other end is connected to one end of the resonant capacitor Cr5 via the secondary coil of the transformer circuit Tr2. One end of the resonant coil Lr6 is connected to the connection point X6 between the switching elements Q11 and Q12 of the sixth arm, and the other end is connected to one end of the resonant capacitor Cr6 via the secondary coil of the transformer circuit Tr3. The other ends of the resonant capacitors Cr4 to Cr6 are connected to form a neutral point Y2. The resonance coil Lr4 may utilize the secondary-side leakage inductance of the transformer circuit Tr1, or may use a reactor different from the transformer circuit Tr1, or may use a combination of the two. The same applies to the resonance coils Lr5 and Lr6.

[0291] Capacitor Co2 is a capacitor for removing output ripple during forward power transmission and also for removing input voltage ripple during reverse power transmission. It is connected between terminals T3 and T4.

[0292] Control unit 115A includes a processing unit that performs various controls for turning on and off switching elements Q1 to Q12, and a storage unit such as a memory. Control unit 115A may be composed of a digital circuit using a microcontroller or DSP, an analog circuit, or a combination of digital and analog circuits.

[0293] During forward power transmission, when the driving frequency of switching elements Q1-Q6 of primary-side switching circuit 111 exceeds a predetermined frequency (equivalent to the "first frequency" of this invention), control unit 115A performs frequency modulation control on primary-side switching circuit 111 and diode rectification control or synchronous rectification control on secondary-side switching circuit 113. When the driving frequency of primary-side switching circuit 111 is below the first frequency, control unit 115A performs boost control (referred to as "frequency boost conversion control" in this embodiment).

[0294] The first frequency is a driving frequency that serves as a threshold for starting frequency step-up conversion control when the input voltage V1 of the primary-side switching circuit 111 is low (the input-to-output voltage ratio is high) and the target output voltage cannot be achieved within the normally used driving frequency range even when frequency modulation control is performed to reduce the driving frequency. The normally used driving frequency range refers to a frequency range greater than the resonant frequency determined by the inductance of the primary-side excitation coil and resonant coils Lr1 to Lr3, and the capacitance of resonant capacitors Cr1 to Cr3, of the transformer circuits Tr1 to Tr3.

[0295] Frequency-boost conversion control calculates the boost amount, a control variable, based on the drive frequency of switching elements Q1-Q6 (the driver side) during forward power transmission. Boost control is initiated in sync with the conduction timing of switching elements Q1-Q6 in primary-side switching circuit 111 during the period when a resonant current flows through primary-side resonant circuit 112. Secondary-side switching circuit 113 is short-circuited only during the boost period corresponding to the boost amount. Frequency-boost conversion control allows a large resonant current to flow through primary-side resonant circuit 112 during the boost period, accumulating a large amount of energy in resonant coils Lr1-Lr3. This energy is then discharged as load current to secondary-side switching circuit 113, thereby achieving a high output voltage V2 even when input voltage V1 is low.

[0296] During reverse power transmission, when the driving frequency of switching elements Q7-Q12 of secondary-side switching circuit 113 is higher than a predetermined frequency (equivalent to the "second frequency" in this invention), control unit 115A performs frequency modulation control on secondary-side switching circuit 113 and diode rectification control or synchronous rectification control on primary-side switching circuit 111. On the other hand, when the driving frequency of secondary-side switching circuit 113 is lower than the second frequency, control unit 115A performs boost control (referred to as "frequency boost conversion control" in this embodiment).

[0297] Regarding the frequency boost conversion control during reverse power transmission, as in the forward power transmission, the boost amount as the control amount is calculated based on the driving frequency of the driving side, i.e., the switching elements Q7 to Q12, during the reverse power transmission, and during the period in which the resonant current flows through the secondary-side resonant circuit 114, the primary-side switching circuit 111 is short-circuited only during the boost period corresponding to the boost amount, in coordination with the conduction timing of the switching elements Q7 to Q12 of the secondary-side switching circuit 113.

[0298] In addition, the second frequency is a driving frequency that serves as a threshold for starting frequency boost conversion control when the output voltage V1 of the primary-side switching circuit 111 is high (the input-output voltage ratio is high) and the target output voltage cannot be reached within the range of the normally used driving frequency even if frequency modulation control is performed to reduce the driving frequency.

[0299] In DC / DC converter 100A, the circuit configurations of the primary and secondary sides are symmetrical, with transformer unit 110 interposed between them. Therefore, control unit 115A controls secondary-side switching circuit 113 during reverse power transmission in the same manner as it controls primary-side switching circuit 111 during forward power transmission. And, in reverse power transmission, control primary-side switching circuit 111 during forward power transmission in the same manner as secondary-side switching circuit 113 during forward power transmission. The specific control details are common during both forward and reverse power transmission, so the following description focuses solely on forward power transmission.

[0300] Figure 20 exemplifies the relationship between the driving frequency f [kHz] of the primary-side switching circuit 111 during forward power transmission and the boost amount ψ [°], which is the control amount of the frequency boost conversion control. Figure 20 The figure shows the boost amount ψ when the input voltage V1 is V11 (for example, V11 = 300 V) and the boost amount ψ when the input voltage V1 is V12 (for example, V12 = 150 V). The boost amount ψ is represented by the value [°] during which any of the switching elements Q7 to Q12 of the secondary-side switching circuit 113 is turned on, short-circuiting the secondary-side switching circuit 113, while the resonant current flows through the primary-side resonant circuit 112, assuming a period T of the drive frequency f of 360°. When the input voltage V1 is relatively low (V12), a larger boost ratio is required, so the boost amount ψ is larger. When the input voltage V1 is relatively high (V11), a smaller boost ratio is sufficient, so the boost amount ψ is smaller.

[0301] When the driving frequency f is higher than the boost start driving frequency fψs [kHz] corresponding to the first frequency, the control unit 115A performs frequency modulation control on the primary side switching circuit 111. When the input voltage V1 is less than the predetermined threshold value (V1th [V]) and the driving frequency f is less than the boost start driving frequency fψs, even if the driving frequency f is reduced by frequency modulation control alone, the output value (for example, the output voltage V2) does not rise to the target value (for example, the target voltage value), so the control unit 115A performs frequency boost conversion control. The boost start driving frequency fψs can be calculated based on the winding ratio of the transformer unit 110, the input voltage V1 and the output voltage V2, and the resonance constant based on the resonant coils Lr1 to Lr3 and the resonant capacitors Cr1 to Cr3. Figure 20 , fψs = 90 [kHz]. Furthermore, when the input voltage V1 is V1th or higher, due to the transformation ratio of the transformer unit 110, even if the input voltage V1 is low (for example, even if V1 = V1th), the output value can be boosted to the target value through frequency modulation alone. Therefore, the drive frequency f does not fall below the boost start drive frequency fψs. Therefore, when the input voltage V1 is V1th or higher, the control unit 115A performs frequency modulation control on the primary-side switching circuit 111.

[0302] When the driving frequency f is higher than the maximum boost driving frequency fψh ( Figure 20 In the case of fψh=67[kHz]), the boost amount ψ increases as the drive frequency f decreases. When the drive frequency f is below the maximum boost drive frequency fψh, the boost amount ψ when the input voltage V1 is V11 (300[V]) is fixed (clamped) to the maximum boost amount ψh(V11), and the boost amount ψ when the input voltage V1 is V12 (150[V]) is fixed (clamped) to the maximum boost amount ψh(V12). Figure 20 , ψh(V11)=15°, ψh(V12)=45°.

[0303] Furthermore, the maximum boost amount ψh(V1x) is set to prevent a situation where, when short-circuiting secondary-side switching circuit 113 during boost control, a large amount of energy is accumulated in a very short period of time in resonant coils Lr1-Lr3. This could lead to excessive boost amount ψ, causing over-boosting and unstable control. By varying boost amount ψ in accordance with the value of input voltage V1, an optimal boost ratio can be achieved even at low input voltages, allowing the output voltage range to be appropriately expanded.

[0304] Figure 21 FIG. 1 shows a frequency-boost conversion control block 120 included in the control unit 115A. The frequency-boost conversion control block 120 includes a first adding unit 121 , a first multiplying unit 122 , and a first clamping unit 123 .

[0305] When the input voltage V1 is lower than the first input voltage (threshold V1th), the first adding unit 121 compares the driving frequency f and the boost start driving frequency fψs. When the driving frequency f is lower than the boost start driving frequency fψs, the first adding unit 121 outputs the difference (fψs-f) between the driving frequency f and the boost start driving frequency fψs to the first multiplying unit 122.

[0306] The first multiplication unit 122 multiplies the difference (fψs-f) by the frequency boost conversion gain Kψ(V1x) to calculate the boost amount ψ, and outputs the boost amount ψ to the first clamp unit 123. The frequency boost conversion gain Kψ(V1x) is calculated based on the input voltage V1 at this time ( Figure 20 In addition, since a higher step-up ratio (larger boost amount ψ) is required when the input voltage V1 is low (the input-output voltage is relatively high), a larger value is preferably used for the frequency boost conversion gain Kψ(V1x). Figure 20 In the example, Kψ(V11)<Kψ(V12).

[0307] The first clamping unit 123 sets the lower limit of the boost amount ψ to 0° and the upper limit to the maximum boost amount ψh(V1x), and outputs the boost amount ψ (ψ=(fψs-f)×Kψ(V1x)≤ψh(V1x)). The maximum boost amount ψh(V1x) is determined by the input voltage V1 at this time (at Figure 20 In the equation ψh(V1x)=ψh(V11) or ψh(V12)). In addition, since a higher step-up ratio (larger boost amount ψ) is required when the input voltage V1 is low (the input-output voltage is relatively high), a larger value is preferably used as the maximum boost amount ψh(V1x). Figure 20 In the example, ψh(V11)<ψh(V12).

[0308] The control unit 115A performs frequency boost conversion control based on the boost amount ψ output from the first clamping unit 123. When the control unit 115A increases the output, since the drive frequency f is the control amount, the control unit 115A controls so that the drive frequency f decreases. Therefore, the difference (fψs-f) output from the first adding unit 121 increases. Before the boost amount ψ reaches the maximum boost amount ψh (V1x), the boost amount ψ increases evenly as the drive frequency f decreases, so the output of the DC / DC converter 100A increases. In addition, Figure 20 and Figure 21In the frequency boost conversion gain Kψ(V1x) and the maximum boost amount ψh(V1x) are changed according to the input voltage V1, but the boost start drive frequency fψs can also be changed according to the input voltage V1. For example, when the input voltage V1 is low, the boost start drive frequency fψs can be set high, and the frequency boost conversion control can be started from a higher drive frequency f. Figure 20 In the example, fψs(V11) < fψs(V12). In this way, a high boost amount ψ can be obtained without using a very high frequency boost conversion gain Kψ(V1x), enabling more stable boost control.

[0309] In addition, the output voltage V2 of the secondary side switching circuit 113 is set to be constant, and the frequency boost conversion gain Kψ(V1x), the maximum boost amount ψh(V1x), and the boost start drive frequency fψs are changed according to the change of the input voltage V1. However, when the input voltage V1 remains unchanged and the output voltage V2 changes, the above values can also be changed according to the change of the output voltage V2. When both or one of the input voltage V1 and the output voltage V2 changes, the above values can also be changed according to the value of the input voltage ratio V2 / V1.

[0310] Figure 22 An example of a flowchart of a control process executed by the control section 115A at the time of forward power transmission is shown.

[0311] The control unit 115A, which has started the control process, first starts frequency modulation control. The control unit 115A obtains input and output information of the DC / DC converter 100A (e.g., input voltage V1, input current to terminal T1, output voltage V2, output current from terminal T3) from the detection circuit (S101).

[0312] The control unit 115A, which has obtained the input and output information, compares the input voltage V1 with a predetermined threshold value V1th [V] to determine whether the input voltage V1 is less than the threshold value V1th (S102). When the input voltage V1 is less than the threshold value V1th (Yes in S102), the control unit 115A reads the control parameters stored in the storage unit (S103) and transfers to step S104. The control parameters include the control parameters of the frequency boost conversion control, namely the boost start drive frequency fψs, the frequency boost conversion gain Kψ(V1x), the maximum boost amount ψh(V1x), etc. In addition, the fixed parameters among the control parameters can be read only once at startup. When the input voltage V1 is above the threshold value V1th (No in S102), the control unit 115A transfers to step S104. In addition, step S102 is provided to speed up the processing and is not essential.

[0313] Shifting to step S104, control unit 115A compares the output value (e.g., output power value) of DC / DC converter 100A with a target value (e.g., target power value) and determines the drive frequency f of switching elements Q1-Q6 based on the difference between the two. Control unit 115A decreases drive frequency f when the output value is less than the target value, and increases drive frequency f when the output value is greater than the target value. The target value can be, for example, a value preset by control unit 115A, a value determined based on the input and output conditions of DC / DC converter 100A, or a value input from an external device. The target value can be variable at any time.

[0314] Next, control unit 115A compares the drive frequency f determined in step S104 with the boost start drive frequency fψs to determine whether the drive frequency f is less than or equal to the boost start drive frequency fψs (S105). If the drive frequency f is greater than the boost start drive frequency fψs (No in S105), control unit 115A sets the drive frequency f to the value determined in step S104, drives switching elements Q1-Q6 at the drive frequency f, and performs diode rectification or synchronous rectification control on secondary-side switching circuit 113 (S106), then proceeding to step S111.

[0315] If the drive frequency f is equal to or lower than the boost start drive frequency fψs (YES in S105 ), the control unit 115A multiplies the difference (fψs-f) between the drive frequency f and the boost start drive frequency fψs by the frequency boost conversion gain Kψ(V1x) to calculate the boost amount ψ ( S107 ). The processing in step S107 is performed by the first adding unit 121 and the first multiplying unit 122 .

[0316] After calculating the supercharging amount ψ, the control unit 115A compares the supercharging amount ψ with the maximum supercharging amount ψh(V1x) to determine whether the supercharging amount ψ is greater than or equal to the maximum supercharging amount ψh(V1x) (S108). If the supercharging amount ψ is greater than or equal to the maximum supercharging amount ψh(V1x) (YES in S108), the control unit 115A sets the supercharging amount ψ to the maximum supercharging amount ψh(V1x) (S109) and proceeds to step S110. On the other hand, if the supercharging amount ψ is less than the maximum supercharging amount ψh(V1x) (NO in S108), the supercharging amount ψ is set to the value calculated in step S107 and the process proceeds to step S110.

[0317] The control unit 115A proceeds to step S110 , sets the driving frequency f to the value determined in step S104 , drives the switching elements Q1 to Q6 at the driving frequency f, and performs frequency boost conversion control based on the set boost amount ψ ( S110 ).

[0318] Next, the control unit 115A determines whether the output value has reached the target value (S111). When the output value has not reached the target value (No in S111), the control unit 115A repeats the processing after step S101. When the output value has reached the target value (Yes in S111), it transfers to step S112. In step S112, the control unit 115A determines whether to continue the control process. For example, the control unit 115A makes the determination of step S112 based on whether an end instruction is received from an external device. For example, when the end instruction is not received, the control unit 115A determines to continue the control process (Yes in S112), repeats the processing after step S101, and when the end instruction is received, determines not to continue the control process (No in S112), and ends the control process.

[0319] Figure 23 1 shows the control timing of the DC / DC converter 100A when the drive frequency f is higher than the boost start drive frequency fψs. Figure 23 In the embodiment, the control unit 115A performs frequency modulation control on the switching elements Q1 to Q6 and performs diode rectification control on the switching elements Q7 to Q12. 1 to 6 in the figure represent the control timing patterns, and patterns 1 to 6 are repeated when performing this control. Figure 23 The dead time is omitted in the figure and will also be omitted in subsequent drawings.

[0320] During frequency modulation control, control unit 115A sets a predetermined dead time and alternately turns switching elements Q1 and Q2 on and off with a phase difference of 180°. It also sets a predetermined dead time and alternately turns switching elements Q3 and Q4 on and off with a phase difference of 180°. It also sets a predetermined dead time and alternately turns switching elements Q5 and Q6 on and off with a phase difference of 180°. The duty ratio of switching elements Q1-Q6 is set to 50%, for example. Strictly speaking, the on-duty ratio of switching elements Q1-Q6 is 50% minus the amount corresponding to the dead time. Furthermore, while the phase difference between the first and second branches is fixed at 120°, the phase difference between the second and third branches is fixed at 120°, and the phase difference between the third branch and the first branch is fixed at 120°, the control unit 115A sets the driving frequency of each switching element Q1 to Q6 to the same driving frequency f, and simultaneously changes the same driving frequency f to the same degree, thereby bringing the output value (e.g., output power value) of the DC / DC converter 100A closer to a predetermined target value (e.g., target power value). The control unit 115A decreases the driving frequency f when the output value is less than the target value, and increases the driving frequency f when the output value is greater than the target value.

[0321] During diode rectification control, the control unit 115A turns the switching elements Q7 to Q12 to the continuously off state, thereby forming the secondary-side switching circuit 113 into a diode bridge circuit of diodes D7 to D12 and performing diode rectification using this diode bridge circuit.

[0322] Figures 24 to 26 Show Figure 23 The main current path diagram during modes 1 to 6. Figures 24 to 26 In the figure, the solid line represents the resonant current flowing through the primary side circuit and the load current flowing through the secondary side circuit, and the dotted line represents the excitation current flowing through the primary side circuit. The same is true in the following figures. Figures 24 to 26 In FIG. 1 , description of the currents when the switching elements Q1 to Q6 are turned on and off (currents during transition) is omitted, and description of the currents during transition is also omitted in subsequent drawings.

[0323] like Figure 24 As shown in (A), during Mode 1, when switching element Q1 is on, a resonant current flows through the first resonant circuit comprising resonant coil Lr1 and resonant capacitor Cr1, and the second resonant circuit comprising resonant coil Lr2 and resonant capacitor Cr2, and a load current flows through the secondary-side circuit via transformer circuits Tr1 and Tr2. Since only the excitation current flows through the third resonant circuit comprising resonant coil Lr3 and resonant capacitor Cr3, no load current flows through the secondary-side coil of transformer circuit Tr3.

[0324] Figure 24 (B) shows the second half of Mode 2. During the second half of Mode 2, only the excitation current flows in the primary circuit (the resonant current flowing during Mode 1 has converged), so no load current flows in the secondary circuit. Furthermore, the resonant period during which the resonant current flows varies depending on the drive frequency f, the input and output conditions of the DC / DC converter 100A, and the load conditions.

[0325] like Figure 25 As shown in (A), during Mode 3, when switching element Q3 is on, resonant current flows through the second and third resonant circuits, and load current flows through the secondary-side circuit via transformer circuits Tr2 and Tr3. Since only the excitation current flows in the first resonant circuit, no load current flows through the secondary-side winding of transformer circuit Tr1.

[0326] Figure 25 (B) shows the second half of Mode 4. In the second half of Mode 4, only the excitation current flows in the primary circuit (the resonant current that flowed during Mode 3 has converged), so no load current flows in the secondary circuit.

[0327] like Figure 26As shown in (A), during Mode 5, when switching element Q5 is on, a resonant current flows through the third resonant circuit and the first resonant circuit, and a load current flows through the secondary-side circuit via transformer circuits Tr3 and Tr1. Since only the excitation current flows in the second resonant circuit, no load current flows through the secondary-side winding of transformer circuit Tr2.

[0328] Figure 26 (B) shows the second half of mode 6. In the second half of mode 6, only the excitation current flows in the primary circuit (the resonant current flowing during mode 5 has converged), so no load current flows in the secondary circuit.

[0329] Figure 27 The control timing of the DC / DC converter 100A is shown when the driving frequency f is equal to or lower than the boost start driving frequency fψs. Figure 27 In the process, the control unit 115A performs diode rectification control and frequency boost conversion control. Figure 23 The difference is that there is a supercharging period ψ in Mode 1-1, Mode 3-1, and Mode 5-1. The supercharging period ψ is a period corresponding to the supercharging amount ψ.

[0330] The control unit 115A during diode rectification control and frequency boost conversion control turns on the switching element Q8 of the fourth branch at the start of the resonance period when the resonant current flows through the first resonant circuit and the second resonant circuit (the turn-on timing of the switching element Q1), thereby generating the boost period ψ of mode 1-1. At the start of the resonance period when the resonant current flows through the second resonant circuit and the third resonant circuit (the turn-on timing of the switching element Q3), the switching element Q10 of the fifth branch turns on at the start of the resonance period when the resonant current flows through the second resonant circuit and the third resonant circuit, thereby generating the boost period ψ of mode 3-1. At the start of the resonance period when the resonant current flows through the third resonant circuit and the first resonant circuit (the turn-on timing of the switching element Q5), the switching element Q12 of the sixth branch turns on at the start of the resonance period when the resonant current flows through the third resonant circuit and the first resonant circuit, thereby generating the boost period ψ of mode 5-1.

[0331] During diode rectification control and frequency boost conversion control, control unit 115A turns off switching element Q8 at the end of boost period ψ in mode 1-1, turns off switching element Q10 at the end of boost period ψ in mode 3-1, and turns off switching element Q12 at the end of boost period ψ in mode 5-1. Switching elements Q7, Q9, and Q11 are continuously off.

[0332] Figures 28 to 30 Show Figure 27 Current path diagram during modes 1-1, 1, 3-1, 3, 5-1, 5. Figure 27 The current path during Mode 2, Mode 4, and Mode 6 is Figure 23The current paths during Mode 2, Mode 4, and Mode 6 are the same and are therefore omitted.

[0333] like Figure 28 As shown in (A), during mode 1-1, when the switching element Q1 is turned on, a resonant current flows through the first and second resonant circuits, and the load current flows through the secondary circuit via the transformer circuits Tr1 and Tr2. In the secondary circuit, since the switching element Q8 is turned on, the secondary switch circuit 113 is short-circuited through the path of the switching element Q8 and the diode D10. As a result, a resonant current larger than the normal resonant current flows through the first and second resonant circuits, and a large amount of energy is stored in the resonant coils Lr1 and Lr2. Figure 28 As shown in (B), during Mode 1, switching element Q8 is turned off, releasing the energy accumulated during Mode 1-1, causing a load current greater than normal to flow through the secondary circuit. Thus, DC / DC converter 100A performs a boost operation during Mode 1-1 and Mode 1.

[0334] like Figure 29 As shown in (A), during mode 3-1, when the switching element Q3 is turned on, a resonant current flows through the second and third resonant circuits, and the load current flows through the secondary circuit via the transformer circuits Tr2 and Tr3. In the secondary circuit, since the switching element Q10 is turned on, the secondary switch circuit 113 is short-circuited through the path of the switching element Q10 and the diode D12. As a result, a resonant current larger than the normal resonant current flows through the second and third resonant circuits, and a large amount of energy is stored in the resonant coils Lr2 and Lr3. Figure 29 As shown in (B), during Mode 3, switching element Q10 is turned off, releasing the energy stored during Mode 3-1, causing a load current greater than normal to flow through the secondary circuit. Thus, DC / DC converter 100A performs a boost operation during Modes 3-1 and 3.

[0335] like Figure 30 As shown in (A), during mode 5-1, when the switching element Q5 is turned on, a resonant current flows through the third resonant circuit and the first resonant circuit, and the load current flows through the secondary circuit via the transformer circuits Tr3 and Tr1. In the secondary circuit, since the switching element Q12 is turned on, the secondary switch circuit 113 is short-circuited through the path of the switching element Q12 and the diode D8. As a result, a resonant current larger than the normal resonant current flows through the third resonant circuit and the first resonant circuit, and a large amount of energy is stored in the resonant coils Lr3 and Lr1. Figure 30As shown in (B), during Mode 5, switching element Q12 is turned off, releasing the energy stored during Mode 5-1, causing a load current greater than normal to flow through the secondary circuit. Thus, DC / DC converter 100A performs a boost operation during Modes 5-1 and 5.

[0336] In this embodiment, control unit 115A synchronizes the on-time of switching element Q8 with the on-time of switching element Q1 to generate the boost period ψ of mode 1-1 at the start of the resonance period. However, the timing for determining the start of the resonance period is not limited to this. Control unit 115A may also detect the load current (current flowing through the fourth to sixth resonant circuits or the fourth to sixth branches) flowing through the secondary-side circuit (e.g., the secondary-side coils of transformer circuits Tr1 and Tr2) or the voltage of the upper or lower arms of each secondary-side branch, and turn on switching element Q8 in synchronization with the increase in this load current or voltage. Furthermore, during the resonance period, switching element Q8 can be turned on after a certain time has passed since switching element Q1 was turned on, taking into account the phase lag of the secondary-side current. However, if mode 1-1 follows mode 1, switching element Q8 is turned on after the load current increases, resulting in hard switching of the load current. Furthermore, during the reverse recovery period of diode D7, secondary-side switching circuit 113 is short-circuited, potentially generating noise. Therefore, it is preferable to generate boost period ψ at the start of the resonance period, as in this embodiment. The same applies to modes 3-1 and 5-1.

[0337] Figure 31 1 shows the control timing of the DC / DC converter 100A when the drive frequency f is higher than the boost start drive frequency fψs. Figure 31 In the process, the control unit 115A performs frequency modulation control on the switching elements Q1 to Q6 and performs synchronous rectification control on the switching elements Q7 to Q12. Figure 23 The FM control is the same.

[0338] During synchronous rectification control, control unit 115A turns on switching elements Q7 and Q10 when the load current begins to flow through the secondary-side circuit in response to the resonant current in mode 1, turns on switching elements Q9 and Q12 when the load current begins to flow through the secondary-side circuit in response to the resonant current in mode 3, and turns on switching elements Q8 and Q11 when the load current begins to flow through the secondary-side circuit in response to the resonant current in mode 5. For example, control unit 115A detects the load current start timing by detecting the load current flowing through each branch of secondary-side switching circuit 113. However, this is not limiting. Control unit 115A may also detect the timing by detecting the resonant current flowing through each branch of secondary-side switching circuit 113 or by detecting the voltage across the current path of switching elements Q7 to Q12.

[0339] During synchronous rectification control, control unit 115A turns off switching elements Q7 and Q10 when the load current converges to zero in mode 1, turns off switching elements Q9 and Q12 when the load current converges to zero in mode 3, and turns off switching elements Q8 and Q11 when the load current converges to zero in mode 5. Control unit 115A detects the timing when the load current converges to zero in the same manner as it detects the timing when the load current starts to flow.

[0340] Figures 32 to 34 Show Figure 31 The main current paths during modes 1 to 6.

[0341] like Figure 32 As shown in (A), during Mode 1, switching elements Q1, Q7, and Q10 are on, causing resonant currents to flow through the first and second resonant circuits, and load currents to flow through the secondary-side circuits of transformer circuits Tr1 and Tr2. Only the excitation current flows through the third resonant circuit, so no load current flows through the secondary-side winding of transformer circuit Tr3.

[0342] Figure 32 (B) shows the second half of Mode 2. During the second half of Mode 2, since switching elements Q7 and Q10 are turned off and synchronous rectification ends, only the excitation current flows in the primary circuit (the resonant current that flowed during Mode 1 converges), and therefore no load current flows in the secondary circuit.

[0343] like Figure 33 As shown in (A), during Mode 3, switching elements Q3, Q9, and Q12 are on, causing resonant current to flow through the second and third resonant circuits, and load current to flow through the secondary-side circuit via transformer circuits Tr2 and Tr3. Since only the excitation current flows through the first resonant circuit, no load current flows through the secondary-side coil of transformer circuit Tr1.

[0344] Figure 33 (B) shows the second half of Mode 4. During the second half of Mode 4, since switching elements Q9 and Q12 are turned off and synchronous rectification is completed, only the excitation current flows in the primary circuit (the resonant current that flowed during Mode 3 has converged), and therefore no load current flows in the secondary circuit.

[0345] like Figure 34 As shown in (A), during Mode 5, switching elements Q5, Q8, and Q11 are on, causing resonant current to flow through the third and first resonant circuits, and load current to flow through the secondary-side circuit via transformer circuits Tr3 and Tr1. Since only the excitation current flows through the second resonant circuit, no load current flows through the secondary-side coil of transformer circuit Tr2.

[0346] Figure 34 (B) shows the second half of Mode 6. During the second half of Mode 6, since switching elements Q8 and Q11 are turned off, synchronous rectification ends, and only the excitation current flows in the primary circuit (the resonant current that flowed during Mode 5 converges). Therefore, no load current flows in the secondary circuit.

[0347] Figure 35 The control timing of the DC / DC converter 100A is shown when the driving frequency f is equal to or lower than the boost start driving frequency fψs. Figure 35 In the process, the control unit 115A performs synchronous rectification control and frequency boost conversion control. Figure 31 The difference is that there is a boost period ψ in Mode 1-1, Mode 3-1 and Mode 5-1.

[0348] The control unit 115A during synchronous rectification control and frequency step-up conversion control turns on the switching element Q8 at the start of the resonance period in which the resonant current flows through the first resonant circuit and the second resonant circuit, while keeping the switching element Q7 off, to generate the boost period ψ of mode 1-1. At the start of the resonance period in which the resonant current flows through the second resonant circuit and the third resonant circuit, the control unit 115A turns on the switching element Q10 at the start of the resonance period in which the resonant current flows through the second resonant circuit and the third resonant circuit, while keeping the switching element Q9 off, to generate the boost period ψ of mode 3-1. At the start of the resonance period in which the resonant current flows through the third resonant circuit and the first resonant circuit, the control unit 115A turns on the switching element Q12 at the start of the resonance period in which the resonant current flows through the third resonant circuit and the first resonant circuit, while keeping the switching element Q11 off, to generate the boost period ψ of mode 5-1.

[0349] During synchronous rectification control and frequency-boost conversion control, control unit 115A turns off switching element Q8 and turns on switching element Q7 at the end of boost period ψ in mode 1-1 to initiate synchronous rectification. At the end of boost period ψ in mode 3-1, control unit 115A turns off switching element Q10 and turns on switching element Q9 to initiate synchronous rectification. At the end of boost period ψ in mode 5-1, control unit 115A turns off switching element Q12 and turns on switching element Q11 to initiate synchronous rectification. Furthermore, dead time may be provided to prevent load short-circuiting due to overlapping off / on periods when transitioning from mode 1-1 to mode 1, from mode 3-1 to mode 3, and from mode 5-1 to mode 5.

[0350] Figures 36 to 38 Show Figure 35 Current path diagram during modes 1-1, 1, 3-1, 3, 5-1, 5. Figure 35 The current path during Mode 2, Mode 4, and Mode 6 is Figure 31 The current paths during Mode 2, Mode 4, and Mode 6 are the same and are therefore omitted.

[0351] like Figure 36 As shown in (A), during mode 1-1, when the switching element Q1 is turned on, the resonant current flows through the first resonant circuit and the second resonant circuit, and the load current flows through the secondary circuit via the transformer circuits Tr1 and Tr2. In the secondary circuit, since the switching elements Q8 and Q10 are turned on, the secondary switch circuit 113 is short-circuited through the path of the switching elements Q8 and Q10 (or the diode D10). As a result, a resonant current larger than the normal resonant current flows through the first resonant circuit and the second resonant circuit, and a large amount of energy is stored in the resonant coils Lr1 and Lr2. Figure 36 As shown in (B), during Mode 1, switching element Q8 is turned off and switching element Q7 is turned on. This releases the energy stored during Mode 1-1, causing a load current greater than normal to flow through the secondary circuit. Thus, DC / DC converter 100A performs a boost operation during Mode 1-1 and Mode 1.

[0352] like Figure 37As shown in (A), during mode 3-1, when the switching element Q3 is turned on, the resonant current flows through the second and third resonant circuits, and the load current flows through the secondary circuit via the transformer circuits Tr2 and Tr3. In the secondary circuit, since the switching elements Q10 and Q12 are turned on, the secondary switch circuit 113 is short-circuited through the path of the switching element Q10 and the switching element Q12 (or the diode D12). As a result, a resonant current larger than the normal resonant current flows through the second and third resonant circuits, and a large amount of energy is stored in the resonant coils Lr2 and Lr3. Figure 37 As shown in (B), during Mode 3, switching element Q10 is turned off and switching element Q9 is turned on. This releases the energy stored during Mode 3-1, causing a load current greater than normal to flow through the secondary circuit. Thus, DC / DC converter 100A performs a boost operation during Modes 3-1 and 3.

[0353] like Figure 38 As shown in (A), during mode 5-1, when switching element Q5 is turned on, a resonant current flows through the third resonant circuit and the first resonant circuit, and the load current flows through the secondary circuit via transformer circuits Tr3 and Tr1. In the secondary circuit, since switching elements Q12 and Q8 are turned on, the secondary switch circuit 113 is short-circuited through the path of switching element Q12 and switching element Q8 (or diode D8). As a result, a resonant current larger than the normal resonant current flows through the third resonant circuit and the first resonant circuit, and a large amount of energy is stored in the resonant coils Lr3 and Lr1. Figure 38 As shown in (B), during Mode 5, switching element Q12 is turned off and switching element Q11 is turned on. This releases the energy stored during Mode 5-1, causing a load current greater than normal to flow through the secondary circuit. Thus, DC / DC converter 100A performs a boost operation during Modes 5-1 and 5.

[0354] In this embodiment, control unit 115A synchronizes the on-time of switching element Q7 with the off-time of switching element Q8, starting synchronous rectification at the end of boost period ψ in mode 1-1. However, this is not limiting; the timing for starting synchronous rectification can be varied as appropriate, as long as boost period ψ occurs within the resonance period. However, if mode 1-1 follows mode 1, switching element Q8 will turn on after the load current increases, causing hard switching of the load current and potentially generating noise. Therefore, it is preferable to start synchronous rectification at the end of boost period ψ, as in this embodiment. The same applies to modes 3-1 and 3, and modes 5-1 and 5.

[0355] [First Modification of Three-Phase Current Resonance DC / DC Converter]

[0356] Figure 39 A three-phase current resonance type DC / DC converter 100B (hereinafter referred to as DC / DC converter 100B) according to a first modification of the second embodiment of the present invention is shown. The DC / DC converter 100B has the same configuration as that of the second embodiment except for a control unit 115B.

[0357] The control unit 115B has Figure 40 The present embodiment is the same as the second embodiment except that the output voltage step-up conversion control block 120 ′ is used instead of the frequency step-up conversion control block 120 of the second embodiment and output voltage step-up conversion control is performed instead of frequency step-up conversion control.

[0358] like Figure 40 As shown, the output voltage boost conversion control block 120 ′ includes a second adding unit 121 ′, a second multiplying unit 122 ′, and a second clamping unit 123 ′.

[0359] The second adding unit 121' compares the output voltage V2 with the target voltage V2t [V]. When the input voltage V1 is less than a predetermined threshold value (V1th) and the driving frequency f is less than the boost start driving frequency fψs, the second adding unit 121' outputs the difference (V2t-V2) between the output voltage V2 and the target voltage V2t to the second multiplying unit 122'.

[0360] The second multiplication unit 122' multiplies the difference (V2t - V2) by the output voltage boost conversion gain K'ψ(V1x) to calculate the boost amount ψ, and outputs this boost amount ψ to the second clamping unit 123'. Like the frequency boost conversion gain Kψ(V1x) of the second embodiment, the output voltage boost conversion gain K'ψ(V1x) is determined by the input voltage V1 at that time (e.g., V1 = V11 or V12).

[0361] The second clamping unit 123' sets the lower limit of the boost amount ψ to 0° and the upper limit to the maximum boost amount ψh(V1x), and outputs the boost amount ψ (ψ = (V2t - V2) × K' ψ(V1x) ≤ ψh(V1x)). As in the second embodiment, the maximum boost amount ψh(V1x) is determined by the input voltage V1 at that time (e.g., ψh(V1x) = ψh(V11) or ψh(V12)).

[0362] Control unit 115B performs output voltage boost conversion control based on the boost amount ψ outputted from second clamp unit 123'. While this variation varies the output voltage boost conversion gain K'ψ(V1x) and the maximum boost amount ψh(V1x) based on input voltage V1, the boost start drive frequency fψs may also be varied based on input voltage V1.

[0363] Figure 41 An example of a flowchart showing a control process executed by the control section 115B at the time of forward power transmission is shown. Figure 22 The difference lies in steps S203, S207, and S208. Steps S201 and S202 are the same as steps S101 and S102, steps S204 to S206 are the same as steps S104 to S106, and steps S209 to S213 are the same as steps S108 to S112. Only steps S203, S207, and S208 are described below.

[0364] In step S203, the control unit 115B reads the control parameters stored in the storage unit. These control parameters include the output voltage boost conversion control parameters, namely, the boost start drive frequency fψs, the output voltage boost conversion gain K'ψ(V1x), and the maximum boost amount ψh(V1x). Fixed parameters among the control parameters may be read only once at startup.

[0365] In step S207, the control unit 115B sets the drive frequency f to the boost start drive frequency fψs. That is, the output voltage boost conversion control differs from the frequency boost conversion control in that the drive frequency f is fixed to the boost start drive frequency fψs.

[0366] In step S208, the control unit 115B multiplies the difference (V2t-V2) between the output voltage V2 and the target voltage V2t by the output voltage boost conversion gain K'ψ(V1x) to calculate the boost amount ψ. Figure 22 In the embodiment, the driving frequency f is changed to determine the boost amount ψ to perform frequency boost conversion control, thereby boosting the output voltage V2. Figure 41 The difference is that the driving frequency f is set to the boost start driving frequency fψs, and the boost amount ψ is determined according to the output voltage V2 itself to perform boost control.

[0367] The DC / DC converter 100B of the first modified example achieves the same advantages as the second embodiment, in addition to the advantages of calculating the boost amount ψ based on the drive frequency f. Specifically, by performing boost control (output voltage boost conversion control) when the drive frequency f is below the boost start drive frequency fψs, DC / DC converter 100B allows a large resonant current to flow through the resonant circuit during the boost period ψ, quickly accumulating a large amount of energy in the resonant coils Lr1 to Lr3 and discharging this energy as load current. This allows a high output voltage V2 to be achieved even when the input voltage V1 is low. As a result, DC / DC converter 100B supports a wide range of output voltages V2 without adding additional components or circuits. Similar advantages can also be achieved during reverse power transmission.

[0368] [Second Modification of Three-Phase Current Resonance DC / DC Converter]

[0369] Figure 42 A three-phase current resonance type DC / DC converter 100C (hereinafter referred to as DC / DC converter 100C) is shown as a second modification of the second embodiment of the present invention. The DC / DC converter 100C has the same configuration as the second embodiment, except that it includes a secondary-side switching circuit 113C and a control unit 115C, and does not include a secondary-side resonant circuit 114.

[0370] The DC / DC converter 100C is an LLC-type three-phase current resonance DC / DC converter that performs forward power transmission from the primary circuit to the secondary circuit and does not perform reverse power transmission from the secondary circuit to the primary circuit. The DC / DC converter 100C does not need to include the capacitor Co1.

[0371] The secondary-side switch circuit 113C is the same as the secondary-side switch circuit 113 of the second embodiment, except that the upper branch of the fourth branch is composed only of the diode D7', the upper branch of the fifth branch is composed only of the diode D9', and the upper branch of the sixth branch is composed only of the diode D11'.

[0372] The control unit 115C has the same configuration as the control unit 115A of the second embodiment, except that it does not perform control during reverse power transmission and does not control the switching elements Q7 , Q9 , and Q11 .

[0373] In DC / DC converter 100C, by performing boost control (frequency boost conversion control) when the drive frequency f is equal to or lower than the boost start drive frequency fψs, a large resonant current flows through the resonant circuit during the boost period ψ. This allows a large amount of energy to be stored in the resonant coils Lr1 to Lr3 in a short period of time and released as load current. This allows a high output voltage V2 to be obtained even when the input voltage V1 is low. Consequently, DC / DC converter 100C supports a wide range of output voltages V2 without requiring additional components or circuitry.

[0374] Furthermore, in the second modified example, similar to the case of diode rectification of the secondary-side switching circuit 113 in the second embodiment, switching element Q8 can be turned on after a certain time has passed since switching element Q1 was turned on. However, if mode 1-1 follows mode 1, switching element Q8 is turned on after the load current increases, and secondary-side switching circuit 113 becomes short-circuited during the reverse recovery period of diode D7, potentially generating noise. Therefore, similar to the second embodiment, it is preferable to generate the boost period ψ at the start of the resonance period. The same applies to modes 3-1 and 5-1.

[0375] In addition, the control unit 115C may have the same configuration as the control unit 115B according to the first modification of the second embodiment.

[0376] [Other Modifications of Three-Phase Current Resonance DC / DC Converter]

[0377] While the embodiment and its modified examples of the three-phase current resonance type DC / DC converter of the present invention have been described above, the present invention is not limited to the above-described embodiment and its modified examples.

[0378] The three-phase current resonance type DC / DC converter of the present invention comprises: a transformer portion including a first transformer circuit, a second transformer circuit, and a third transformer circuit, each transformer circuit including a primary-side coil and a secondary-side coil; a primary-side switching circuit including a first branch, a second branch, and a third branch connected in parallel, each branch including an upper branch and a lower branch connected in series, each branch including a switching element, a reverse-connected diode, and a partial resonant capacitor connected in parallel; a primary-side resonant circuit including a first resonant circuit, a second resonant circuit, and a third resonant circuit, the first resonant circuit being connected to the first branch and the primary-side coil of the first transformer circuit, the second resonant circuit being connected to the second branch and the primary-side coil of the second transformer circuit, and the third resonant circuit being connected to the third branch and the primary-side coil of the third transformer circuit, each resonant circuit including a resonant coil and a resonant capacitor; and a secondary-side switching circuit including a fourth branch, a fifth branch, and a sixth branch connected in parallel. Each branch includes a switching element, a reverse-connected diode, and a partial resonant capacitor connected in parallel; and a control unit that controls the primary-side switching circuit and the secondary-side switching circuit. The three-phase current resonant DC / DC converter performs forward power transmission from the primary-side switching circuit to the secondary-side switching circuit. The control unit performs frequency modulation control to control the driving frequency based on the output of the secondary-side switching circuit when the driving frequency of the primary-side switching circuit is higher than a first frequency, and performs boost control to generate a boost period in which the secondary-side switching circuit is short-circuited when the driving frequency is lower than the first frequency. The structure can be changed as appropriate as long as the control unit during the boost control causes any switching element in the secondary-side switching circuit to conduct to generate the boost period during the period when the resonant current flows through the first resonant circuit and the second resonant circuit, the period when the resonant current flows through the second resonant circuit and the third resonant circuit, and the period when the resonant current flows through the third resonant circuit and the first resonant circuit.

[0379] For example, in the second embodiment and the first variant of the second embodiment, since the secondary side of the first transformer circuit is connected to the fourth branch, the secondary side of the second transformer circuit is connected to the fifth branch, and the secondary side of the third transformer circuit is connected to the sixth branch of the secondary-side switching circuit, the control unit during the boost control can be configured as follows: during the period when the resonant current flows through the first resonant circuit and the second resonant circuit, the switching element of the lower branch of the fourth branch or the upper branch of the fifth branch is turned on to produce a boost period; during the period when the resonant current flows through the second resonant circuit and the third resonant circuit, the switching element of the lower branch of the fifth branch or the upper branch of the sixth branch is turned on to produce a boost period; during the period when the resonant current flows through the third resonant circuit and the first resonant circuit, the switching element of the lower branch of the sixth branch or the upper branch of the fourth branch is turned on to produce a boost period.

[0380] Furthermore, similarly, in the second variation of the second embodiment, the upper arms of the fourth, fifth, and sixth branches are configured as diodes, and the lower arms are configured as switching elements. However, the diodes and switching elements of each branch may be interchanged as appropriate. For example, one of the lower arm of the fourth branch and the upper arm of the fifth branch may be configured as a switching element, a reverse-connected diode, and a partial resonant capacitor connected in parallel, while the other may be configured as a diode alone. Alternatively, one of the lower arm of the fifth branch and the upper arm of the sixth branch may be configured as a switching element, a reverse-connected diode, and a partial resonant capacitor connected in parallel, while the other may be configured as a diode alone. Alternatively, one of the lower arm of the sixth branch and the upper arm of the fourth branch may be configured as a switching element, a reverse-connected diode, and a partial resonant capacitor connected in parallel, while the other may be configured as a diode alone. In this case, the control unit during boost control can be constructed as follows: during the period when the resonant current flows through the first resonant circuit and the second resonant circuit, the switching element of the lower branch of the fourth branch or the upper branch of the fifth branch is turned on to produce a boost period; during the period when the resonant current flows through the second resonant circuit and the third resonant circuit, the switching element of the lower branch of the fifth branch or the upper branch of the sixth branch is turned on to produce a boost period; during the period when the resonant current flows through the third resonant circuit and the first resonant circuit, the switching element of the lower branch of the sixth branch or the upper branch of the fourth branch is turned on to produce a boost period.

[0381] For example, in the second embodiment and its second modified example, the boost amount ψ is calculated based on the drive frequency f, and in the first modified example of the second embodiment, the boost amount ψ is calculated based on the output voltage V2. However, the boost amount ψ may be calculated based on an output other than the output voltage V2 (e.g., output current or output power). In other words, the control unit during boost control can be configured to calculate the boost amount by multiplying the difference between the output value of the secondary-side circuit and a predetermined target value by the output boost conversion gain, and determine the length of the boost period based on this boost amount.

Claims

1. A single-phase current resonant DC / DC converter, It is characterized in that Including: main circuit part and control part, The main circuit unit comprises: Transformer circuit; a primary-side switching circuit, disposed on the primary side of the transformer circuit, comprising at least one primary-side branch, wherein the primary-side branch comprises a group of primary-side switching elements connected in series; a primary-side resonant circuit, disposed between the primary-side branch and the transformer circuit, comprising a resonant coil and a resonant capacitor; and The secondary-side rectifier circuit is provided on the secondary side of the transformer circuit and includes at least one secondary-side branch. The secondary-side branch includes a set of rectifier units connected in series. The rectifier units include diodes or secondary-side switching elements. The control unit controls the primary-side switching element and / or the secondary-side switching element. The control unit executes: Frequency modulation control, when the driving frequency of the primary-side switching element is between a first frequency and a second frequency lower than the first frequency, controlling the output of the main circuit portion using the driving frequency as a control variable; and Control of at least one of output suppression conversion control and output increase conversion control, wherein the output suppression conversion control is a control in which, when the driving frequency exceeds the first frequency, the output is reduced by using a first control amount calculated using the difference between the driving frequency and the first frequency as the control amount, and the output increase conversion control is a control in which, when the driving frequency is lower than the second frequency, the output is increased by using a second control amount calculated using the difference between the second frequency and the driving frequency as the control amount.

2. The single-phase current resonance DC / DC converter according to claim 1, wherein: The output suppression conversion control includes: Frequency PWM conversion control, based on the first control amount, controlling the pulse width of the PWM signal for the primary side switching element or the secondary side switching element, or Frequency intermittent conversion control, based on the first control amount, controls the standby period during which the switching operation of the primary side switching element is not performed, or The frequency phase shift conversion control controls the phase shift amount between the primary-side branches based on the first control amount.

3. The single-phase current resonance DC / DC converter according to claim 1, wherein: The output rising conversion control is a frequency boosting conversion control in which, based on the second control amount, the conduction period of the secondary-side switching element of the secondary-side rectifier circuit is controlled while the resonant current flows in the primary-side resonant circuit, thereby short-circuiting the secondary-side rectifier circuit.

4. The single-phase current resonance DC / DC converter according to claim 1, wherein: The first control amount is calculated by calculating the difference between the driving frequency and the first frequency and a predetermined first gain. The control unit during the output suppression conversion control makes the value of the first frequency and / or the first gain variable according to input / output conditions related to the input voltage, input / output voltage ratio, or output power of the main circuit unit.

5. The single-phase current resonance DC / DC converter according to claim 1, wherein: The second control amount is calculated by calculating the difference between the second frequency and the driving frequency and a predetermined second gain. The control unit during the output step-up conversion control makes the value of the second frequency and / or the second gain variable according to input / output conditions related to the input voltage, input-output voltage ratio, or output power of the main circuit unit.

6. The single-phase current resonance DC / DC converter according to claim 1, wherein: The control unit during the output suppression conversion control performs control so that the first control amount does not exceed a predetermined first maximum control amount. The control unit during the output increase conversion control performs control so that the second control amount does not exceed a predetermined second maximum control amount.

7. The single-phase current resonance type DC / DC converter according to claim 1, wherein: The main circuit unit includes a resonant coil and a resonant capacitor. The primary-side switching circuit includes two primary-side branches consisting of a first branch and a second branch connected in parallel. The secondary-side rectifier circuit includes two secondary-side branches consisting of a third branch and a fourth branch connected in parallel, and the rectifier unit of each branch is composed of the secondary-side switching element, or only the rectifier units constituting the upper and lower branches of the third branch or the fourth branch are composed of the secondary-side switching element, or only the rectifier units constituting the lower branch of the third branch and the upper branch or the lower branch of the fourth branch are composed of the secondary-side switching element, and a reverse-connected diode is connected in parallel on the current path of the secondary-side switching element.

8. The single-phase current resonance type DC / DC converter according to claim 7, wherein: The control unit performs frequency phase shift conversion control as the output suppression conversion control, During the frequency phase shift conversion control, the control unit determines the driving frequency by comparing the output value of the output with the target value, and uses the value obtained by multiplying the difference between the driving frequency and the first frequency by a predetermined first gain as the phase shift amount, so that there is a phase difference corresponding to the phase shift amount between the first branch and the second branch.

9. The single-phase current resonance DC / DC converter according to claim 7, wherein: The control unit performs frequency boost conversion control as the output increase conversion control, During the frequency boost conversion control, the control unit determines the driving frequency by comparing the output value of the output with a target value, and uses the value obtained by multiplying the difference between the second frequency and the driving frequency by a predetermined second gain as the boost amount, so that the secondary-side switching element has a conduction period corresponding to the boost amount.

10. The single-phase current resonance DC / DC converter according to claim 1, wherein: The main circuit unit includes a resonant coil and a resonant capacitor. The primary-side switching circuit includes two primary-side branches consisting of a first branch and a second branch connected in parallel. The secondary-side rectifier circuit includes two secondary-side branches consisting of a third branch and a fourth branch connected in parallel, and the rectifier unit of each branch is composed of the secondary-side switching element. The control unit performs frequency PWM conversion control as the output suppression conversion control, During the frequency PWM conversion control, the control unit determines the driving frequency by comparing the output value of the output with the target value, and uses the difference between the driving frequency and the first frequency to calculate the phase difference as the first control quantity, so that the conduction timing of the secondary side switching element is staggered relative to the conduction timing of the primary side switching element by an amount corresponding to the phase difference.

11. A three-phase current resonance DC / DC converter, characterized in that: have: a transformer unit including a first transformer circuit, a second transformer circuit, and a third transformer circuit, each transformer circuit including a primary-side coil and a secondary-side coil; a primary-side switching circuit comprising a first branch, a second branch, and a third branch connected in parallel, each branch comprising an upper branch and a lower branch connected in series, each branch comprising a switching element, a reverse-connected diode, and a portion of a resonant capacitor connected in parallel; a primary-side resonant circuit comprising a first resonant circuit, a second resonant circuit, and a third resonant circuit, wherein the first resonant circuit is connected to the first branch and the primary-side coil of the first transformer circuit, the second resonant circuit is connected to the second branch and the primary-side coil of the second transformer circuit, and the third resonant circuit is connected to the third branch and the primary-side coil of the third transformer circuit, each resonant circuit comprising a resonant coil and a resonant capacitor; The secondary-side switching circuit includes a fourth branch, a fifth branch, and a sixth branch connected in parallel, each branch including a switching element, a reverse-connected diode, and a portion of a resonant capacitor connected in parallel; as well as a control unit that controls the primary-side switching circuit and the secondary-side switching circuit, The three-phase current resonance type DC / DC converter performs forward power transmission from the primary side switching circuit to the secondary side switching circuit, wherein When the driving frequency of the primary-side switching circuit is higher than a first frequency, the control unit performs frequency modulation control to control the driving frequency according to the output of the secondary-side switching circuit. The control unit performs boost control for generating a boost period in which the secondary-side switching circuit is short-circuited when the driving frequency is equal to or lower than the first frequency. During the boost control, the control unit turns on any of the switching elements in the secondary-side switching circuit to generate the boost period during a period in which the resonant current flows through the first resonant circuit and the second resonant circuit, a period in which the resonant current flows through the second resonant circuit and the third resonant circuit, and a period in which the resonant current flows through the third resonant circuit and the first resonant circuit.

12. The three-phase current resonance DC / DC converter according to claim 11, wherein: In the transformer unit, The secondary side coil of the first transformer circuit is connected to the fourth branch. The secondary side coil of the second transformer circuit is connected to the fifth branch. The secondary side coil of the third transformer circuit is connected to the sixth branch. In the secondary-side switching circuit, each of the fourth branch, the fifth branch, and the sixth branch includes an upper branch and a lower branch connected in series, and each branch includes the switching element, the reverse-connected diode, and the partial resonant capacitor connected in parallel. The control unit during the boost control: During a period in which a resonant current flows through the first resonant circuit and the second resonant circuit, the switching element of the lower arm of the fourth branch or the switching element of the upper arm of the fifth branch is turned on to generate the boost period. During a period in which a resonant current flows through the second resonant circuit and the third resonant circuit, the switching element of the lower arm of the fifth branch or the switching element of the upper arm of the sixth branch is turned on to generate the boost period. During a period in which the resonant current flows in the third resonant circuit and the first resonant circuit, the switching element of the lower arm of the sixth branch or the switching element of the upper arm of the fourth branch is turned on to generate the boost period.

13. The three-phase current resonance type DC / DC converter according to claim 11, wherein: During the boost control, the control unit calculates a boost amount by multiplying a difference between the driving frequency and the first frequency by a frequency boost conversion gain, and determines a length of the boost period based on the boost amount.

14. The three-phase current resonance type DC / DC converter according to claim 11, wherein: During the boost control, the control unit calculates a boost amount by multiplying a difference between an output value of the secondary-side switching circuit and a predetermined target value by an output boost conversion gain, and determines a length of the boost period based on the boost amount.

15. The three-phase current resonance type DC / DC converter according to claim 11, wherein: The control unit during the boost control starts the boost period at the timing when the period during which the resonant current flows starts.

16. The three-phase current resonance type DC / DC converter according to claim 11, wherein: In the secondary-side switching circuit, each of the fourth branch, the fifth branch, and the sixth branch includes an upper branch and a lower branch connected in series, and each branch includes the switching element, the reverse-connected diode, and the partial resonant capacitor connected in parallel. The control unit controls the secondary-side switching circuit to turn on any of the switching elements in each branch to generate a synchronous rectification period. The control unit in the boost control and the synchronous rectification control starts the boost period at the start of the period in which the resonant current flows, and starts the synchronous rectification period at the end of the boost period.

17. The three-phase current resonance type DC / DC converter according to claim 11, wherein: The control unit makes the maximum value of the boost period variable according to the input-output voltage ratio so that the boost period becomes longer when the input-output voltage ratio of the input voltage of the primary-side switching circuit to the output voltage of the secondary-side switching circuit is higher.

18. The three-phase current resonance DC / DC converter according to claim 11, wherein: In the secondary side switching circuit, Each of the fourth branch, the fifth branch, and the sixth branch includes an upper branch and a lower branch connected in series. One of the lower branch of the fourth branch and the upper branch of the fifth branch is composed of the switching element, the reverse-connected diode, and the partial resonant capacitor connected in parallel, and the other is composed of only a diode. One of the lower branch of the fifth branch and the upper branch of the sixth branch is composed of the switching element, the reverse-connected diode, and the partial resonant capacitor connected in parallel, and the other is composed of only a diode. One of the lower branch of the sixth branch and the upper branch of the fourth branch is composed of the switching element, the reverse-connected diode, and the partial resonant capacitor connected in parallel, and the other is composed of only a diode. The control unit during the boost control: During a period in which a resonant current flows through the first resonant circuit and the second resonant circuit, the switching element of the lower arm of the fourth branch or the switching element of the upper arm of the fifth branch is turned on to generate the boost period. During a period in which a resonant current flows through the second resonant circuit and the third resonant circuit, the switching element of the lower arm of the fifth branch or the switching element of the upper arm of the sixth branch is turned on to generate the boost period. During a period in which the resonant current flows in the third resonant circuit and the first resonant circuit, the switching element of the lower arm of the sixth branch or the switching element of the upper arm of the fourth branch is turned on to generate the boost period.

19. The three-phase current resonance DC / DC converter according to claim 11, wherein: The secondary-side resonant circuit further comprises a fourth resonant circuit, a fifth resonant circuit, and a sixth resonant circuit, wherein the fourth resonant circuit is connected to the fourth branch and the secondary-side coil of the first transformer circuit, the fifth resonant circuit is connected to the fifth branch and the secondary-side coil of the second transformer circuit, and the sixth resonant circuit is connected to the sixth branch and the secondary-side coil of the third transformer circuit, each resonant circuit comprising a resonant coil and a resonant capacitor. In the secondary-side switching circuit, each of the fourth branch, the fifth branch, and the sixth branch includes an upper branch and a lower branch connected in series, and each branch includes the switching element, the reverse-connected diode, and the partial resonant capacitor connected in parallel. When performing reverse power transmission from the secondary-side switching circuit to the primary-side switching circuit, the control unit: When the driving frequency of the secondary-side switching circuit is higher than the second frequency, frequency modulation control is performed to control the driving frequency according to the output of the primary-side switching circuit. When the driving frequency is equal to or lower than the second frequency, a boost control is performed to generate a boost period in which the primary-side switching circuit is short-circuited. During the boost control, the control unit turns on any of the switching elements in the primary-side switching circuit to generate the boost period during a period in which the resonant current flows through the fourth resonant circuit and the fifth resonant circuit, a period in which the resonant current flows through the fifth resonant circuit and the sixth resonant circuit, and a period in which the resonant current flows through the sixth resonant circuit and the fourth resonant circuit.

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