External phase shift angle determination method and system, frequency conversion phase shift control method and chip

Abstract

The invention relates to an external phase shift angle determination method and system, a frequency conversion phase shift control method and a chip, and the method comprises the steps: obtaining an external phase shift angle based on a general solution of a time domain differential equation of a resonant inductor current and a resonant capacitor voltage of a DBSRC equivalent circuit and the symmetry of the DBSRC equivalent circuit; determining a first current value of the resonant inductor at the first moment and a second current value of the resonant inductor at the second moment; based on the first current value when the primary bridge circuit meets the ZVS soft switching requirement, obtaining a first relational expression of the external phase shift angle about the input voltage and the output voltage of the DBSRC and the switching frequency of the switching tube; based on a second current value when the secondary-side bridge circuit meets the ZVS soft switching requirement, obtaining a second relational expression of the external phase shift angle with respect to the input voltage, the output voltage and the switching frequency; according to the first relational expression, the second relational expression and the steady-state voltage value of the output end of the DBSRC, the expression of the external phase shift angle about the switching frequency, the steady-state voltage value and the input voltage is determined; the scheme is simple, the accuracy is high, and the portability is high.

Description

TECHNICAL FIELD

[0001] The present application relates to the technical field of double bridge series resonant converter, and particularly relates to a method and system for determining an external phase angle, a variable frequency phase shift control method and a chip. BACKGROUND

[0002] The double bridge series resonant converter (DBSRC) involves control of an external phase angle in optimizing output gain and efficiency.

[0003] In related technologies, methods for optimizing the efficiency of DBSRC mainly include loop current power optimization, peak current optimization or effective value current optimization, etc. However, the calculation of these optimization methods is usually complex, and a computer needs to be used to solve the problem with the help of multiple mathematical calculation software to obtain the external phase angle. In addition, the inductance and other parameters of the circuit need to be known and solved by multiple mathematical calculation software, so when the circuit parameters are replaced, the method needs to be recalculated, and the portability of the method is low.

[0004] In addition, in related technologies, the derivation of the external phase angle is approximated by fundamental wave equivalence and frequency domain analysis. When the switching frequency fs of the circuit and the resonant frequency fr of the circuit differ greatly, the calculation accuracy of the voltage and current will decrease due to the neglect of the influence of high-order harmonics by the fundamental wave equivalence, thereby affecting the control effect. SUMMARY

[0005] Therefore, it is necessary to provide a simple, accurate and portable method and system for determining an external phase angle, a variable frequency phase shift control method and a computer chip, which are applied to DBSRC to optimize output gain and efficiency.

[0006] In a first aspect, an embodiment of the present application provides a method for determining an external phase angle, which is applied to a double bridge series resonant converter, and the method comprises the following steps.

[0007] Based on a general solution of a time domain differential equation of a resonant inductor current of a DBSRC equivalent circuit of the double bridge series resonant converter, a general solution of a time domain differential equation of a resonant capacitor voltage of the DBSRC equivalent circuit and symmetry of the DBSRC equivalent circuit, a first current value of the resonant inductor at a first time and a second current value of the resonant inductor at a second time are determined.

[0008] Based on the first current value when the primary bridge circuit meets the ZVS soft switching requirement, a first relationship of the external phase angle with respect to an input voltage, an output voltage and a switching frequency of a switching tube of the double bridge series resonant converter is obtained.

[0009] obtaining a second relationship of the phase-shifted angle with respect to the input voltage, the output voltage and the switching frequency based on the second current value when the secondary bridge circuit meets the ZVS soft switching requirement;

[0010] determining an expression of the phase-shifted angle with respect to the switching frequency, the steady-state voltage value and the input voltage according to the first relationship, the second relationship and a steady-state voltage value of an output terminal of the double-bridge series resonant conversion circuit;

[0011] wherein the first time point is a time point when a first upper switch tube of a first left bridge arm of the primary bridge circuit starts to turn on, and the second time point is a time point when a second upper switch tube of a second left bridge arm of the secondary bridge circuit starts to turn on in a conduction process of the first upper switch tube.

[0012] Based on the same inventive concept, in a second aspect, embodiments of the present application provide a variable frequency phase-shifted control method applied to a double-bridge series resonant conversion circuit, and the control method comprises:

[0013] sampling an output terminal of the double-bridge series resonant conversion circuit to obtain an actual voltage value of the output terminal at a current time point;

[0014] obtaining a first switching frequency of a switch tube when a voltage of the output terminal reaches a steady-state voltage value based on the actual voltage value and a PI control principle;

[0015] obtaining a required switching frequency of the double-bridge series resonant conversion circuit in a variable frequency phase-shifted control process based on the first switching frequency and a preset maximum switching frequency;

[0016] obtaining a required phase-shifted angle in a variable frequency phase-shifted control process at a current time point according to the required switching frequency, an input voltage of an input terminal of the double-bridge series resonant conversion circuit at the current time point, the steady-state voltage value of the output terminal and the expression obtained based on the determination method of the phase-shifted angle of the first aspect;

[0017] obtaining a PWM driving signal of a switch tube of the primary bridge circuit and a PWM driving signal of a switch tube of the secondary bridge circuit according to a preset duty cycle value of the switch tube, the required switching frequency and the required phase-shifted angle.

[0018] Based on the same inventive concept, in a third aspect, embodiments of the present application provide a phase-shifted angle determination system, which comprises:

[0019] The double-bridge series resonant conversion circuit comprises a primary bridge circuit, a resonant network and a secondary bridge circuit; an input end of the primary bridge circuit is connected to an input power supply providing an input voltage, an input end of the resonant network is connected to an output end of the primary bridge circuit, an input end of the secondary bridge circuit is connected to an output end of the resonant network, and an output end of the secondary bridge circuit provides an output voltage for a load; the resonant network comprises a transformer, a resonant inductor and a resonant capacitor.

[0020] The execution unit is configured to collect the input voltage and an output voltage of the output end of the secondary bridge circuit, and to execute the steps of the method according to the first aspect when the output voltage reaches a set steady-state voltage value.

[0021] Based on the same inventive concept, in a fourth aspect, the embodiments of the present application provide a computer chip, comprising a memory and a processor, the memory stores a computer program, and the processor executes the computer program to implement the steps of the method according to the first aspect and the second aspect.

[0022] The above-mentioned phase-shift angle determination method and system, variable frequency phase-shift control method and computer chip first determine a first current value of the resonant inductor at a first time and a second current value of the resonant inductor at a second time based on a general solution of a time-domain differential equation of a resonant inductor current of a DBSRC equivalent circuit of the DBSRC, a general solution of a time-domain differential equation of a resonant capacitor voltage of the DBSRC equivalent circuit and symmetry of the DBSRC equivalent circuit; then, based on the first current value when the primary bridge circuit meets the ZVS soft switching requirement, a first relationship between the phase-shift angle and the input voltage, the output voltage and the switching frequency of the switching tube of the DBSRC is obtained, and based on the second current value when the secondary bridge circuit meets the ZVS soft switching requirement, a second relationship between the phase-shift angle and the input voltage, the output voltage and the switching frequency is obtained; finally, according to the first relationship, the second relationship and a steady-state voltage value of the output end of the DBSRC, an expression of the phase-shift angle with respect to the switching frequency, the steady-state voltage value and the input voltage is determined.

[0023] Therefore, compared with the fundamental equivalent and frequency domain analysis in the related art, the time domain method is used for formula derivation in the embodiment of the application, which can ensure high accuracy of optimization; compared with the method for efficiency optimization of DBSRC in the related art, the embodiment of the application obtains a physical measurement expression of the phase shift angle, which can adapt to DBSRC of different parameters, and for DBSRC circuits of different circuit parameters, the phase shift angle can be directly obtained through the known input voltage, required output voltage and switching frequency of the current circuit, the calculation of the phase shift angle is relatively simple, and it is not necessary to solve through mathematical software such as matlab and mathcad in the prior art, and the portability is high in DBSRC of different parameters, which can be applied to calculation of DSP (Digital Signal Processor), ARM (Advanced RISC Machines) and other single-chip microcomputers in each control cycle, thereby ensuring the real-time performance of circuit control. In addition, the embodiment of the application is optimized by using critical ZVS, and the soft switching condition of the circuit is ensured through real-time calculation, thereby reducing the calculation amount and improving the efficiency of the circuit. BRIEF DESCRIPTION OF DRAWINGS

[0024] In order to more clearly illustrate the technical solutions of the embodiments of the application or the prior art, the drawings needed to be used in the embodiments or the prior art description will be briefly introduced. Obviously, the drawings in the following description are only some embodiments of the application, and other drawings can be obtained by those skilled in the art without creative labor.

[0025] Figure 1A FIG. 1 is a schematic diagram of a topology structure of a double-bridge series resonant conversion circuit according to an embodiment of the application;

[0026] Figure 1B FIG. 2 is a schematic diagram of a topology structure of a double-bridge series resonant conversion circuit according to another embodiment of the application;

[0027] Figure 1C FIG. 3 is an equivalent circuit diagram of DBSRC according to an embodiment of the application; Figure 1A

[0028] Figure 2 FIG. 5 is a partial driving signal waveform diagram, a resonant inductance current waveform diagram and a resonant capacitance voltage waveform diagram of the double-bridge series resonant conversion circuit in FIG. 4; Figure 1A

[0029] FIG. 6 is a flowchart of a method for determining a phase shift angle according to an embodiment of the application; Figure 3

[0030] FIG. 7 is a flowchart of step S301 in the method for determining a phase shift angle according to an embodiment of the application; Figure 4 ​​

[0031] Figure 5 Flowchart of step S302 in the method for determining the lead-out phase angle of an embodiment;

[0032] Figure 6 Flowchart of step S302 in the method for determining the lead-out phase angle of an embodiment;

[0033] Figure 7 Flowchart of the method for variable frequency phase-shift control of an embodiment;

[0034] Figure 8 Control schematic involved in the method for variable frequency phase-shift control of an embodiment;

[0035] Figure 9 Circuit working state schematic obtained from simulation based on the method for determining the lead-out phase angle of an embodiment;

[0036] Figure 10 Output gain schematic obtained from simulation based on the method for determining the lead-out phase angle of an embodiment. DETAILED DESCRIPTION

[0037] In order to facilitate the understanding of the present application, the present application will be described more fully below with reference to the accompanying drawings. The embodiments of the present application are shown in the drawings. However, the present application can be implemented in many different forms and is not limited to the embodiments described herein. On the contrary, the purpose of providing these embodiments is to make the disclosure of the present application more thorough and comprehensive.

[0038] Unless otherwise defined, all technical and scientific terms used herein have the same meaning as commonly understood by one of ordinary skill in the art to which the present application belongs. The terminology used in the specification of the present application is only for the purpose of describing specific embodiments of the present application and is not intended to limit the present application.

[0039] It can be understood that the terms "first", "second", and the like can be used herein to describe various elements, but these elements are not limited by these terms. These terms are only used to distinguish the first element from another element. For example, without departing from the scope of the present application, the first resistor can be referred to as the second resistor, and similarly, the second resistor can be referred to as the first resistor. The first resistor and the second resistor are both resistors, but they are not the same resistor. For example, in the embodiments of the present application, the first current value can be referred to as the second current value, and similarly, the second current value can be referred to as the first current value. The first current value and the second current value are both current values, but they are not the same current value.

[0040] It can be understood that, in the following embodiments, “connection” should be understood as “electrical connection”, “communication connection” and the like if the circuits, modules, units and the like connected with each other have transmission of electrical signals or data.

[0041] It can be understood that “at least one” means one or more, and “multiple” means two or more. “At least part of an element” means part or all of the element.

[0042] As used herein, the singular forms “a”, “an” and “the” can include plural forms unless the context clearly indicates otherwise. It should also be understood that the term “comprise / comprising” or “have / having” or the like specifies the presence of stated features, integers, steps, operations, components, parts, or combinations thereof, but does not exclude the presence or addition of one or more other features, integers, steps, operations, components, parts, or combinations thereof.

[0043] Reference Figure 1A A topology of a double-bridge series resonant conversion circuit is exemplarily provided for embodiments of the present application. Figure 1B It is a resonant type push-pull voltage doubling topology. Figure 1A For example, the double-bridge series resonant conversion circuit includes a primary bridge circuit, a secondary bridge circuit, a resonant network (including a transformer, a resonant inductor Ls and a resonant capacitor Cs), an output capacitor and an output resistor Ro. The input end of the primary bridge circuit is connected to an input power supply providing an input voltage, the input end of the resonant network is connected to the output end of the primary bridge circuit, in some exemplary embodiments, the primary winding of the transformer is connected to the output end of the primary bridge circuit, the input end of the secondary bridge circuit is connected to the output end of the resonant network, the output end of the secondary bridge circuit provides an output voltage for a load, in some exemplary embodiments, the two ends of the secondary winding of the transformer are respectively connected to the input end of the secondary bridge circuit through the resonant inductor Ls and the resonant capacitor Cs. Of course, the connection mode of the resonant inductor Ls and the resonant capacitor Cs is not limited to this, and the connection mode of the resonant inductor Ls and the resonant capacitor Cs is not specifically limited in the embodiments of the present application.

[0044] In some exemplary embodiments, the primary bridge circuit comprises a first upper switch Q1, a first lower switch Q2, a third upper switch Q4 and a third lower switch Q3. The first upper switch Q1 and the first lower switch Q2 are connected in series to form a first left bridge arm of the primary bridge circuit, and the third upper switch Q4 and the third lower switch Q3 are connected in series to form a first right bridge arm of the primary bridge circuit. The first left bridge arm and the first right bridge arm are connected in parallel. The two ends of the primary winding of the transformer are connected to the midpoint of the first left bridge arm and the midpoint of the first right bridge arm, respectively. The midpoint of the first left bridge arm is the connection point of the first upper switch Q1 and the first lower switch Q2, and the midpoint of the first right bridge arm is the connection point of the third upper switch Q4 and the third lower switch Q3.

[0045] In some exemplary embodiments, the secondary bridge circuit comprises a second upper switch Q5, a second lower switch Q6, a fourth upper switch Q8 and a fourth lower switch Q7. The second upper switch Q5 and the second lower switch Q6 are connected in series to form a second left bridge arm of the secondary bridge circuit, and the fourth upper switch Q8 and the fourth lower switch Q7 are connected in series to form a second right bridge arm of the secondary bridge circuit. The second left bridge arm and the second right bridge arm are connected in parallel. The two ends of the secondary winding of the transformer are connected to the midpoint of the second left bridge arm and the midpoint of the second right bridge arm, respectively. The midpoint of the second left bridge arm is the connection point of the second upper switch Q5 and the second lower switch Q6, and the midpoint of the first right bridge arm is the connection point of the fourth upper switch Q8 and the fourth lower switch Q7.

[0046] Figure 1C For Figure 1A , the DBSRC equivalent circuit of the double-bridge series resonant conversion circuit is an RC circuit. The driving signals of the above-mentioned switches and the circuit parameters are described below with reference to Figure 1A and Figure 2 . Vi represents the input voltage of the input end of the double-bridge series resonant conversion circuit, and Vo represents the output voltage of the output end of the double-bridge series resonant conversion circuit. The output end capacitor and the output end resistor Ro are connected in parallel to the output end of the double-bridge series resonant conversion circuit. The capacitor connected in parallel to each switch is the parasitic capacitance Coss of the switch. In the primary bridge circuit, the driving signals of the control ends of the first upper switch Q1 and the third lower switch Q3 are the same and are denoted as the first PWM driving signal g1. The waveform diagram of the first PWM driving signal g1 is shown in the first waveform diagram of Figure 2 . The driving signals of the control ends of the first lower switch Q2 and the third upper switch Q4 are the same and are denoted as the second PWM driving signal g2. The first PWM driving signal g1 and the second PWM driving signal g2 are complementary.

[0047] In the secondary bridge circuit, the driving signals of the control ends of the second upper switch Q5 and the fourth lower switch Q7 are the same and are denoted as the third PWM driving signal g3, and a waveform diagram of the third PWM driving signal g3 is shown in the second waveform diagram in Figure 2 ; and the external phase angle between the first PWM driving signal g1 and the third PWM driving signal g3 is θ. The driving signals of the control ends of the second lower switch Q6 and the fourth upper switch Q8 are the same and are denoted as the fourth PWM driving signal g4, and the third PWM driving signal g3 and the fourth PWM driving signal g4 drive complementarily. In the embodiment of the present application, the duty cycles of the PWM driving signals of the switches in the primary bridge circuit and the secondary bridge circuit are 50% after ignoring the dead time, and the internal phase angles of the primary bridge circuit and the secondary bridge circuit are both 0; the switching frequencies of the switches in the primary bridge circuit and the secondary bridge circuit are fs, and the switching frequency fs is a variable parameter.

[0048] In an exemplary embodiment, referring to Figure 1C and Figure 3 , a method for determining an external phase angle is provided, which is applied to, for example but not limited to, the exemplary schematic double-bridge series resonant conversion circuit Figure 1A and Figure 1B , and the method can include the following steps S301-S304.

[0049] S301, determining a first current value of the resonant inductor at a first time and a second current value of the resonant inductor at a second time based on a general solution of a time-domain differential equation of a resonant inductor current of a DBSRC equivalent circuit of the double-bridge series resonant conversion circuit, a general solution of a time-domain differential equation of a resonant capacitor voltage of the DBSRC equivalent circuit, and a symmetry of the DBSRC equivalent circuit; wherein the first time is a time when a first upper switch of a first left bridge arm of the primary bridge circuit starts to turn on, and the second time is a time when a second upper switch of a second left bridge arm of the secondary bridge circuit starts to turn on in a conduction process of the first upper switch.

[0050] As shown in Figure 2 , the first time t0 is a time when the first upper switch Q1 starts to turn on. The second time t1 is a time when the second upper switch Q5 starts to turn on in a conduction process of the first upper switch Q1. The third time t2 is a time when the first switch Q1 starts to turn off in a conduction process of the second upper switch Q5. The first time t0 and the second time t1 are the start time and the end time of a first time period (t0-t1) respectively, and the start time and the end time of a second time period (t1-t2) are the second time t1 and the third time t2 respectively.

[0051] and iL0 is a first current value of the resonant inductor Ls at a first time t0. iL1 is a second current value of the resonant inductor Ls at a second time t1. iL2 is a current value of the resonant inductor Ls at a third time t2. Uc0 is a first voltage value of the resonant capacitor Cs at the first time t0. Uc1 is a second voltage value of the resonant capacitor Cs at the second time t1. Uc2 is a voltage value of the resonant capacitor Cs at the third time t2.

[0052] S302, obtaining a first relationship of the phase shift angle with respect to the input voltage, the output voltage and the switching frequency of the switching tube of the double-bridge series resonant conversion circuit based on a first current value when the primary side bridge circuit meets the ZVS soft switching requirement.

[0053] S303, obtaining a second relationship of the phase shift angle with respect to the input voltage, the output voltage and the switching frequency based on a second current value when the secondary side bridge circuit meets the ZVS soft switching requirement.

[0054] S304, determining an expression of the phase shift angle with respect to the switching frequency, the steady-state voltage value and the input voltage according to the first relationship, the second relationship and the steady-state voltage value of the output end of the double-bridge series resonant conversion circuit.

[0055] In the embodiments of the present application, the ZVS soft switching is specifically zero voltage switching (ZVS).

[0056] Compared with the fundamental wave equivalence and the frequency domain analysis in the related art, the time domain method is used for formula derivation in the embodiments of the present application, which can ensure high accuracy of optimization. Compared with the method of DBSRC efficiency optimization in the related art, the physical measurement expression of the phase shift angle is obtained in the embodiments of the present application, which can adapt to different parameters of DBSRC. For the DBSRC circuit with different circuit parameters, the phase shift angle can be directly obtained through the known input voltage, the required output voltage and the switching frequency of the current circuit. The calculation of the phase shift angle is relatively simple, and does not need to be solved by using mathematical software such as matlab and mathcad. The portability is high in the DBSRC with different parameters, and the calculation can be performed in each control period by using single-chip microcomputer chips such as DSP and ARM, which ensures the real-time performance of circuit control. Moreover, the critical ZVS is used for optimization in the embodiments of the present application, which ensures the soft switching condition of the circuit through real-time calculation, reduces the calculation amount and improves the efficiency of the circuit.

[0057] In one exemplary embodiment, the above-mentioned step S301 can include the following steps S401-S404 in combination with reference to Figure 2 and Figure 4 The above-mentioned step S301 can include the following steps S401-S404.

[0058] S401, obtaining general solutions: obtaining a first general solution iL1(t) of a time-domain differential equation of a resonance inductance current of the DBSRC equivalent circuit in a first time period (t0-t1) and a second general solution iL2(t) of the time-domain differential equation of the resonance inductance current of the DBSRC equivalent circuit in a second time period (t1-t2), and obtaining a third general solution Uc1(t) of a time-domain differential equation of a resonance capacitance voltage of the DBSRC equivalent circuit in the first time period (t0-t1) and a fourth general solution Uc2(t) of the time-domain differential equation of the resonance capacitance voltage of the DBSRC equivalent circuit in the second time period (t1-t2).

[0059] In step S401, the first general solution iL1(t) and the third general solution Uc1(t) are both a first-order function of a first current value iL0 and a first voltage value Uc0 of the resonance capacitance voltage at a first time t0 as a first-type coefficient. The second general solution iL2(t) and the fourth general solution Uc2(t) are both a first-order function of a second current value iL1 and a second voltage value Uc1 of the resonance capacitance voltage at a second time t1 as a second-type coefficient.

[0060] It is also worth mentioning that the first general solution iL1(t) and the second general solution iL2(t) are both functions of the resonance inductance Ls at different times about the input voltage Vi, the output voltage Vo, and the phase shift angle θ. The third general solution Uc1(t) and the fourth general solution Uc2(t) are both functions of the resonance capacitance Cs at different times about the input voltage Vi, the output voltage Vo, and the phase shift angle θ.

[0061] Referring to Figure 2 , Figure 2 the third waveform diagram in FIG. 6 is a waveform diagram of the resonance inductance current I(Ls) about time t, Figure 2 the fourth waveform diagram in FIG. 6 is a waveform diagram of the resonance capacitance voltage V(Cs) about time t. Specifically, the first general solution iL1(t) and the second general solution iL2(t) correspond to the resonance inductance current about time t in the third waveform diagram in FIG. 6 in the first time period (t0-t1) and the second time period (t1-t2) respectively, and the third general solution Uc1(t) and the fourth general solution Uc2(t) correspond to the resonance inductance voltage about time t in the fourth waveform diagram in FIG. 6 in the first time period (t0-t1) and the second time period (t1-t2) respectively. Figure 2 the third waveform diagram in FIG. 6 is a waveform diagram of the resonance inductance current I(Ls) about time t, Figure 2 the fourth waveform diagram in FIG. 6 is a waveform diagram of the resonance capacitance voltage V(Cs) about time t. Specifically, the first general solution iL1(t) and the second general solution iL2(t) correspond to the resonance inductance current about time t in the third waveform diagram in FIG. 6 in the first time period (t0-t1) and the second time period (t1-t2) respectively, and the third general solution Uc1(t) and the fourth general solution Uc2(t) correspond to the resonance inductance voltage about time t in the fourth waveform diagram in FIG. 6 in the first time period (t0-t1) and the second time period (t1-t2) respectively.

[0062] In step S401, based on the DBSRC equivalent circuit, the first general solution iL1(t), the second general solution iL2(t), the third general solution Uc1(t), and the fourth general solution Uc2(t) can be directly obtained according to the principle of a second-order series RLC circuit. The first general solution iL1(t), the second general solution iL2(t), the third general solution Uc1(t), and the fourth general solution Uc2(t) are as follows:

[0063] The first general solution iL1(t) is:

[0064]

[0065] The second general solution iL2(t) is:

[0066]

[0067] The third general solution Uc1(t) is:

[0068]

[0069] The fourth general solution Uc2(t) is:

[0070]

[0071] In the above general solutions, Vi is an input voltage, Ls represents an inductance value of a resonance inductor, Cs represents a capacitance value of a resonance capacitor, Ntr is a turns ratio of a transformer of the double-bridge series resonance conversion circuit, Vo is an output voltage, cos is a cosine function, and sin is a sine function.

[0072] S402, obtaining general current and voltage values: obtaining a first general current value of the first general solution iL1(t) at a first time t0, a second general current value of the first general solution iL1(t) at a second time t1, a third general current value of the second general solution iL2(t) at a third time t2, a first general voltage value of the third general solution Uc1(t) at the first time t0, a second general voltage value of the third general solution Uc1(t) at the second time t1, and a third general voltage value of the fourth general solution Uc2(t) at the third time t2.

[0073] In step S402, in order to analyze one period, the first time t0 is set as a starting time, and the starting time is 0, that is, t0=0. The second time t1 is determined by the external phase angle θ and the switching frequency fs; specifically, when t0 is the starting time, the second time t1 is a time difference from the starting time, and the time difference corresponds to a time period value corresponding to the angle of the external phase angle θ. That is, when t0=0, t1=θ / (2πfs), and the third time t2 is T / 2=1 / (2fs), and T is one switching period.

[0074] Thus, t=t0 is substituted into the above first general solution iL1(t) to obtain the first general current value iL1(t=t0) of the first general solution iL1(t) at the first time t0. Since the starting time t0=0, iL0 is the first current value of the resonance inductor Ls at the first time t0, as shown above, and combined with the waveform diagram Figure 2It can be seen that iL1(t=t0=0)=iL0. Similarly, substituting t=t1 into the first general solution iL1(t) above, the second general current value iL1(t=t1) of the first general solution iL1(t) at the second time t1 is obtained, as described above, iL1 is the second current value of the resonant inductor Ls at the second time t1, and combined with the waveform diagram Figure 2 It can be seen that iL1(t=t1)=iL1. Similarly, substituting t=t2 into the second general solution iL2(t) above, the third general current value iL2(t=t2) of the second general solution iL2(t) at the third time t2 is obtained, as described above, iL2 is the current value of the resonant inductor Ls at the third time t2, and combined with the waveform diagram Figure 2 It can be seen that iL2(t=t2)=iL2.

[0075] Similarly, substituting t=t1 into the third general solution Uc1(t) above, the second general voltage value Uc1(t=t1) of the third general solution Uc1(t) at the second time t1 is obtained, as described above, Uc1 is the second voltage value of the resonant capacitor Cs at the second time t1, and combined with the waveform diagram Figure 2 It can be seen that Uc1(t=t0=0)=Uc0. Similarly, substituting t=t1 into the third general solution Uc1(t) above, the second general voltage value Uc1(t=t1) of the third general solution Uc1(t) at the second time t1 is obtained, as described above, Uc1 is the second voltage value of the resonant capacitor Cs at the second time t1, and combined with the waveform diagram Figure 2 It can be seen that Uc1(t=t1)=Uc1. Similarly, substituting t=t2 into the fourth general solution Uc2(t) above, the third general voltage value Uc2(t=t2) of the fourth general solution Uc2(t) at the third time t2 is obtained, as described above, Uc2 is the voltage value of the resonant capacitor Cs at the third time t2, and combined with the waveform diagram Figure 2 It can be seen that Uc2(t=t2)=Uc2.

[0076] S403, determine the symmetry condition of the resonant inductor current of the DBSRC equivalent circuit and the symmetry condition of the resonant capacitor voltage of the DBSRC equivalent circuit: the third general current value iL2(t=t2) and the first general current value iL1(t=t0=0) are opposite numbers, and the third general voltage value Uc2(t=t2) and the first general voltage value Uc1(t=t0=0) are opposite numbers.

[0077] That is, iL2(t=t2)=-iL1(t=t0), that is, iL2=-iL0.

[0078] Similarly, Uc2(t=t2)=-Uc1(t=t0), that is, Uc2=-Uc0.

[0079] S404, according to the general solution current voltage value and symmetry condition, multiple multiple linear equations are obtained by simultaneous, and the first current value iL0 of the first coefficient value, the first voltage value Uc0 and the second current value iL1 of the first coefficient value, the second voltage value Uc1 of the second coefficient value are obtained by solving the equation.

[0080] The first current value iL0 of the first coefficient value is:

[0081]

[0082] The second current value iL1 of the second coefficient value is:

[0083]

[0084] The first voltage value Uc0 of the first coefficient value is:

[0085]

[0086] The second voltage value Uc1 of the second coefficient value is:

[0087]

[0088] Wherein, in the current voltage value of each coefficient value, Vi is the input voltage, Ls represents the inductance value of the resonance inductance, Cs represents the capacitance value of the resonance capacitance, Ntr is the turns ratio of the transformer of the double-bridge series resonance conversion circuit, Vo is the output voltage, fs is the switching frequency, θ is the external phase angle, tan is the tangent function, sin is the sine function, and cos is the cosine function.

[0089] In the embodiment of the application, the time domain method is used for formula derivation, which can ensure high accuracy of optimization. Compared with the method for optimizing the efficiency of DBSRC in the related art, the embodiment of the application obtains the general solution, obtains the general solution current voltage value, determines the symmetry condition, and obtains multiple multiple linear equations by simultaneous, thereby obtaining the physical measurement expression of the external phase angle, which can adapt to DBSRC with different parameters, has high portability, and the calculation of the external phase angle is relatively simple, which can be applied to the calculation of DSP, ARM and other single-chip microcomputers in each control period, thereby ensuring the real-time performance of circuit control. Moreover, the critical ZVS is used for optimization in the embodiment of the application, the soft switching condition of the circuit is ensured by real-time calculation, the calculation amount is reduced, and the efficiency of the circuit is improved.

[0090] In one exemplary embodiment, with reference to Figure 5 , step S302 can include steps S501-S502.

[0091] S501, obtaining a first equation that a first current value iL0 satisfies when a critical condition of the primary side bridge circuit satisfying the ZVS soft switching requirement is met: iL0=0.

[0092] Wherein, when the parasitic capacitance Coss of the switch tube is small enough, the first current value iL0 of the primary side bridge circuit satisfying the ZVS soft switching requirement is less than 0, that is:

[0093]

[0094] Normalized phase shift angle (0<D<1), resonant frequency All parameters in the above formula (1) are greater than 0, so according to the above formula (1), it can be deduced that:

[0095] (2)

[0096] S502, obtaining a first relationship according to the first equation.

[0097] That is, when the critical condition of the primary side bridge circuit satisfying the ZVS soft switching requirement is met, that is, iL0=0, that is, the inequality (2) is equal to 0, and thus the first relationship is obtained:

[0098] ,

[0099] In the above first relationship, Dmin is the minimum normalized outer phase shift angle, fr is the resonant frequency, fs is the switching frequency, Vi is the input voltage, Vo is the output voltage, Ntr is the turns ratio of the transformer of the double-bridge series resonant conversion circuit, sin is the sine function, and asin is the inverse sine function.

[0100] In the embodiment of the application, the minimum normalized value Dmin of the outer phase shift angle is calculated based on the above first relationship, so that all power tubes in the circuit can realize ZVS opening, thereby improving the efficiency of the circuit.

[0101] In an exemplary embodiment, with reference to Figure 6 , step S302 can further include steps S601-S602.

[0102] S601, obtaining a second equation that a second current value iL1 satisfies when a critical condition of the secondary side bridge circuit satisfying the ZVS soft switching requirement is met: iL1=0.

[0103] Wherein, when the parasitic capacitance Coss of the switch tube is small enough, the first current value iL1 of the secondary side bridge circuit satisfying the ZVS soft switching requirement is greater than 0, that is:

[0104] (3)

[0105] Normalized phase-shift angle (0 < D < 1), resonant frequency All parameters in the above formula (3) are greater than 0, and thus, according to the above formula (3), it can be deduced that:

[0106] (4)

[0107] S602, a second relationship is obtained according to a second equation.

[0108] That is, when the critical condition of the secondary bridge circuit satisfying the ZVS soft switching requirement is iL1=0, that is, the inequality (4) is equal to 0, and thus the second relationship is obtained as:

[0109] , ;

[0110] In the above second relationship, Dmin is the minimum normalized outer phase-shift angle, fr is the resonant frequency, fs is the switching frequency, Vi is the input voltage, Vo is the output voltage, Ntr is the turns ratio of the transformer of the double-bridge series resonant conversion circuit, sin is the sine function, and asin is the inverse sine function.

[0111] In the embodiments of the present application, the minimum normalized value Dmin of the outer phase-shift angle is calculated based on the above second relationship, which can make all power tubes in the circuit achieve ZVS turn-on, thereby improving the efficiency of the circuit.

[0112] In an exemplary embodiment, the steady-state voltage value Vo_set refers to the output voltage of the resonant conversion circuit provided to the load. For simplicity of calculation, ZVS is optimized only after the circuit is in a steady state (the circuit enters a steady state when the output voltage Vo reaches the set steady-state voltage value Vo_set after PI control, which is when the circuit enters a steady state). At this time, Vo=Vo_set, and Vo_set is the set output voltage of the DAB circuit required, based on the first relationship and the second relationship, the expression of the outer phase-shift angle with respect to the switching frequency, the steady-state voltage value, and the input voltage is determined, including:

[0113] , _set

[0114] , _set

[0115]

[0116] (5)

[0117] Wherein, M can be understood as an intermediate quantity, used to replace the relationship between the input voltage, the steady-state voltage set value and the turns ratio; Dmin is the minimum normalized phase shift angle, fr is the resonant frequency, fs is the switching frequency, Vi is the input voltage, Vo_set is the steady-state voltage value, Ntr is the turns ratio of the transformer of the double-bridge series resonant conversion circuit, sin is the sine function, asin is the inverse sine function; The parameter is related to both the resonant frequency fr and the switching frequency fs.

[0118] In summary, after the circuit enters the steady state, that is, when the phase shift angle D > Dmin, all the switching tubes can meet the requirement of ZVS.

[0119] The scheme of the embodiment of the present application calculates the minimum normalized value Dmin of the phase shift angle based on formula (5), which can make all the power tubes in the circuit realize ZVS turn-on, thereby improving the efficiency of the circuit (because the derivation process of formula (5) needs to meet the requirement of ZVS soft switching, so the transmission efficiency of the circuit is improved). The phase shift angle in formula (5) is only related to the output gain and is independent of the output load (i.e., the power of the circuit), thereby reducing the complexity of calculation and reducing the dependence of control on the accuracy of current sampling. The calculation of the phase shift angle is relatively simple, and a single-chip microcomputer chip such as DSP or ARM can be used to calculate it in each control period, thereby ensuring the real-time performance of the circuit control.

[0120] Based on the above, the embodiment of the present application can determine a first expression of the output power Po of the output end of the double-bridge series resonant conversion circuit based on the piecewise calculus of the first general solution iL1(t) and the second general solution iL2(t), and determine the gain of the output voltage Vo of the output end according to the first expression of the output power Po and a second expression of the output power Po of the output end determined based on Ohm's law.

[0121] The first expression of the output power Po is:

[0122]

[0123] The first expression of the output power Po is simplified as:

[0124]

[0125] The second expression of the output power Po is:

[0126]

[0127] Based on the fact that the output power based on the first expression is equal to the output power based on the second expression, it can be deduced that the gain of the output voltage Vo is:

[0128] (6)

[0129] After the circuit enters steady state, i.e. when the external phase shift angle D > Dmin, all the switch tubes can meet the requirement of ZVS, and the gain of the output voltage Vo is simplified as:

[0130] (7)

[0131] Since M is a constant less than 1, so When fs > fr, 0 < θr < π / 2, according to the simplified formula (7) of the gain of the output voltage Vo, Vo monotonically decreases with fs, so the steady-state voltage value Vo_set can be obtained by closed-loop adjustment of fs, which indicates the feasibility of realizing the output gain of the circuit through closed-loop control.

[0132] Based on the same inventive concept, in one exemplary embodiment, in combination with Figure 7 and Figure 8 , a variable frequency phase shift control method is provided, which is applied to, for example but not limited to Figure 1A and Figure 1B an exemplary schematic double-bridge series resonant conversion circuit, which can include the following steps S701~S705.

[0133] S701, sampling the output end of the double-bridge series resonant conversion circuit to obtain the actual voltage value of the output end at the current time.

[0134] S702, based on the actual voltage value and the PI control principle, obtaining the first switching frequency of the switch tube when the voltage of the output end reaches the steady-state voltage value.

[0135] S703, based on the first switching frequency fs1 and the preset maximum switching frequency fs_max, obtaining the required switching frequency fs of the double-bridge series resonant conversion circuit in the variable frequency phase shift control process.

[0136] S704, according to the required switching frequency, the input voltage of the input end of the double-bridge series resonant conversion circuit at the current time, the steady-state voltage value of the output end, and the expression based on the determination method of the external phase shift angle, obtaining the required external phase shift angle in the variable frequency phase shift control process at the current time.

[0137] S705, according to the preset duty cycle value of the switch tube, the required switching frequency and the required external phase shift angle, obtaining the PWM driving signal of the switch tube of the primary bridge circuit and the PWM driving signal of the switch tube of the secondary bridge circuit.

[0138] wherein, Figure 8Vout is the actual voltage value of the output voltage of the output terminal of the DBSRC collected, Vo-set is the set steady-state voltage value of the output terminal of the DBSRC, the output voltage Vout of the DBSRC is subjected to PI control, so that the output voltage Vout of the DBSRC reaches Vo-set, a first switching frequency fs1 is obtained, and the maximum switching frequency fs_max allowed by the DBSRC is subtracted by the first switching frequency fs1 to obtain the required switching frequency fs in the variable frequency phase shift control of the DBSRC. The input voltage Vi of the DBSRC and the set steady-state voltage value Vo-set of the DBSRC are brought into the above formula (5) to obtain the minimum normalized value Dmin of the outer phase shift angle θ, and Dmin is added by a proper angle margin d0 to compensate for the effects of dead time and parasitic capacitance Coss and the like, to obtain the required outer phase shift angle D of the outer phase shift angle θ in the variable frequency phase shift control. Optionally, the duty cycle of all switching tubes can be set to 50%, and according to the switching frequency fs of the switching tube and the required outer phase shift angle D, the PWM drive signals of the switching tubes Q1, Q2, Q3 and Q4 of the primary side of the transformer of the DBSRC and the switching tubes Q5, Q6, Q7 and Q8 of the secondary side of the transformer can be obtained.

[0139] In addition, when the output load is changed to a power supply, the DBSRC circuit supports bidirectional power transmission: when D<1, power is transmitted from the output to the output, and when D>1, power is transmitted from the output to the input.

[0140] Optionally, referring to Figure 8 , step S704 can include: inputting the required switching frequency, the input voltage at the current time and the steady-state voltage value into the expression to obtain the initial outer phase shift angle Dmin; and obtaining the required outer phase shift angle D according to the sum of the initial outer phase shift angle Dmin and a preset angle value d0.

[0141] In the embodiment of the application, the PI closed-loop control is used to substitute the required switching frequency fs obtained by the output into the above formula (5) to calculate the minimum phase shift angle Dmin (i.e. the initial outer phase shift angle Dmin), and Dmin is added by a proper angle margin d0 to compensate for the effects of dead time and parasitic capacitance Coss and the like, wherein d0 is an experimental test value, such as 0.02, and the size of d0 is not limited in the embodiment of the application. According to the required switching frequency fs and the required outer phase shift angle D, the required PWM drive signal can be generated to make the circuit meet the requirements of output gain and ZVS.

[0142] Exemplarily, the technical solutions of the embodiments of the application are simulated as follows:

[0143] The circuit parameters of a DBSRC circuit C are: the inductance value of the resonant inductor Ls=64uH, the capacitance value of the resonant capacitor Cs=80nF, the turn ratio of the transformer Ntr=1 / 8, the output resistance of the DBSRC circuit output end Ro=53.3Ω, and the steady-state voltage value Vo_set=400V. A closed-loop control model is built according to the scheme of the embodiment of the application, and the simulation waveform is as shown in Figure 9 .

[0144] As can be seen from Figure 9 , when Vi<Ntr×Vo_set, iL0≈0 and iL1>0, that is, all the switch tubes can be turned on with ZVS, and the primary-side switch tube is critical ZVS; when Vi≥Ntr×Vo_set, iL0<0 and iL1≈0, that is, all the switch tubes can be turned on with ZVS, and the secondary-side switch tube is critical ZVS.

[0145] Based on the above parameters, the function curve of the output gain Vo×Ntr / Vi versus the switching frequency fs is as shown in Figure 10 , Figure 10 where the horizontal axis is the frequency, and the vertical axis is the output voltage gain. The steady-state voltage Vo_set at the output end of the resonant conversion circuit in the above formula is a fixed value, so M is a constant. Wherein, Figure 10 the first function curve GM(fs) in the above formula (7) and the parameters of the DBSRC circuit C is drawn according to the above output voltage gain formula (7) and the parameters of the DBSRC circuit C, Figure 10 the second function curve G1(fs) in the above formula (6) and the expression of the phase angle of the embodiment of the application is drawn according to the above output voltage gain formula (6), the expression of the phase angle of the embodiment of the application and the parameters of the DBSRC circuit C, Figure 10 and the third function curve G2(fs) in the above formula (6) and the first relationship of the phase angle is drawn according to the above output voltage gain formula (6), the first relationship of the phase angle and the parameters of the DBSRC circuit C.

[0146] As can be known from Figure 10 , the second function curve G1(fs) is a monotonically decreasing function of the output voltage gain with respect to the switching frequency fs, and the third function curve G2(fs) is a monotonically increasing function of the output voltage gain with respect to the switching frequency fs, that is, G is not monotonous in the full gain range and needs to be regulated in sections. The first function curve GM(fs) is a monotonically decreasing function of the output voltage gain with respect to the switching frequency fs, so a single PI can be used for regulation, and the PI control regulation is more convenient. Figure 9 When the switching frequency changes from 70kHz to 150kHz in the above formula (7), the output gain GM is regulated between 0.25 and 1.5, which can meet the design requirements of a wide range of DC-DC of lithium batteries and the like.

[0147] Based on the same inventive concept, the embodiment of the present application also provides an outward moving phase angle determination system, which comprises:

[0148] The double-bridge series resonant conversion circuit comprises a primary bridge circuit, a resonant network and a secondary bridge circuit; an input end of the primary bridge circuit is connected to an input power supply providing an input voltage, an input end of the resonant network is connected to an output end of the primary bridge circuit, an input end of the secondary bridge circuit is connected to an output end of the resonant network, and an output end of the secondary bridge circuit provides an output voltage for a load; the resonant network comprises a transformer, a resonant inductor and a resonant capacitor;

[0149] The execution unit is configured to collect an input voltage and an output voltage at the output end of the secondary bridge circuit, and to execute the steps of the outward moving phase angle determination method of any of the embodiments when the output voltage reaches a set steady-state voltage value.

[0150] The outward moving phase angle determination system provided by the embodiment of the present application and the outward moving phase angle determination method of any of the above embodiments belong to the same inventive concept, can solve the same technical problem, and achieve the same technical effect, and repeated contents will not be described here.

[0151] Based on the same inventive concept, the embodiment of the present application also provides a computer chip comprising a memory and a processor, wherein the memory stores a computer program, and the processor executes the steps of the outward moving phase angle determination method of any of the embodiments when executing the computer program.

[0152] Based on the same inventive concept, the embodiment of the present application also provides a computer chip comprising a memory and a processor, wherein the memory stores a computer program, and the processor executes the steps of the frequency conversion phase control method of any of the embodiments when executing the computer program.

[0153] Based on the same inventive concept, the embodiment of the present application also provides a computer device comprising a memory and a processor, wherein the memory stores a computer program, and the processor executes the steps of the outward moving phase angle determination method of any of the embodiments when executing the computer program.

[0154] Based on the same inventive concept, the embodiment of the present application also provides a computer device comprising a memory and a processor, wherein the memory stores a computer program, and the processor executes the steps of the frequency conversion phase control method of any of the embodiments when executing the computer program.

[0155] Based on the same inventive concept, the embodiment of the present application also provides a computer readable storage medium, which stores a computer program, and the computer program is executed by a processor to implement the steps of the outward moving phase angle determination method of any of the embodiments.

[0156] Based on the same inventive concept, the embodiment of the present application further provides a computer readable storage medium, which stores a computer program, and the computer program is executed by a processor to implement the steps of the variable frequency phase shift control method of any embodiment.

[0157] Based on the same inventive concept, the embodiment of the present application further provides a computer program product, which comprises a computer program, and the computer program is executed by a processor to implement the steps of the determination method of the external phase angle of any embodiment.

[0158] Based on the same inventive concept, the embodiment of the present application further provides a computer program product, which comprises a computer program, and the computer program is executed by a processor to implement the steps of the variable frequency phase shift control method of any embodiment.

[0159] In the description of the present specification, the description referring to the terms "some embodiments", "other embodiments", and the like means that the specific features, structures, materials or characteristics described in connection with the embodiments or examples are contained in at least one embodiment or example of the present application. In the present specification, the illustrative description of the above terms does not necessarily mean the same embodiment or example.

[0160] The technical features of the above-described embodiments can be combined in any manner. In order to make the description concise, all possible combinations of the technical features in the above-described embodiments are not described, however, as long as the combinations of the technical features do not contradict each other, they should be considered as the scope of the present application.

[0161] The above-described embodiments only express several implementation manners of the present application, and the description is more specific and detailed, but it should not be understood as the limitation of the scope of the present application. It should be pointed out that, for those skilled in the art, without departing from the concept of the present application, a number of modifications and improvements can be made, which are all within the protection scope of the present application. Therefore, the protection scope of the present application should be subject to the appended claims.

Claims

1. A method for determining an external phase angle, applied to a dual-bridge series resonant converter circuit, characterized in that: The method comprises: Determining a first current value of the resonant inductor at a first moment and a second current value of the resonant inductor at a second moment based on a general solution of a time-domain differential equation of a resonant inductor current in a DBSRC equivalent circuit of the dual-bridge series resonant converter circuit, a general solution of a time-domain differential equation of a resonant capacitor voltage in the DBSRC equivalent circuit, and the symmetry of the DBSRC equivalent circuit; Based on the first current value when the primary bridge circuit meets the ZVS soft switching requirement, a first relationship between the external phase shift angle and the input voltage, output voltage and switching frequency of the dual-bridge series resonant conversion circuit is obtained; Obtaining a second relationship between the external phase shift angle and the input voltage, the output voltage, and the switching frequency based on the second current value when the secondary bridge circuit meets the ZVS soft switching requirement; Determine an expression for the external phase shift angle with respect to the switching frequency, the steady-state voltage value, and the input voltage based on the first relational expression, the second relational expression, and the steady-state voltage value at the output end of the dual-bridge series resonant converter circuit; Among them, the first moment is the moment when the first upper switch tube of the first left bridge arm of the primary bridge circuit starts to turn on, and the second moment is the moment when the second upper switch tube of the second left bridge arm of the secondary bridge circuit starts to turn on during the conduction process of the first upper switch tube.

2. The method according to claim 1, characterized in that Determining a first current value of the resonant inductor at a first moment and a second current value of the resonant inductor at a second moment based on a general solution of a time-domain differential equation of a resonant inductor current of a DBSRC equivalent circuit of the dual-bridge series resonant conversion circuit, a general solution of a time-domain differential equation of a resonant capacitor voltage of the DBSRC equivalent circuit, and the symmetry of the DBSRC equivalent circuit includes: Obtaining a general solution: obtaining a first general solution of the time-domain differential equation of the resonant inductor current of the DBSRC equivalent circuit in the first time period and a second general solution in the second time period, respectively; obtaining a third general solution of the time-domain differential equation of the resonant capacitor voltage of the DBSRC equivalent circuit in the first time period and a fourth general solution in the second time period, respectively; wherein the first general solution and the third general solution are both linear functions of first-type coefficients with the first current value and the first voltage value of the resonant capacitor voltage at the first moment, and the second general solution and the fourth general solution are both linear functions of second-type coefficients with the second current value and the second voltage value of the resonant capacitor voltage at the second moment, respectively; the first moment and the second moment are the starting moment and the ending moment of the first time period, respectively; the starting moment of the second time period is the second moment; the ending moment of the second time period is recorded as the third moment; and the third moment is the moment when the first upper switch tube begins to turn off during the conduction process of the second upper switch tube; Obtaining through-current current and voltage values: respectively obtaining a first through-current value of the first through-current formula at a first moment, a second through-current value of the first through-current formula at a second moment, a third through-current value of the second through-current formula at a third moment, a first through-current voltage value of the third through-current formula at a first moment, a second through-current voltage value of the third through-current formula at a second moment, and a third through-current voltage value of the fourth through-current formula at a third moment, wherein the first moment t0 is a starting moment t0=0, the second moment t1 is determined by the external phase shift angle and the switching frequency, and the third moment t2 is T / 2, where T is one switching cycle; Determining a symmetry condition of the resonant inductor current of the DBSRC equivalent circuit and a symmetry condition of the resonant capacitor voltage of the DBSRC equivalent circuit: the third on-state current value and the first on-state current value are inverse numbers of each other, and the third on-state voltage value and the first on-state voltage value are inverse numbers of each other; According to the general solution current and voltage values ​​and the symmetry condition, a plurality of multivariate linear equations are obtained simultaneously, and the first type coefficient values ​​and the second type coefficient values ​​are obtained by solving the equations.

3. The method according to claim 2, characterized in that The first general solution and the second general solution are both inductor current functions of the resonant inductor with respect to the input voltage, the output voltage and the external phase shift angle at different times; The third general solution and the fourth general solution are both capacitance-voltage functions of the resonant capacitor with respect to the input voltage, the output voltage, and the external shift phase angle at different times.

4. The method according to claim 2, characterized in that The first current value iL0 of the first type of coefficient value is: ； The second current value iL1 of the second type coefficient value is: ； Among them, Vi is the input voltage, Ls represents the inductance of the resonant inductor, Cs represents the capacitance of the resonant capacitor, Ntr is the turns ratio of the transformer of the dual-bridge series resonant conversion circuit, Vo is the output voltage, fs is the switching frequency, θ is the external shift phase angle, tan is the tangent function, sin is the sine function, and cos is the cosine function.

5. The method according to any one of claims 1 to 4, It is characterized by: in, The first current value when the primary bridge circuit meets the ZVS soft switching requirement is used to obtain a first relationship between the external phase shift angle and the input voltage, output voltage, and switching frequency of the dual-bridge series resonant conversion circuit, including: Obtaining a first equation that the first current value iL0 satisfies when the primary bridge circuit meets a critical condition for ZVS soft switching requirements: iL0=0; According to the first equation, the first relationship is obtained as follows: ， ； The second current value when the secondary bridge circuit meets the ZVS soft switching requirement is used to obtain a second relationship between the external phase shift angle and the input voltage, the output voltage, and the switching frequency, including: Obtaining a second equation that the second current value iL1 satisfies when the secondary bridge circuit meets the critical condition of the ZVS soft switching requirement: iL1=0; According to the second equation, the second relational expression is obtained as follows: ， ； Among them, Dmin is the minimum normalized external phase shift angle, fr is the resonant frequency, fs is the switching frequency, Vi is the input voltage, Vo is the output voltage, Ntr is the turns ratio of the transformer of the dual-bridge series resonant conversion circuit, sin is the sine function, and asin is the inverse sine function.

6. The method according to claim 5, characterized in that When the output voltage is a set steady-state voltage value, determining an expression of the external phase shift angle with respect to the switching frequency, the steady-state voltage value, and the input voltage based on the first relationship and the second relationship includes: ； ； ， ； ， ； Wherein, Vo_set is the steady-state voltage value.

7. A variable frequency phase shift control method, applied to a dual-bridge series resonant converter circuit, characterized in that: The control method includes: Sampling the output end of the dual-bridge series resonant conversion circuit to obtain an actual voltage value of the output end at a current moment; Based on the actual voltage value and the PI control principle, obtaining a first switching frequency of the switch tube when the voltage at the output end reaches a steady-state voltage value; Based on the first switching frequency and a preset maximum switching frequency, obtaining a required switching frequency of the dual-bridge series resonant conversion circuit during a variable frequency phase shift control process; Obtaining the required external phase shift angle in the variable frequency phase shift control process at the current moment based on the required switching frequency, the input voltage at the input end of the dual-bridge series resonant converter circuit at the current moment, the steady-state voltage value of the output end, and the expression based on the method for determining the external phase shift angle according to any one of claims 1 to 6; According to the preset duty cycle value of the switch tube, the required switching frequency and the required external shift phase angle, the PWM drive signal of the switch tube of the primary bridge circuit and the PWM drive signal of the switch tube of the secondary bridge circuit are obtained.

8. The method according to claim 7, characterized in that Obtaining the required external phase shift angle in the variable frequency phase shift control process at the current moment according to the required switching frequency, the input voltage at the current moment, the steady-state voltage value of the output terminal, and the expression, including: Inputting the required switching frequency, the input voltage at the current moment, and the steady-state voltage value into the expression to obtain an initial external phase shift angle; The required external phase shift angle is obtained according to the sum of the initial external phase shift angle and a preset angle value.

9. A system for determining an external phase angle, characterized in that: include: A double-bridge series resonant converter circuit, comprising a primary-side bridge circuit, a resonant network and a secondary-side bridge circuit; The input end of the primary bridge circuit is connected to an input power supply providing an input voltage, the input end of the resonant network is connected to the output end of the primary bridge circuit, the input end of the secondary bridge circuit is connected to the output end of the resonant network, and the output end of the secondary bridge circuit provides an output voltage to a load; The resonant network includes a transformer, a resonant inductor and a resonant capacitor; An execution unit is used to collect the input voltage and the output voltage of the output end of the secondary bridge circuit, and is also used to execute the steps of the method according to any one of claims 1 to 6 when the output voltage reaches a set steady-state voltage value.

10. A computer chip comprising a memory and a processor, wherein the memory stores a computer program, wherein: When the processor executes the computer program, the steps of the method according to any one of claims 1 to 8 are implemented.

Images