Wireless power transmission anti-offset efficiency optimization method for combined SCC compensation

Abstract

The invention discloses a wireless power transmission anti-offset efficiency optimization method for combined SCC compensation. The method comprises the following steps: step 1, constructing a topological circuit of a magnetic coupling wireless power transmission system; step 2, acquiring output voltage Vdc of the direct-current power supply, outputting the output voltage Vdc to a primary side high-frequency inverter module, and outputting 85kHz alternating-current square waves to a primary side coil by a high-frequency inverter; step 3, calculating loop current and input impedance; 4, giving the phase difference between the resonant frequency of the system and the output voltage and current of the inverter, and constructing an input impedance angle under the control of the compensation parameter; step 5, controlling a parallel single-tube controllable switch capacitor C1 of the primary side to enable the capacitor C1 to resonate with a compensation inductor, controlling a series full-wave controllable switch capacitor Cp of the primary side to enable an input impedance angle to be close to zero and greater than zero, enabling a circuit to be inductive, realizing an inverter ZVS and improving the overall transmission efficiency of the system; the system has the characteristics of continuous adjustment of series-parallel compensation, good anti-offset performance, small loss of the parallel SCC switch and low system cost.

Description

Technical Field

[0001] This invention belongs to the field of wireless power transmission, and particularly relates to a method for optimizing the anti-migration efficiency of wireless power transmission using combined SCC compensation. Background Technology

[0002] Wireless power transfer technology, as a breakthrough in energy transmission methods that overcomes traditional wire connections, is becoming a global research hotspot in the technology field due to its contactless, highly secure, and easy-to-deploy characteristics. Magnetic coupling wireless power transfer achieves energy transfer through magnetic field resonant coupling, demonstrating irreplaceable advantages in scenarios such as dynamic charging of electric vehicles, power supply for smart medical devices, and industrial IoT sensor networks. In the field of new energy vehicles, dynamic wireless charging technology can achieve continuous power supply during vehicle operation, completely changing the reliance on traditional charging piles and promoting the construction of a "cableless" transportation ecosystem. In the medical field, wireless power supply for implantable medical devices can avoid the infection risks associated with wire implantation and extend the lifespan of the devices. However, this technology faces complex engineering challenges in practical applications. One of the core issues is the problem of resonance detuning under dynamic operating conditions. When the transmission distance fluctuates, the load changes dynamically, or environmental electromagnetic interference occurs, the self-inductance and mutual inductance parameters of the coupling mechanism will drift, causing the system resonant frequency to deviate from the optimal operating point, leading to problems such as a sharp drop in transmission efficiency and unstable power output, which seriously restricts its reliability in dynamic scenarios.

[0003] While using compensated topology parameter design can ensure high efficiency and stable output of the WPT system to a certain extent, its effectiveness is limited when coil offset resistance or soft switching is required. When the transmission distance changes, appropriate control strategies are needed to maintain the WPT system at maximum transmission efficiency. Controllable switched capacitor technology, due to its compact structure and flexible adjustment, has become a key solution to offset resonance detuning problems. By adjusting the combination state or charging / discharging time of the capacitor array, this technology can dynamically adjust the equivalent capacitance value of the compensation network, thereby tracking changes in the system's resonant frequency. For example, in an LCC-S type compensated topology, the resonant state can be controlled by adjusting the parallel compensation capacitor through switching capacitors. However, using dual switching transistors as parallel compensation switching capacitors without zero-crossing detection can lead to a surge in circulating current stress on the switching transistors, increasing the complexity of capacitor parameter tuning. Differentiated configurations of switched capacitor arrays offer simplicity and versatility, but their discrete adjustment limits adjustment accuracy and increases device cost and integration difficulty, thus presenting certain limitations. Summary of the Invention

[0004] To overcome the shortcomings of the prior art, the present invention aims to provide a method for optimizing the anti-offset efficiency of wireless power transmission with combined SCC compensation, which aims to solve the technical problem of low coil offset efficiency and stability in wireless power transmission systems. It features continuous adjustment of series and parallel compensation, good anti-offset performance, low switching loss of parallel SCC, and low system cost.

[0005] To achieve the above objectives, the technical solution adopted by the present invention is as follows:

[0006] A method for optimizing the anti-migration efficiency of wireless power transfer using combined SCC compensation includes the following steps:

[0007] Step 1: Construct the topology of the magnetically coupled wireless power transfer system;

[0008] Step 2, obtain the output voltage V of the DC power supply. dc It is then output to the primary-side high-frequency inverter module, which outputs an 85kHz AC square wave to the primary-side coil.

[0009] Step 3: Calculate the loop current and input impedance using the primary side compensation parameters and reflection impedance;

[0010] Step 4: Given the system resonant frequency and the phase difference between the inverter output voltage and current, construct the input impedance angle under the control of compensation parameters;

[0011] Step 5: By controlling the parallel single-tube controllable switching capacitor C1 on the primary side to make it resonate with the compensation inductor, the series full-wave controllable switching capacitor C on the primary side is controlled. p By controlling the input impedance angle to be close to zero and greater than zero, the circuit becomes inductive, achieving ZVS in the inverter and improving the overall transmission efficiency of the system.

[0012] The circuit topology of the magnetically coupled wireless power transfer system described in step 1 further includes:

[0013] DC power supply, primary-side high-frequency inverter module, transmitting coil module, receiving coil module, secondary-side rectifier module, load, inverter PWM drive module, switched capacitor PWM drive module, controller, primary-side parallel switched capacitor and primary-side series switched capacitor;

[0014] The DC power supply is connected to the input of the primary-side high-frequency inverter module. The output of the high-frequency inverter is connected to the input of the transmitting coil module. The output of the transmitting coil module is positioned opposite to the input of the receiving coil module. The output of the receiving coil module is connected to the input of the secondary-side rectifier module. The output of the secondary-side rectifier module is connected to the system load. The output of the controller is connected to the input of the inverter PWM drive module and the input of the switched-capacitor PWM drive module. The output of the inverter PWM drive module is connected to the input of the primary-side high-frequency inverter module. The output of the switched-capacitor PWM drive module is connected to the input of the primary-side parallel switched-capacitor and the primary-side series switched-capacitor. The controller outputs control signals to the inverter PWM drive module and the switched-capacitor PWM drive module. The inverter PWM drive module outputs PWM signals to the primary-side high-frequency inverter module. The switched-capacitor PWM drive module outputs PWM signals to the primary-side parallel switched-capacitor and the primary-side series switched-capacitor. The primary-side parallel switched-capacitor is connected in parallel in the primary-side LCC topology, and the primary-side series switched-capacitor is connected in series in the LCC topology as a compensation capacitor.

[0015] The primary side DC power supply is controlled to obtain a suitable output voltage. The high-frequency voltage is obtained by the primary side high-frequency inverter through PWM control. The voltage is then passed through the topology compensation coil structure to the output of the transmitting coil module and then to the input voltage of the receiving coil module. The secondary side rectifier module converts the high-frequency AC power into DC power. The secondary side load is connected to the output terminal of the secondary side rectifier module to transmit power.

[0016] Primary-side parallel switched capacitor, primary-side series switched capacitor and secondary-side capacitor C s As a compensation capacitor for the transmitting and receiving coil modules, it is controlled by the switching transistor S in the parallel switching capacitor on the primary side. C1 By maintaining the on / off state with a certain duty cycle, the equivalent capacitance of the parallel switching capacitor on the primary side is changed, thereby controlling the first switching transistor S in the series switching capacitor on the primary side. p1 and the second switching transistor S p2 By maintaining the on / off state with a certain duty cycle, changing the size of the equivalent capacitance of the primary-side series switched capacitor, a suitable capacitance value is obtained. By changing the capacitance value of the primary-side parallel switched capacitor, the target resonant frequency is obtained. The input impedance angle of the wireless power transmission system is changed by changing the capacitance value of the primary-side series switched capacitor.

[0017] The equivalent circuit of the magnetically coupled wireless power transfer system is determined, and according to Kirchhoff's voltage and current laws, a set of equations relating the input impedance, output efficiency, and compensation parameters are obtained.

[0018] The specific steps of step 2 are as follows:

[0019] The switching transistor S controls the primary-side parallel switching capacitor and the primary-side series switching capacitor. C1and the first switching transistor S p1 and the second switching transistor S p2 Maintain the on / off state with a certain duty cycle, turn on the voltage source and keep the system working normally;

[0020] Obtain the output voltage V of the DC power supply (1) dc The amplitude is U dc After passing through a high-frequency inverter, an AC square wave with an amplitude of U is obtained. in When the duty cycle of the full-bridge inverter is 50%, the voltage U in Sampling is performed, and the input voltage U of the resonant network is... in With input DC voltage U dc The relationship can be expressed as:

[0021]

[0022] The acquired high-frequency voltage passes through an LCC resonant topology compensation circuit and reaches the transmitting coil. Using Kirchhoff's voltage theorem, the following specific equation is obtained:

[0023]

[0024] In the above formula, R eq U is the equivalent resistance of the stage circuit after the rectifier input. s U p X is the induced voltage in the primary and secondary coils generated by the mutual inductance M; L1 X is the reactance of the resonant inductor L1. Lp The inductance L of the primary coil p Reactance, X Ls For the secondary coil inductance L s Reactance, X C1 X is the reactance of the primary-side parallel compensation capacitor C1. Cp The primary-side series compensation capacitor C p Reactance, X Cs For the secondary side series compensation capacitor C s The reactance, I L1 I is the output current of the high-frequency inverter. p I is the primary coil current. s This refers to the secondary coil current, and its specific meaning is as follows:

[0025]

[0026] In the formula, ω is the angular frequency 2πf s f s L is the switching frequency of the high-frequency inverter, L1 is the resonant inductor, and L... p L is the inductance of the primary coil. s C1 is the secondary coil inductance, and C2 is the primary side parallel compensation capacitor.p For primary-side series compensation capacitor, C s The capacitor is a series compensation capacitor on the secondary side, M is the mutual inductance generated by the primary and secondary coils, and R is... L This is the output load resistor.

[0027] 4. The method for optimizing the anti-migration efficiency of wireless power transfer with combined SCC compensation according to claim 1, characterized in that, the specific method for calculating the loop current and input impedance through the primary-side compensation parameters and reflection impedance in step 3 is as follows:

[0028] Select the secondary compensation capacitor C under the resonance of the secondary coil inductance. s To make it resonate with the secondary coil L at the target resonant frequency s Compensation is performed, and the reflection impedance Z ref for:

[0029]

[0030] In the formula, ω is the angular frequency 2πf s f s R is the switching frequency of the high-frequency inverter, M is the mutual inductance generated by the primary and secondary windings, and R is the switching frequency of the high-frequency inverter. eq This is the equivalent resistance of the circuit following the rectifier input.

[0031] Parameters a and b represent the degree of resonance of the primary-side parallel switched capacitor and the primary-side series switched capacitor relative to the resonance compensation, respectively. The specific equations are as follows:

[0032]

[0033] In the formula, ω is the angular frequency 2πf s f s L is the switching frequency of the high-frequency inverter, L1 is the resonant inductor, and L... p C is the inductance of the primary coil. p X is a primary-side series compensation capacitor. C1 The reactance of the primary-side parallel compensation capacitor C1,

[0034] The inductor current I is obtained from the compensation parameters and Kirchhoff's voltage theorem. L1 Primary coil current I p Secondary coil current I s equation:

[0035]

[0036] In the formula, U in ω is the output voltage of the high-frequency inverter, and ω is the angular frequency 2πf. s f sX is the switching frequency of the high-frequency inverter, a is the compensation parameter for the resonance degree of the primary-side parallel switched capacitor, b is the compensation parameter for the resonance degree of the primary-side series switched capacitor, and X is the switching frequency of the high-frequency inverter. L1 Z is the reactance of the resonant inductor L1. ref R is the reflection impedance, M is the mutual inductance generated by the primary and secondary coils, and R is the reflection impedance. eq This is the equivalent resistance of the circuit following the rectifier input.

[0037] For inductor current I L1 Sampling is performed, and the real part Re(I) L1 The imaginary part Im(I) represents the component in phase with the input voltage and the active component. L1 ) represents the components orthogonal to the input voltage and the reactive component, Im(I L1 When )>0, the current lags behind the voltage, and the circuit exhibits inductive load; Im(I L1 When ) < 0, the current leads the voltage, and the circuit exhibits capacitive load; Im(I L1 When )=0, the circuit is a purely resistive load, at which point a=1, b=1, Im(I L1 When the system is in resonance state, the circuit is a purely resistive load, the reactive power consumption is minimal, and the efficiency is highest. Therefore, the reactive power consumption of the system can be controlled by adjusting the parallel compensation capacitor and the series compensation capacitor, so that the system efficiency can reach the best.

[0038] The input impedance Z is defined by the compensation parameter. in for:

[0039]

[0040] In the formula, ω is the angular frequency 2πf s f s Z is the switching frequency of the high-frequency inverter, a is the compensation parameter for the resonance degree of the primary-side parallel switched capacitor, b is the compensation parameter for the resonance degree of the primary-side series switched capacitor, L1 is the resonant inductance, and Z is the resonant inductance. ref This is the reflection impedance.

[0041] 5. The method for optimizing the anti-migration efficiency of wireless power transfer with combined SCC compensation according to claim 1, characterized in that, the specific method for constructing the input impedance angle under the control of compensation parameters in step 4 is as follows:

[0042] The input impedance angle θ, obtained by defining parameters a and b, is expressed as:

[0043]

[0044] in:

[0045]

[0046] In the formula, ω is the angular frequency 2πf s f s Here, is the switching frequency of the high-frequency inverter; 'a' is the compensation parameter for the resonance level of the primary-side parallel switched capacitor; 'b' is the compensation parameter for the resonance level of the primary-side series switched capacitor; 'L1' is the resonant inductance; 'M' is the mutual inductance generated by the primary and secondary windings; and 'R' is the resonant inductance. eq This is the equivalent resistance of the circuit following the rectifier input.

[0047] The input impedance angle θ reflects the phase difference between the inverter voltage and current. When θ > 0, the current lags the voltage, the circuit is an inductive load, and the bridge arm current is still positive when the switch turn-off command arrives. This is a hard turn-off, or zero-voltage turn-on (ZVS). When θ < 0, the current leads the voltage, the circuit is a capacitive load, and the bridge arm current has reversed before the turn-off command arrives. This is a zero-voltage turn-off (ZVS), or hard turn-on. When θ = 0, the circuit is a purely resistive load. At this time, a = 1 and b = 1. When the system is in a resonant state, the circuit is a purely resistive load, but it cannot achieve either zero-voltage turn-on or zero-voltage turn-off. Therefore, the parallel compensation capacitor and the series compensation capacitor can be adjusted to control the circuit to be inductive or capacitive in order to achieve ZVS.

[0048] 6. The method for optimizing the anti-migration efficiency of wireless power transfer with combined SCC compensation according to claim 1, characterized in that step 5 is specifically implemented as follows:

[0049] Full-wave SCC equivalent capacitance C full-eq It can be represented as:

[0050]

[0051] In the formula, α is the conduction angle, and C sp For the switching transistor S p1 and S p2 The parallel capacitor after the source is connected in series, C divp The voltage divider capacitor is connected in series after SCC.

[0052] Single-tube SCC equivalent capacitance C single-eq It can be represented as:

[0053]

[0054] In the formula, α is the conduction angle, and C s1 For the switching transistor S c1 and C div1 The parallel capacitor after series connection, C div1 For the switching transistor S c1 The voltage divider capacitors connected in series later.

[0055] The system input power P obtained from the defined parameters a and b in Represented as:

[0056]

[0057] in:

[0058]

[0059] In the formula, U in I is the output voltage of the high-frequency inverter. L1 ω is the output current of the high-frequency inverter, and ω is the angular frequency 2πf. s f s Here, is the switching frequency of the high-frequency inverter; 'a' is the compensation parameter for the resonance level of the primary-side parallel switched capacitor; 'b' is the compensation parameter for the resonance level of the primary-side series switched capacitor; 'L1' is the resonant inductance; 'M' is the mutual inductance generated by the primary and secondary windings; and 'R' is the resonant inductance. eq This is the equivalent resistance of the circuit following the rectifier input.

[0060] The system output power P obtained from defining parameters a and b o Represented as:

[0061]

[0062] in:

[0063]

[0064] In the formula, U in I is the output voltage of the high-frequency inverter. s ω is the secondary coil current, and ω is the angular frequency 2πf. s f s Here, is the switching frequency of the high-frequency inverter; 'a' is the compensation parameter for the resonance level of the primary-side parallel switched capacitor; 'b' is the compensation parameter for the resonance level of the primary-side series switched capacitor; 'L1' is the resonant inductance; 'M' is the mutual inductance generated by the primary and secondary windings; and 'R' is the resonant inductance. eq This is the equivalent resistance of the circuit following the rectifier input.

[0065] The system transmission efficiency η, obtained by defining parameters a and b, is expressed as:

[0066]

[0067] in:

[0068]

[0069] In the formula, ω is the angular frequency 2πf s f s P is the switching frequency of the high-frequency inverter. in P is the system input power. oThe system output power is given by 'a', which is the compensation parameter for the resonance degree of the primary-side parallel switched capacitor; 'b' is the compensation parameter for the resonance degree of the primary-side series switched capacitor; 'L1' is the resonant inductance; 'M' is the mutual inductance generated by the primary and secondary coils; and 'R' is the system output power. eq This is the equivalent resistance of the circuit following the rectifier input.

[0070] When coil misalignment causes changes in mutual inductance or drift in compensation parameters, adjusting a=1 can achieve a zero input impedance angle state to optimize system efficiency. Adjusting b controls the input impedance angle to make the circuit inductive, thus achieving ZVS for the inverter switching transistor. This involves controlling the parallel switching capacitor on the primary side to resonate with the compensation inductor. Zero-crossing detection of the primary coil current I... p A trigger pulse is generated to control the primary-side series-connected switched capacitor, thereby adjusting the input impedance angle L. s Completely by C p Compensation, current I in the transmitting coil p A load-independent output voltage is achieved on a resistive load.

[0071] During system operation, the coupling state changes continuously with the deflection of the coil. In order to transmit electrical energy in a certain resonant state, fixed frequency control and phase difference control are adopted to stabilize the phase difference between the resonant frequency and the inverter output voltage and current. The inverter output voltage and current are collected to calculate the phase difference and resonant frequency, which are compared with the target phase difference and resonant frequency. Through PID control, excitation pulses with corresponding trigger angles are given to the primary-side parallel switching capacitor and the primary-side series switching capacitor to control the resonant state of the system.

[0072] The beneficial effects of this invention are:

[0073] This invention provides a method for optimizing the anti-offset efficiency of wireless power transfer with combined SCC compensation. It proposes a topology and parameter configuration method for combined SCC compensation wireless power transfer, employing an LCC-S type topology with a compensation model incorporating a primary-side parallel single-tube and series full-wave combined switch-controlled capacitor (SCC). The single-tube SCC addresses the stress problem of surged circulating current in the switching transistor when adjusting the parallel compensation capacitor without zero-crossing detection in a dual-tube SCC. The influence of compensation parameters on system efficiency under varying input impedance, coil offset, and coupling coefficient is analyzed. The parallel SCC and inductor resonance are controlled by detecting the resonant frequency; the input impedance angle is adjusted by detecting the phase difference between the inverter output voltage and current to achieve zero-voltage switching of the inverter and improve system output efficiency. This method demonstrates that the SCC-compensated mechanism exhibits better anti-offset performance than ordinary compensation and is applicable to wireless power transfer systems such as wireless charging for electric vehicles and wireless power supply for home appliances, showing promising engineering application prospects. This invention improves efficiency and stability in the face of resonant mismatch caused by coil offset, requiring only primary-side parameter acquisition and control, eliminating the need for secondary-side communication transmission control. The parallel compensation uses a single-tube switched capacitor, resolving the surge in circulating current that occurs when using dual tubes without zero-crossing detection, thus saving system costs and reducing the difficulty of parameter tuning and control. The specific advantages of this invention are as follows:

[0074] 1) Significantly improves the system's efficiency and stability under dynamic offset conditions, with excellent anti-offset performance.

[0075] This invention innovatively introduces a combination of parallel single-tube and series full-wave controllable switched capacitors (SCCs) into the primary-side LCC compensation network, achieving independent, continuous, and precise adjustment of the resonant frequency and input impedance angle. When coil misalignment causes the mutual inductance M to drop from 20μH to 6μH (equivalent horizontal misalignment of 90mm), the system efficiency using this method is consistently higher than that of a system without SCC compensation, and it effectively controls the inverter output voltage-current phase difference to an optimal near-zero degree. This solves the problem of efficiency drop and detuning that easily occurs in traditional fixed-parameter compensation topologies when coupling conditions change, greatly enhancing the robustness of wireless power transfer systems in practical applications such as dynamic charging of electric vehicles.

[0076] 2) The innovative use of a single-tube SCC for parallel compensation fundamentally solves the problems of circulating current surge and switching stress.

[0077] To address the problem of excessive switching transistor stress caused by the lack of zero-crossing detection in existing dual-transistor SCCs when adjusting parallel capacitors, this invention creatively employs a single-transistor SCC structure for parallel branch compensation. Simulation results clearly show that under the same offset condition (M=6μH), the dual-transistor SCC scheme exhibits a circulating current surge exceeding 200A, while the single-transistor SCC scheme used in this invention successfully limits the circulating current to a safe range of 30A. This design not only significantly reduces the electrical stress and failure risk of the switching transistor, improving system reliability, but also simplifies the drive circuit by eliminating one switching transistor, reducing system cost and control complexity.

[0078] 3) Precisely achieves zero-voltage switching (ZVS) in the inverter, effectively reducing switching losses and electromagnetic interference.

[0079] This invention dynamically adjusts the input impedance angle θ of the system by controlling the series full-wave SCC, keeping it in an inductive state slightly greater than zero (θ > 0). This control strategy ensures that the voltage across the inverter switch is reduced to zero (V_DS=0) before it is turned on, achieving true zero-voltage turn-on (ZVS). Achieving ZVS significantly reduces inverter switching losses, improves energy conversion efficiency, and effectively suppresses voltage and current spikes and electromagnetic interference (EMI), thus improving the system's electromagnetic compatibility.

[0080] 4) Employing a primary-side control strategy eliminates the need for secondary-side communication, resulting in a simpler system structure, lower cost, and easier deployment.

[0081] The control strategy of this invention requires feedback signals (such as resonant frequency, input voltage and current phase) all taken from the primary side, eliminating the need to establish a complex wireless communication link with the secondary receiver. This "primary-side-only control" scheme avoids the cost, power consumption, and reliability issues associated with additional communication modules, simplifies the system architecture, and makes the method more practical and scalable in application scenarios that require powering multiple receivers or where the receiver positions are not fixed (such as AGVs and robots).

[0082] 5) It also features load-independent constant voltage output characteristics, broadening its application scenarios.

[0083] Due to the adoption of a specific LCC-S compensation topology, when the primary-side series compensation capacitor Cp and the secondary-side coil inductance Ls are in perfect resonance, the transmitting coil current Ip is independent of the load size. Combined with the control method of this invention, a stable output voltage that is approximately independent of the load can be achieved on resistive loads. This characteristic makes the system highly suitable for applications requiring stable output voltage, such as constant-voltage charging of batteries, thus enhancing its application value.

[0084] In summary, this invention, through topological innovation of combined SCC**, stress optimization of single-tube parallel compensation, and intelligent control strategy based on primary side parameters, collaboratively solves the core technical bottlenecks of wireless power transmission systems in terms of anti-offset, high efficiency, and high reliability, providing advanced and practical technical solutions for the wide application of wireless charging for electric vehicles, smart homes, medical devices, and other fields. Attached Figure Description

[0085] Figure 1 This is the topology of the magnetically coupled wireless power transmission system in the method of the present invention.

[0086] Figure 2 In the method of this invention, Im(I) L1 A three-dimensional mapping diagram of the changes in compensation coefficients a and b.

[0087] Figure 3 This is a three-dimensional mapping of θ with respect to the compensation coefficients a and b in the method of this invention.

[0088] Figure 4 The diagram shows the current waveforms of the parallel-compensated switched capacitor dual-tube and single-tube switching transistors in the method of this invention.

[0089] Figure 5 This refers to the variation of η with respect to b and k when the compensation parameter a=1 in the method of this invention.

[0090] Figure 6 This refers to the variation of η with respect to a and k when the compensation parameter b=1 in the method of this invention.

[0091] Figure 7 The graphs show the system efficiency versus mutual inductance values ​​with and without SCC compensation in the method of this invention.

[0092] Figure 8 The graphs show the changes in the phase difference between the inverter output voltage and current with and without SCC compensation in the method of this invention.

[0093] Figure 9 The diagram shows the inverter output voltage and current waveforms and the rectified output voltage waveforms in the initial state of the method of the present invention.

[0094] Figure 10 The diagram shows the inverter output voltage and current waveforms and the rectified output voltage waveforms under the condition of 6μH mutual inductance and no SCC compensation in the method of the present invention.

[0095] Figure 11 The diagram shows the inverter output voltage and current waveforms and the rectified output voltage waveforms with SCC compensation and a mutual inductance of 6μH in the method of this invention.

[0096] Figure 12The primary coil current I in the method of this invention p S of the primary-side series switched capacitor (11) p1 S p2 Drive signal and V Csp Waveform diagram.

[0097] Figure 13 The diagram shows the zero-voltage switching (ZVS) waveforms of inverter switching transistors S1, S2, S3, and S4 in the method of this invention. Detailed Implementation

[0098] To make the objectives, technical solutions, and advantages of the embodiments of the present invention clearer, the technical solutions of the embodiments of the present invention will be clearly and completely described below with reference to the accompanying drawings. Obviously, the described embodiments are only some embodiments of the present invention, not all embodiments. Based on the embodiments of the present invention, all other embodiments obtained by those skilled in the art without creative effort are within the scope of protection of the present invention.

[0099] A method for optimizing the anti-migration efficiency of wireless power transfer using combined SCC compensation includes the following steps:

[0100] Step 1: Construct the circuit topology diagram of the magnetically coupled wireless power transfer system;

[0101] refer to Figure 1 The circuit topology of the magnetically coupled wireless power transmission system shown includes: a DC power supply 1, a primary-side high-frequency inverter module 2, a transmitting coil module 3, a receiving coil module 4, a secondary-side rectifier module 5, a load 6, an inverter PWM drive module 7, a switched capacitor PWM drive module 8, a controller 9, a primary-side parallel switched capacitor 10, and a primary-side series switched capacitor 11.

[0102] In this configuration, DC power supply 1 is connected to the input terminal of primary-side high-frequency inverter module 2, the output terminal of high-frequency inverter 2 is connected to the input terminal of transmitting coil module 3, the output terminal of transmitting coil module 3 is positioned opposite to the input terminal of receiving coil module 4, the output terminal of receiving coil module 4 is connected to the input terminal of secondary-side rectifier module 5, the output terminal of secondary-side rectifier module 5 is connected to system load 6, controller 9 outputs control signals to inverter PWM drive module 7 and switched capacitor PWM drive module 8, inverter PWM drive module 7 outputs PWM to primary-side high-frequency inverter module 2, switched capacitor PWM drive module 8 outputs PWM to primary-side parallel switched capacitor 10 and primary-side series switched capacitor 11, primary-side parallel switched capacitor 10 is connected in parallel in the primary-side LCC topology, and primary-side series switched capacitor 11 is connected in series in the LCC topology as a compensation capacitor;

[0103] The DC power supply 1 includes a DC power supply V.dc Its output has one positive terminal and one negative terminal;

[0104] The primary-side high-frequency inverter module 2 includes four MOSFET switches: S1, S2, S3, and S4. The drains of MOSFET switches S1 and S3 are connected together to form the first input terminal, the sources of MOSFET switches S2 and S4 are connected together to form the second input terminal, the source of S1 and the drain of S2 are connected together to form the first output terminal, and the source of MOSFET switch S3 and the drain of MOSFET switch S4 are connected together to form the second output terminal.

[0105] The transmitting coil module 3 includes a coil inductor L p and coil resistance R p L p With R p The terminals are connected in series, and the upper and lower terminals are used as input ports.

[0106] The receiving coil module 4 includes a coil inductor L s and coil resistance R s L s With R s Connected in series, the upper and lower terminals serve as output ports;

[0107] The secondary rectifier module 5 includes four diodes (D1, D2, D3, and D4) and an output filter capacitor C. L The anode of diode D1 and the cathode of diode D2 are connected together to form the first input terminal; the anode of diode D3 and the cathode of diode D4 are connected together to form the second input terminal; the cathodes of diode D1 and D3 are connected together to form the first output terminal; and the anodes of diode D2 and D4 are connected together to form the second output terminal.

[0108] The load 6 includes a load resistor R. L ;

[0109] The inverter PWM drive module 7 includes a drive circuit, whose four output ports are respectively connected to the gates of four MOSFET switching transistors S1, S2, S3, and S4.

[0110] The switched capacitor PWM drive module 8 includes two drive circuits, one of which has an output port connected to a MOSFET switch S. C1 The gate of the other drive circuit is connected to the two output ports of the MOSFET switch S. p1 and S p2 The gate;

[0111] The controller 9 includes resonant frequency, voltage and current sampling, phase shift control, PWM modulation, SCC-C1-PWM, and SCC-C p -PWM, where SCC-C1-PWM is controlled after the resonant frequency is sampled, and SCC-C is controlled after the voltage and current are sampled. p -PWM, PWM modulation after phase-shift control, SCC-C1-PWM and SCC-C p - The PWM is connected to the switch-capacitor PWM module 8 drive circuit;

[0112] The primary-side parallel switched capacitor 10 includes a MOSFET switch S C1 Capacitor C s1 Voltage divider capacitor C div1 S C1 With C div1 After being connected in series with C s1 in parallel;

[0113] The primary-side series switched capacitor 11 includes a MOSFET switch S p1 and S p1 Capacitor C sp Voltage divider capacitor C divp S p1 The source pole and S p1 After the source is connected in series with C sp Parallel connection, then the whole system is connected to C. divp Series connection.

[0114] The primary-side DC input voltage V of the system dc The voltage is 15V, the system switching frequency is 85kHz, and the load R L The Ω is 10Ω, and the transmitting coil L p Receiver coil L s Wound from Litz wire, all with an inductance of 77μH, and L p With L s The mutual inductance M between them is 20μH, the inductance of the compensating inductor L1 is 22μH, and the initial value of the compensating capacitor C1 is 159nF. p The initial value is set to 64nF, C s The value is 45 nF;

[0115] Step 2: Obtain the output voltage V of the DC power supply. dc The signal is then output to the primary-side high-frequency inverter module, which outputs an 85kHz AC square wave to the primary-side coil. The specific steps include:

[0116] The switching transistor S controls the primary-side parallel switching capacitor 10 and the primary-side series switching capacitor 11. C1 and the first switching transistor S p1 and the second switching transistor Sp2 The system is kept in the on / off state with a certain duty cycle, so that the conduction angle of the primary-side parallel switching capacitor 10 is 37.7° and the conduction angle of the primary-side series switching capacitor 11 is 135.1°. This is the initial state of the system. The voltage source is turned on and the system is judged to be working normally.

[0117] Obtain the output voltage V of DC power supply 1 dc The amplitude is U dc After passing through a high-frequency inverter, an AC square wave with an amplitude of U is obtained. in When the duty cycle of the full-bridge inverter is 50%, the voltage U in Sampling is performed, and the input voltage U of the resonant network is... in With input DC voltage U dc The relationship can be expressed as:

[0118]

[0119] U can be obtained from the above formula. in =13.5V∠0°;

[0120] The acquired high-frequency voltage is fed to the transmitting coil through an LCC resonant topology compensation circuit. The following specific equation is obtained by applying Kirchhoff's voltage theorem.

[0121]

[0122] In the above formula, R eq U is the equivalent resistance of the stage circuit after the rectifier input. s U p X is the induced voltage in the primary and secondary coils generated by the mutual inductance M; L1 X is the reactance of the resonant inductor L1. Lp The inductance L of the primary coil p Reactance, X Ls For the secondary coil inductance L s Reactance, X C1 X is the reactance of the primary-side parallel compensation capacitor C1. Cp The primary-side series compensation capacitor C p Reactance, X Cs For the secondary side series compensation capacitor C s The reactance, I L1 I is the output current of the high-frequency inverter. p I is the primary coil current. s This refers to the secondary coil current, and its specific meaning is as follows:

[0123]

[0124] In the formula, ω is the angular frequency 2πf s f sL is the switching frequency of the high-frequency inverter, L1 is the resonant inductor, and L... p L is the inductance of the primary coil. s C1 is the secondary coil inductance, and C2 is the primary side parallel compensation capacitor. p For primary-side series compensation capacitor, C s The capacitor is a series compensation capacitor on the secondary side, M is the mutual inductance generated by the primary and secondary coils, and R is... L This is the output load resistor.

[0125] Step 3: Calculate the loop current and input impedance using the primary-side compensation parameters and reflection impedance. This includes the following steps:

[0126] Select the secondary compensation capacitor C under the resonance of the secondary coil inductance. s To make it resonate with the secondary coil L at the target resonant frequency s After compensation, the reflection impedance is:

[0127]

[0128] In the formula, ω is the angular frequency 2πf s f s R is the switching frequency of the high-frequency inverter, M is the mutual inductance generated by the primary and secondary windings, and R is the switching frequency of the high-frequency inverter. eq This is the equivalent resistance of the circuit following the rectifier input.

[0129] Parameters a and b are defined to represent the degree of resonance of the primary-side parallel switched capacitor and the primary-side series switched capacitor relative to the resonance compensation, respectively. The specific equation is as follows:

[0130]

[0131] In the formula, ω is the angular frequency 2πf s f s L is the switching frequency of the high-frequency inverter, L1 is the resonant inductor, and L... p C is the inductance of the primary coil. p X is a primary-side series compensation capacitor. C1 The reactance of the primary-side parallel compensation capacitor C1,

[0132] The inductor current I is obtained from the compensation parameters and Kirchhoff's voltage theorem. L1 Primary coil current I p Secondary coil current I s equation:

[0133]

[0134] In the formula, U in ω is the output voltage of the high-frequency inverter, and ω is the angular frequency 2πf. s f sX is the switching frequency of the high-frequency inverter, a is the compensation parameter for the resonance degree of the primary-side parallel switched capacitor, b is the compensation parameter for the resonance degree of the primary-side series switched capacitor, and X is the switching frequency of the high-frequency inverter. L1 Z is the reactance of the resonant inductor L1. ref R is the reflection impedance, M is the mutual inductance generated by the primary and secondary coils, and R is the reflection impedance. eq This is the equivalent resistance of the circuit following the rectifier input.

[0135] When the compensation parameter is in the initial state, i.e., a=b=1, I is obtained. L1 =1.38A, I p =1.15A∠-90°, I s =-1.51A;

[0136] For inductor current I L1 Sampling is performed, and the real part Re(I) L1 The imaginary part Im(I) represents the component in phase with the input voltage and the active component. L1 ) represents components orthogonal to the input voltage and reactive components, such as Figure 2 As shown, Im(I) L1 A three-dimensional surface with respect to the variations of a and b, Im(I L1 When )>0, the current lags behind the voltage, and the circuit exhibits inductive load; Im(I L1 When ) < 0, the current leads the voltage, and the circuit exhibits capacitive load; Im(I L1 When )=0, the circuit is a purely resistive load. Figure 2 Plot a=1, b=1 and Im(I) L1 The intersection of the curves )=0 indicates that the system is in a resonant state. At this point, the circuit is a purely resistive load, the system has the minimum reactive power consumption and the highest efficiency. Therefore, the reactive power consumption of the system can be controlled by adjusting the parallel compensation capacitor and the series compensation capacitor, so that the system efficiency can reach the best.

[0137] The input impedance Z is defined by the compensation parameter. in for:

[0138]

[0139] In the formula, ω is the angular frequency 2πf s f s Z is the switching frequency of the high-frequency inverter, a is the compensation parameter for the resonance degree of the primary-side parallel switched capacitor, b is the compensation parameter for the resonance degree of the primary-side series switched capacitor, L1 is the resonant inductance, and Z is the resonant inductance. ref For reflection impedance,

[0140] Step 4: Given the system resonant frequency and the phase difference between the inverter output voltage and current, construct the input impedance angle under the control of compensation parameters, specifically including the following steps:

[0141] The input impedance angle θ, obtained by defining parameters a and b, is expressed as:

[0142]

[0143] in:

[0144]

[0145] In the formula, ω is the angular frequency 2πf s f s Here, is the switching frequency of the high-frequency inverter; 'a' is the compensation parameter for the resonance level of the primary-side parallel switched capacitor; 'b' is the compensation parameter for the resonance level of the primary-side series switched capacitor; 'L1' is the resonant inductance; 'M' is the mutual inductance generated by the primary and secondary windings; and 'R' is the resonant inductance. eq This is the equivalent resistance of the circuit following the rectifier input.

[0146] like Figure 3 The three-dimensional surface showing the input impedance angle θ varying with respect to a and b reflects the phase difference relationship between the inverter voltage and current. When θ > 0, the current lags the voltage, the circuit is an inductive load, and the bridge arm current is still positive when the switch turn-off command arrives; this is hard turn-off, zero-voltage turn-on (ZVS). When θ < 0, the current leads the voltage, the circuit is a capacitive load, and the bridge arm current has reversed before the turn-off command arrives; this is zero-voltage turn-off (ZVS), hard turn-on. When θ = 0, the circuit is a purely resistive load. Figure 3 The intersection of the curves a=1, b=1 and θ=0 is plotted. When the system is in resonance, the circuit is a purely resistive load, but it cannot achieve zero voltage turn-on or zero voltage turn-off. Therefore, the parallel compensation capacitor and the series compensation capacitor can be adjusted to control the circuit to be inductive or capacitive in order to achieve ZVS.

[0147] Step 5: Control the parallel single-tube controllable switching capacitor C1 on the primary side to make it resonate with the compensation inductor, and control the series full-wave controllable switching capacitor C on the primary side. p By controlling the input impedance angle to be close to zero and greater than zero, the circuit becomes inductive, achieving zero-voltage switching (ZVS) of the inverter switching transistors and improving the overall system transmission efficiency. This specifically includes the following steps:

[0148] Full-wave SCC equivalent capacitance C full-eq It can be represented as:

[0149]

[0150] In the formula, α is the conduction angle, and C sp For the switching transistor S p1 and S p2 The parallel capacitor after the source is connected in series, C divpThe voltage divider capacitor is connected in series after SCC.

[0151] Based on the conduction angle of 135.1° and the equivalent capacitance of 64nF of the primary-side series-connected switching capacitor 11, C is calculated. sp =182nF, C divp =68nF.

[0152] Single-tube SCC equivalent capacitance C single-eq It can be represented as:

[0153]

[0154] In the formula, α is the conduction angle, and C s1 For the switching transistor S c1 and C div1 The parallel capacitor after series connection, C div1 For the switching transistor S c1 The voltage divider capacitors connected in series later.

[0155] Based on the conduction angle of the primary-side parallel switching capacitor 10 (37.7°) and its equivalent capacitance of 159nF, C is calculated. s1 =156nF, C div1 =4.4nF.

[0156] By using a parallel compensation capacitor on the primary side, the cost of components and the complexity of control can be reduced without using zero-crossing detection, such as... Figure 4 The figure shows the current I of switch transistor 1 when using a dual-transistor switched capacitor with a mutual inductance of 6μH. DC1 , Switching transistor 2 current I DC2 The switching transistor S of a single-tube switched capacitor C1 Current I C1 Waveform. From Figure 4 The waveforms show that a surge in circulating current occurred when using a dual-tube switched capacitor for regulation, with the maximum surge current exceeding 200A. However, when using a single-tube switched capacitor for regulation, the circulating current was smaller, with a maximum of no more than 30A, which is within the stress range of the switching transistor. Therefore, using a single-tube switched capacitor instead of a dual-tube switched capacitor to regulate the parallel compensation capacitor without zero-crossing detection can reduce the switching stress requirements and lower device costs and control complexity.

[0157] The system input power P obtained from the defined parameters a and b in Represented as:

[0158]

[0159] in:

[0160]

[0161] In the formula, U in I is the output voltage of the high-frequency inverter. L1ω is the output current of the high-frequency inverter, and ω is the angular frequency 2πf. s f s Here, is the switching frequency of the high-frequency inverter; 'a' is the compensation parameter for the resonance level of the primary-side parallel switched capacitor; 'b' is the compensation parameter for the resonance level of the primary-side series switched capacitor; 'L1' is the resonant inductance; 'M' is the mutual inductance generated by the primary and secondary windings; and 'R' is the resonant inductance. eq This is the equivalent resistance of the circuit following the rectifier input.

[0162] When the compensation parameter is in the initial state, i.e., a=b=1, P is obtained. in =17.75W.

[0163] The system output power P obtained from defining parameters a and b o Represented as:

[0164]

[0165] in:

[0166]

[0167] In the formula, U in I is the output voltage of the high-frequency inverter. s ω is the secondary coil current, and ω is the angular frequency 2πf. s f s Here, is the switching frequency of the high-frequency inverter; 'a' is the compensation parameter for the resonance level of the primary-side parallel switched capacitor; 'b' is the compensation parameter for the resonance level of the primary-side series switched capacitor; 'L1' is the resonant inductance; 'M' is the mutual inductance generated by the primary and secondary windings; and 'R' is the resonant inductance. eq This is the equivalent resistance of the circuit following the rectifier input.

[0168] When the compensation parameter is in the initial state, i.e., a=b=1, P is obtained. o =16.16W.

[0169] The system transmission efficiency η, obtained by defining parameters a and b, is expressed as:

[0170]

[0171] in:

[0172]

[0173] In the formula, ω is the angular frequency 2πf s f s P is the switching frequency of the high-frequency inverter. in P is the system input power. oThe system output power is given by 'a', which is the compensation parameter for the resonance degree of the primary-side parallel switched capacitor; 'b' is the compensation parameter for the resonance degree of the primary-side series switched capacitor; 'L1' is the resonant inductance; 'M' is the mutual inductance generated by the primary and secondary coils; and 'R' is the system output power. eq This is the equivalent resistance of the circuit following the rectifier input.

[0174] When the compensation parameter is in the initial state, i.e., a=b=1, η=91% is obtained.

[0175] Figure 5 The system efficiency η is a three-dimensional surface with respect to the changes in compensation parameter b and coupling coefficient k when a=1. The compensation parameter b changes from 0 to 2, and the coupling coefficient k changes from 0.1 to 0.5. The system efficiency η is mainly affected by k and less affected by b. Figure 6 The system efficiency η is a three-dimensional surface representing the system efficiency η when b=1, with respect to the variations in the compensation parameter a and the coupling coefficient k. The compensation parameter a varies from 0 to 4.5, and the coupling coefficient k varies from 0.1 to 0.5. Within a certain compensation range, such as a varying from 0 to 2, the system efficiency η is optimal when a=1 when k varies. Therefore, from... Figure 5 and Figure 6 It can be seen that, under the influence of coil offset causing changes in mutual inductance or drift in compensation parameters, the zero input impedance angle state can be achieved by adjusting a=1 to optimize system efficiency. Furthermore, adjusting b controls the input impedance angle to make the circuit inductive, thus achieving ZVS in the inverter. This involves controlling the primary-side parallel switching capacitor 10 to resonate with the compensation inductor; and detecting the zero-crossing primary coil current I. p A trigger pulse is generated to control the primary-side series-connected switched capacitor 11, thereby adjusting the input impedance angle. s Completely by C p Compensation, current I in the transmitting coil p A load-independent output voltage is achieved on a resistive load.

[0176] In the initial state, the offset of the magnetically coupled coils leads to a decrease in mutual inductance. A Simulink simulation model is built, considering the resistance loss of the primary and secondary coils, i.e., R... p =R s =0.369Ω, the simulated mutual inductance value gradually decreases from 20μH, controlling the primary-side parallel switching capacitor 10 and the primary-side series switching capacitor 11, the operation is as follows. Figure 7 The graph shows the efficiency curves with and without SCC compensation as a function of mutual inductance. The continuous decrease of the coil represents the continuous increase of the offset distance. The mutual inductance decreases from 20μH to 6μH, which means the horizontal offset distance increases from 0mm to 90mm. As the mutual inductance decreases and the offset distance increases, the system efficiency decreases. The overall efficiency of the system with SCC compensation is higher than that without SCC compensation, which shows the effectiveness of the proposed SCC compensation method in improving system efficiency.

[0177] like Figure 8 The figure shows the output voltage and current phase difference curves of inverters with and without SCC compensation as a function of mutual inductance. The continuous decrease of the coil represents the continuous increase of the offset distance. The mutual inductance decreases from 20μH to 6μH, which means the horizontal offset distance increases from 0mm to 90mm. As the mutual inductance decreases, i.e. the offset distance increases, the phase difference increases without SCC compensation, while the phase difference is about 0 with SCC compensation. This shows the effectiveness of the proposed SCC compensation control method for input impedance angle control.

[0178] like Figure 9 The figure shows the inverter output voltage and current waveforms and the rectified output voltage waveform in the initial state. At this time, the mutual inductance is 20μH, that is, the coupling coefficient is 0.260, the rectified output voltage is 12.7V, the output power is 16.13W, and the system efficiency is 81.69%. It can be seen from the inverter output voltage and current waveforms that their phases are basically aligned.

[0179] When the coil offset results in a mutual inductance of 6μH, i.e., a coupling coefficient of 0.078, such as... Figure 10 The figure shows the inverter output voltage and current waveforms and the rectified output voltage waveform without SCC compensation. At this time, the mutual inductance is 6μH, that is, the coupling coefficient is 0.078, the rectified output voltage is 3.82V, the output power is 1.45W, and the system efficiency is 31.4%. It can be seen from the inverter output voltage and current waveforms that their phases are misaligned, indicating that the coil offset will cause a phase difference in the inverter output voltage and current.

[0180] When the coil offset results in a mutual inductance of 6μH, i.e., a coupling coefficient of 0.078, such as... Figure 11 The figure shows the inverter output voltage and current waveforms and the rectified output voltage waveform under SCC compensation. At this time, the mutual inductance is 6μH, i.e., the coupling coefficient is 0.078, the rectified output voltage is 3.97V, the output power is 1.57W, and the system efficiency is 46.6%. From the fundamental waveform of the inverter output voltage and current, it can be seen that their phases are basically aligned, indicating that the proposed SCC compensation method can control the system resonance state to align the phases of the inverter output voltage and current, thereby improving the system efficiency.

[0181] like Figure 12 The figure shows the primary coil current I. p The S of the primary-side series switched capacitor 11 p1 S p2 Drive signal and SCC capacitor C sp V Csp The waveform shows that the SCC switch drive signal S... p1 and S p2 The conduction time and the primary coil current I p The zero-crossing point correspondence is used to reduce switching losses, and from VCsp The waveform shows that C sp The charging and discharging trends are consistent with the theoretical model analysis, verifying the normal operation of the SCC's regulation function.

[0182] like Figure 13 The diagram shows the zero-voltage switching (ZVS) waveforms of inverter switches S1, S2, S3, and S4. The waveforms include the drive signals S for switches S1, S2, S3, and S4. G1 ~S G4 Current I S1 ~I S4 and the voltage V at both ends S1 ~V S4 As can be seen from the waveform, in the driving signal S G1 S G4 Before S1 and S4 are turned on, the voltage V across them is... S1 V S4 It has dropped to 0, the turn-on time and current I S1 I S4 The product is zero, achieving zero-voltage turn-on; in the drive signal S G2 S G3 The voltage V across S2 and S3 will be present for a period of time after S2 and S3 are turned off. S2 V S3 The value remains 0, and the turn-off time is related to the current I. S2 I S3 The product is zero, achieving zero-voltage turn-off, indicating that the proposed SCC compensation control method can effectively achieve ZVS for the inverter switching transistors.

[0183] As a further technical solution of the present invention: During the operation of the system, the coupling state changes continuously with the deflection of the coil. In order to transmit electrical energy in a certain resonant state, fixed frequency control and phase difference control are adopted to stabilize the phase difference between the resonant frequency and the inverter output voltage and current. The inverter output voltage and current are collected to calculate the phase difference and resonant frequency, which are compared with the target phase difference and resonant frequency. Through PID control, excitation pulses are given to the two switched capacitors at the corresponding trigger angles to control the resonant state of the system.

[0184] It will be apparent to those skilled in the art that the present invention is not limited to the details of the exemplary embodiments described above, and that the invention can be implemented in other specific forms without departing from the spirit or essential characteristics of the invention. Therefore, the embodiments should be considered in all respects as exemplary and non-limiting, and the scope of the invention is defined by the appended claims rather than the foregoing description. Thus, it is intended that all variations falling within the meaning and scope of equivalents of the claims be included within the present invention.

[0185] Furthermore, it should be understood that although this specification describes embodiments, not every embodiment contains only one independent technical solution. This way of describing the specification is only for clarity. Those skilled in the art should regard the specification as a whole, and the technical solutions in each embodiment have been appropriately combined to form other embodiments that are easy for those skilled in the art to understand.

Claims

1. A method for optimizing the anti-migration efficiency of wireless power transfer using combined SCC compensation, characterized in that, Includes the following steps: Step 1: Construct the topology of the magnetically coupled wireless power transfer system; Step 2, obtain the output voltage V of the DC power supply. dc It is then output to the primary-side high-frequency inverter module, which outputs an 85kHz AC square wave to the primary-side coil. Step 3: Calculate the loop current and input impedance using the primary side compensation parameters and reflection impedance; Step 4: Given the system resonant frequency and the phase difference between the inverter output voltage and current, construct the input impedance angle under the control of compensation parameters; Step 5: By controlling the parallel single-tube controllable switching capacitor C1 on the primary side to make it resonate with the compensation inductor, the series full-wave controllable switching capacitor C on the primary side is controlled. p By controlling the circuit to make the input impedance angle close to zero and greater than zero, the circuit becomes inductive, thereby achieving zero-voltage switching of the inverter switching transistors and improving the overall transmission efficiency of the system.

2. The method for optimizing the anti-migration efficiency of wireless power transfer with combined SCC compensation according to claim 1, characterized in that, The topology of the magnetically coupled wireless power transfer system described in step 1 further includes: DC power supply, primary-side high-frequency inverter module, transmitting coil module, receiving coil module, secondary-side rectifier module, load, inverter PWM drive module, switched capacitor PWM drive module, controller, primary-side parallel switched capacitor and primary-side series switched capacitor; The DC power supply is connected to the input of the primary-side high-frequency inverter module. The output of the high-frequency inverter is connected to the input of the transmitting coil module. The output of the transmitting coil module is positioned opposite to the input of the receiving coil module. The output of the receiving coil module is connected to the input of the secondary-side rectifier module. The output of the secondary-side rectifier module is connected to the system load. The output of the controller is connected to the input of the inverter PWM drive module and the input of the switched-capacitor PWM drive module. The output of the inverter PWM drive module is connected to the input of the primary-side high-frequency inverter module. The output of the switched-capacitor PWM drive module is connected to the input of the primary-side parallel switched-capacitor and the primary-side series switched-capacitor. The controller outputs control signals to the inverter PWM drive module and the switched-capacitor PWM drive module. The inverter PWM drive module outputs PWM signals to the primary-side high-frequency inverter module. The switched-capacitor PWM drive module outputs PWM signals to the primary-side parallel switched-capacitor and the primary-side series switched-capacitor. The primary-side parallel switched-capacitor is connected in parallel in the primary-side LCC topology, and the primary-side series switched-capacitor is connected in series in the LCC topology as a compensation capacitor. The primary side DC power supply is controlled to obtain a suitable output voltage. The high-frequency voltage is obtained by the primary side high-frequency inverter through PWM control. The voltage is then passed through the topology compensation coil structure to the output of the transmitting coil module and then to the input voltage of the receiving coil module. The secondary side rectifier module converts the high-frequency AC power into DC power. The secondary side load is connected to the output terminal of the secondary side rectifier module to transmit power. Primary-side parallel switched capacitor, primary-side series switched capacitor and secondary-side capacitor C s As a compensation capacitor for the transmitting and receiving coil modules, it is controlled by the switching transistor S in the primary-side parallel switching capacitor. C1 By maintaining the on / off state with a certain duty cycle, the equivalent capacitance of the parallel switching capacitor on the primary side is changed, thereby controlling the first switching transistor S in the series switching capacitor on the primary side. p1 and the second switching transistor S p2 By maintaining the on / off state with a certain duty cycle, changing the size of the equivalent capacitance of the primary-side series switched capacitor, a suitable capacitance value is obtained. By changing the capacitance value of the primary-side parallel switched capacitor, the target resonant frequency is obtained. The input impedance angle of the wireless power transmission system is changed by changing the capacitance value of the primary-side series switched capacitor. The equivalent circuit of the magnetically coupled wireless power transfer system is determined, and according to Kirchhoff's voltage and current laws, a set of equations relating the input impedance, output efficiency, and compensation parameters are obtained.

3. The method for optimizing the anti-migration efficiency of wireless power transfer with combined SCC compensation according to claim 1, characterized in that, The specific steps of step 2 are as follows: The switching transistor S controls the primary-side parallel switching capacitor and the primary-side series switching capacitor. C1 and the first switching transistor S p1 and the second switching transistor S p2 Maintain the on / off state with a certain duty cycle, turn on the voltage source and keep the system working normally; Obtain the output voltage V of the DC power supply (1) dc The amplitude is U dc After passing through a high-frequency inverter, an AC square wave with an amplitude of U is obtained. in When the duty cycle of the full-bridge inverter is 50%, the voltage U in Sampling is performed, and the input voltage U of the resonant network is... in With input DC voltage U dc The relationship can be expressed as: ； The acquired high-frequency voltage passes through an LCC resonant topology compensation circuit and reaches the transmitting coil. Using Kirchhoff's voltage theorem, the following specific equation is obtained: ； In the above formula, R eq U is the equivalent resistance of the stage circuit after the rectifier input. s U p X is the induced voltage in the primary and secondary coils generated by the mutual inductance M; L1 X is the reactance of the resonant inductor L1. Lp The inductance L of the primary coil p Reactance, X Ls For the secondary coil inductance L s Reactance, X C1 X is the reactance of the primary-side parallel compensation capacitor C1. Cp The primary-side series compensation capacitor C p Reactance, X Cs For the secondary side series compensation capacitor C s The reactance, I L1 I is the output current of the high-frequency inverter. p I is the primary coil current. s This refers to the secondary coil current, and its specific meaning is as follows: ； In the formula, ω is the angular frequency 2πf s f s L is the switching frequency of the high-frequency inverter, L1 is the resonant inductor, and L... p L is the inductance of the primary coil. s C1 is the secondary coil inductance, and C2 is the primary side parallel compensation capacitor. p For primary-side series compensation capacitor, C s The capacitor is a series compensation capacitor on the secondary side, M is the mutual inductance generated by the primary and secondary coils, and R is... L This is the output load resistor.

4. The method for optimizing the anti-migration efficiency of wireless power transfer with combined SCC compensation according to claim 1, characterized in that, Step 3, which involves calculating the loop current and input impedance using the primary-side compensation parameters and reflection impedance, is specifically implemented as follows: Select the secondary compensation capacitor C under the resonance of the secondary coil inductance. s To make it resonate with the secondary coil L at the target resonant frequency s Compensation is performed, and the reflection impedance Z ref for: ； In the formula, ω is the angular frequency 2πf s f s R is the switching frequency of the high-frequency inverter, M is the mutual inductance generated by the primary and secondary windings, and R is the switching frequency of the high-frequency inverter. eq This is the equivalent resistance of the circuit following the rectifier input. Parameters a and b represent the degree of resonance of the primary-side parallel switched capacitor and the primary-side series switched capacitor relative to the resonance compensation, respectively. The specific equations are as follows: ； In the formula, ω is the angular frequency 2πf s f s L is the switching frequency of the high-frequency inverter, L1 is the resonant inductor, and L... p C is the inductance of the primary coil. p X is a primary-side series compensation capacitor. C1 The reactance of the primary-side parallel compensation capacitor C1, The inductor current I is obtained from the compensation parameters and Kirchhoff's voltage theorem. L1 Primary coil current I p Secondary coil current I s equation: ； In the formula, U in ω is the output voltage of the high-frequency inverter, and ω is the angular frequency 2πf. s f s X is the switching frequency of the high-frequency inverter, a is the compensation parameter for the resonance degree of the primary-side parallel switched capacitor, b is the compensation parameter for the resonance degree of the primary-side series switched capacitor, and X is the switching frequency of the high-frequency inverter. L1 Z is the reactance of the resonant inductor L1. ref R is the reflection impedance, M is the mutual inductance generated by the primary and secondary coils, and R is the reflection impedance. eq This is the equivalent resistance of the circuit following the rectifier input. For inductor current I L1 Sampling is performed, and the real part Re(I) L1 The imaginary part Im(I) represents the component in phase with the input voltage and the active component. L1 ) represents the components orthogonal to the input voltage and the reactive component, Im(I L1 When )>0, the current lags behind the voltage, and the circuit exhibits inductive load; Im(I L1 When ) < 0, the current leads the voltage, and the circuit exhibits capacitive load; Im(I L1 When )=0, the circuit is a purely resistive load, at which point a=1, b=1, Im(I L1 When the system is in resonance state, the circuit is a purely resistive load, the reactive power consumption is minimal, and the efficiency is highest. Therefore, the reactive power consumption of the system can be controlled by adjusting the parallel compensation capacitor and the series compensation capacitor, so that the system efficiency can reach the best. The input impedance Z is defined by the compensation parameter. in for: ； In the formula, ω is the angular frequency 2πf s f s Z is the switching frequency of the high-frequency inverter, a is the compensation parameter for the resonance degree of the primary-side parallel switched capacitor, b is the compensation parameter for the resonance degree of the primary-side series switched capacitor, L1 is the resonant inductance, and Z is the resonant inductance. ref This is the reflection impedance.

5. The method for optimizing the anti-migration efficiency of wireless power transfer with combined SCC compensation according to claim 1, characterized in that, Step 4 describes the construction of the input impedance angle under the control of compensation parameters. The specific steps are as follows: The input impedance angle θ, obtained by defining parameters a and b, is expressed as: ； in: ； In the formula, ω is the angular frequency 2πf s f s Here, is the switching frequency of the high-frequency inverter; 'a' is the compensation parameter for the resonance level of the primary-side parallel switched capacitor; 'b' is the compensation parameter for the resonance level of the primary-side series switched capacitor; 'L1' is the resonant inductance; 'M' is the mutual inductance generated by the primary and secondary coils; and 'R' is the resonant inductance. eq This is the equivalent resistance of the circuit following the rectifier input. The input impedance angle θ reflects the phase difference between the inverter voltage and current. When θ > 0, the current lags the voltage, the circuit is an inductive load, and the bridge arm current is still positive when the switch turn-off command arrives. This is a hard turn-off, or zero-voltage turn-on (ZVS). When θ < 0, the current leads the voltage, the circuit is a capacitive load, and the bridge arm current has reversed before the turn-off command arrives. This is a zero-voltage turn-off (ZVS), or hard turn-on. When θ = 0, the circuit is a purely resistive load. At this time, a = 1 and b = 1. When the system is in a resonant state, the circuit is a purely resistive load, but it cannot achieve either zero-voltage turn-on or zero-voltage turn-off. Therefore, the parallel compensation capacitor and the series compensation capacitor can be adjusted to control the circuit to be inductive or capacitive in order to achieve ZVS.

6. The method for optimizing the anti-migration efficiency of wireless power transfer with combined SCC compensation according to claim 1, characterized in that, Step 5, as described above, is specifically implemented as follows: Full-wave SCC equivalent capacitance C full-eq It can be represented as: ； In the formula, α is the conduction angle, and C sp For the switching transistor S p1 and S p2 The parallel capacitor after the source is connected in series, C divp This is a voltage divider capacitor connected in series after SCC. Single-tube SCC equivalent capacitance C single-eq It can be represented as: ； In the formula, α is the conduction angle, and C s1 For the switching transistor S c1 and C div1 The parallel capacitor after series connection, C div1 For the switching transistor S c1 The voltage divider capacitors connected in series afterward. The system input power P obtained from the defined parameters a and b in Represented as: ； in: ； In the formula, U in I is the output voltage of the high-frequency inverter. L1 ω is the output current of the high-frequency inverter, and ω is the angular frequency 2πf. s f s Here, is the switching frequency of the high-frequency inverter; 'a' is the compensation parameter for the resonance level of the primary-side parallel switched capacitor; 'b' is the compensation parameter for the resonance level of the primary-side series switched capacitor; 'L1' is the resonant inductance; 'M' is the mutual inductance generated by the primary and secondary coils; and 'R' is the resonant inductance. eq This is the equivalent resistance of the circuit following the rectifier input. The system output power P obtained from defining parameters a and b o Represented as: ； in: ； In the formula, U in I is the output voltage of the high-frequency inverter. s ω is the secondary coil current, and ω is the angular frequency 2πf. s f s Here, is the switching frequency of the high-frequency inverter; 'a' is the compensation parameter for the resonance level of the primary-side parallel switched capacitor; 'b' is the compensation parameter for the resonance level of the primary-side series switched capacitor; 'L1' is the resonant inductance; 'M' is the mutual inductance generated by the primary and secondary coils; and 'R' is the resonant inductance. eq This is the equivalent resistance of the circuit following the rectifier input. The system transmission efficiency η, obtained by defining parameters a and b, is expressed as: ； in: ； In the formula, ω is the angular frequency 2πf s f s P is the switching frequency of the high-frequency inverter. in P is the system input power. o The system output power is given by , a is the compensation parameter for the resonance degree of the primary-side parallel switched capacitor, b is the compensation parameter for the resonance degree of the primary-side series switched capacitor, L1 is the resonant inductance, M is the mutual inductance generated by the primary and secondary coils, and R... eq This is the equivalent resistance of the circuit following the rectifier input. When coil misalignment causes changes in mutual inductance or drift in compensation parameters, adjusting a=1 can achieve a zero input impedance angle state to optimize system efficiency. Adjusting b controls the input impedance angle to make the circuit inductive, thus achieving ZVS for the inverter switching transistor. This involves controlling the parallel switching capacitor on the primary side to resonate with the compensation inductor. Zero-crossing detection of the primary coil current I... p A trigger pulse is generated to control the primary-side series-connected switched capacitor, thereby adjusting the input impedance angle L. s Completely by C p Compensation, current I in the transmitting coil p A load-independent output voltage is achieved on a resistive load.

7. The method for optimizing the anti-migration efficiency of wireless power transfer with combined SCC compensation according to claim 1, characterized in that, During system operation, the coupling state changes continuously with the deflection of the coil. In order to transmit electrical energy in a certain resonant state, fixed frequency control and phase difference control are adopted to stabilize the phase difference between the resonant frequency and the inverter output voltage and current. The inverter output voltage and current are collected to calculate the phase difference and resonant frequency, which are compared with the target phase difference and resonant frequency. Through PID control, excitation pulses with corresponding trigger angles are given to the primary-side parallel switching capacitor and the primary-side series switching capacitor to control the resonant state of the system.

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