High-boost coupling inductor DC-DC converter adopting switched capacitor and coupling inductor technology

Abstract

The invention provides a high-boost coupling inductor DC-DC converter adopting a switched capacitor and coupling inductor technology, and relates to the technical field of converters. The converter comprises an input inductor, a coupling inductor, a switch, diodes D1-D6, capacitors C1-C3 and output capacitors Co1 and Co2. The coupling inductance equivalent model comprises a magnetization inductor, a primary leakage inductor and an ideal transformer; the converter is formed by cascading a primary two-stage boost converter and a secondary switched capacitor structure, the two-stage boost converter comprises capacitors C1, Co1, a diode D3 and a primary side of an ideal transformer, and in the switched capacitor structure, capacitors C2 and C3 realize parallel charging and series discharging; a diode in the switched capacitor unit realizes zero-current switching through leakage inductance of a coupling inductor, and leakage inductance energy is fed back to a load. According to the converter, a switched capacitor structure is cascaded on the basis of a two-stage boost converter, so that the gain of the converter is improved.

Description

Technical Field

[0001] This application relates to the field of converter technology, and in particular to a high boost coupled inductor DC-DC converter employing switched capacitors and coupled inductors. Background Technology

[0002] Against the backdrop of rapid global economic development, fossil energy is becoming increasingly depleted. At the same time, the environmental pollution caused by its combustion process is becoming more and more serious, which has promoted the accelerated development and widespread application of new energy sources such as wind power and solar energy. New energy has become a key direction for solving the dual challenges of energy and environment. In recent years, the installed capacity and scale of renewable energy power generation have been continuously expanding. Among them, photovoltaic power generation has received widespread attention due to its zero emission and convenient access. Although traditional DC-DC converters can meet the usage requirements in applications such as photovoltaic array boosting, there are still shortcomings. For example, the voltage stress of the power switch is the same as the output voltage[6], which leads to the need to use power switches with high rated voltage stress. Such switches are often accompanied by large on-resistance, which in turn leads to increased power loss. In addition, photovoltaic power generation is intermittent and fluctuating. Traditional DC-DC converters need to work at the limit duty cycle to achieve high voltage gain, and under such circumstances, their regulation capability is insufficient when performing maximum power point tracking (MPPT).

[0003] To address these issues, existing technologies have proposed various solutions to improve gain. Currently commonly used boost technologies include cascading technology, switched capacitors, coupled inductors, and isolation transformers.

[0004] The paper "A. Torkan and M. Ehsani, “A novel nonisolated Z-source DC–DC converter for photovoltaic applications,” IEEE Trans. Ind. Appl., vol. 54, no. 5, pp. 4574–4583, Sep. 2018.” proposes a nonisolated converter integrating a Z-source network, coupled inductor, and voltage multiplication technology. This converter exhibits higher voltage gain compared to traditional Z-source converters and can be implemented at low switching duty cycles. It demonstrates lower normalized voltage stress on the main switch and diodes, eliminating the reverse recovery problem. However, the converter's input and output are not grounded, increasing the difficulty of application. Furthermore, it suffers from an excessive number of electrical components, high cost, and low efficiency.

[0005] The paper "Y. Wang et al., “A Family of Y-Source DC / DC Converter Based on Switched Inductor,” IEEE Trans. Ind. Appl., vol. 55, no. 2, pp. 1587–1597, Mar. 2019, doi: 10.1109 / TIA.2018.2871658.” proposes a Y-source converter. Although this converter improves the voltage gain, the problem of high voltage stress on the switching transistor remains unresolved. Due to the limited number of turns in the coupling inductor, the converter's duty cycle range is less than 0.5.

[0006] The paper "B. Faridpak, M. Bayat, M. Nasiri, R. Samanbakhsh, and M. Farrokhhifar, “Improved hybrid switched inductor / switched capacitor DC–DC converters,” IEEE Trans. Power Electron., vol. 36, no. 3, pp. 3053–3062, Mar. 2021., proposes three improved non-isolated hybrid switched inductor / switched capacitor DC-DC converters by integrating active and passive switched inductors, switched capacitors, and auxiliary switching technologies. This converter features continuous inductor current and high voltage gain. However, the semiconductor devices in this converter experience significant current stress, and the use of switched inductors further increases the voltage stress on the semiconductor devices.

[0007] The 2-stage boost converter (QBC) evolved from cascaded boost converters, but it solves the fundamental problems of loss accumulation, stress concentration, and control complexity in cascaded structures through three core technologies: coupled inductor integration, shared multiplier units, and leakage inductor energy recovery. The core advantages of QBC include ultra-high gain, high efficiency, low stress, continuous input current, and a common ground structure.

[0008] The paper "Y. Wang, Y. Qiu, Q. Bian, Y. Guan, and D. Xu, “A single switch quadratic boost high step-up DC–DC converter,” IEEE Trans. Ind. Electron., vol. 66, no. 6, pp. 4387–4397, Jun. 2019.” proposes a converter combining a QBC, coupled inductor, and multiplier unit. This converter exhibits low voltage stress on its semiconductor devices. However, the use of a high-frequency transformer in its structure leads to higher converter losses and limitations in power density.

[0009] The paper "A. Ajami, H. Ardi, and A. Farakhor, “A novel high step-up DC / DC converter based on integrating coupled inductor and switched-capacitor techniques for renewable energy applications,” IEEE Trans. Power Electron., vol. 30, no. 8, pp. 4255–4263, Aug. 2015., proposes an ultra-high voltage gain converter with a similar structure to that described in "Y. Wang, Y. Qiu, Q. Bian, Y. Guan, and D. Xu, “A single switch quadratic boost high step-up DC–DC converter,” IEEE Trans. Ind. Electron., vol. 66, no. 6, pp. 4387–4397, Jun. 2019.,". It employs a coupled inductor and two switches, reducing voltage stress on the switches. While it offers advantages such as smaller required inductance, lower switching stress, and higher voltage gain, its structure has drawbacks. For example, excessive switching current stress in the converter during its operating mode can lead to increased losses. Summary of the Invention

[0010] This application provides a high-boost coupled-inductor DC-DC converter employing switched capacitors and coupled inductors. This converter cascades a switched capacitor structure onto a two-stage boost converter, thereby increasing the converter's gain. Simultaneously, the introduction of coupled inductors further enhances the gain, reduces voltage stress on the switching devices, achieves zero-crossing-switching (ZCS), recovers leakage inductance energy, and improves the converter's efficiency.

[0011] This application provides a high-boost coupled-inductor DC-DC converter employing switched capacitor and coupled-inductor technology, including an input inductor L. in The components include a coupling inductor, a switch S1, diodes D1, D2, D3, D4, D5 and D6, capacitors C1, C2 and C3, and output capacitors Co1 and Co2; the coupling inductor includes a magnetizing inductor L. m , primary leakage inductance L k And an ideal transformer, wherein the turns ratio of the ideal transformer is n=N2 / N1, and the coupling coefficient is k=L. m / (L m +L k The converter consists of a primary two-stage boost converter and a secondary switched capacitor structure cascaded together. The two-stage boost converter includes capacitors C1 and Co1, diode D3, and the primary side of an ideal transformer. The switched capacitor structure includes diodes D4, D5, and D6, capacitors C2 and C3, and an output capacitor Co2. Capacitors C2 and C3 are connected in parallel for charging and in series for discharging. The diodes in the switched capacitor unit achieve zero-current switching through the leakage inductance of the coupled inductor, and the leakage inductance energy is fed back to the load. The voltage gain of the converter in continuous operation mode satisfies formula (8): (8) In the formula, G CCM Voltage gain; D Duty cycle; n For variable ratios.

[0012] Furthermore, in continuous operation mode, the converter includes five operating modes within one switching cycle, namely: Mode 1: At time t0, switch S1 is turned on, capacitor C1 discharges, and energy is transferred from the primary winding of the coupled inductor to the secondary winding. The input voltage V in Through diode D1, the input inductor L in During charging, output capacitors Co1 and Co2 supply power to the load, and at time t1, the current of diodes D4-D5 drops to 0. Mode 2: At time t1, switch S1 remains on, diodes D4 and D5 achieve zero-current soft switching to turn off, capacitors C2 and C3 switch from charging to discharging, charging output capacitor Co2, and switch S1 turns off at time t2. Mode 3: At time t2, switch S1 and diode D1 are off, diodes D2 and D3 are on, and input inductor L... in Discharge, magnetizing inductor L m Energy received, leakage inductance L k Energy is output to capacitor Co1, and the secondary winding current is 0 at time t3; Mode 4: At time t3, diode D6 is turned off, and magnetizing inductor L... m Energy is released, charging capacitors C2 and C3 through the coupling inductor. Diodes D4 and D5 conduct. At time t4, the leakage inductance L... k The current drops to 0; Mode 5: Magnetized Inductor L m Energy is continuously released to charge the switched capacitor, the input current and secondary winding current decrease, and the next switching cycle begins at time t5.

[0013] Furthermore, the converter also includes an intermittent operating mode, when the magnetized inductor L... m When the current is discontinuous, it enters the discontinuous operating mode, and the voltage gain in the discontinuous operating mode satisfies formula (20): (20) In the formula, G DCM Voltage gain in discontinuous operation mode; τ Lm is the normalized time constant of the magnetizing inductance; The intermittent operating mode includes five operating modes within one switching cycle: mode 1, mode 2, mode 3, mode 4, and mode 5. Modes 1, 2, and 3 correspond to modes 1, 2, and 3 of the continuous operating mode, respectively. In mode 4, the magnetizing inductor L... m After the energy is released, the current i Lm Reduced to zero, in the fifth mode, the magnetizing inductance L m The energy is completely discharged, only diode D2 is turned on, the rest of the semiconductor devices are turned off, the input current continues to decrease, and the output capacitor supplies power to the load.

[0014] Furthermore, the converter also includes a boundary conduction mode, when the magnetized inductor L m When the current is critically continuous, it enters the boundary conduction mode. At this time, the voltage gain of the continuous operation mode and the intermittent operation mode are equal, and the normalized time constant of the boundary magnetizing inductance satisfies formula (21): (twenty one) In the formula, τ LmB is the normalized time constant of the boundary magnetization inductance.

[0015] Furthermore, the input inductor L in The design satisfies formula (22): (twenty two) In the formula, I L For input inductor current ripple; L For input inductance; f s The switching frequency; V in Input voltage; for; I L for; The magnetizing inductor L m The design satisfies formula (23): (twenty three) In the formula, I m The current ripple of the magnetized inductor; I m The current in the magnetizing inductor; R For output resistance; This is the voltage across the magnetizing inductance when the MOSFET is turned on; The coupled inductor has an air gap, and its effective permeability and corrected inductance coefficient satisfy formula (25), while the number of turns in the primary winding satisfies formula (26). (25) (26) In the formula, μ eff It is the effective permeability; μ r It is the relative permeability of the magnetic core; l m It is the length of the magnetic circuit; δ It is the air gap thickness; N 1 represents the number of turns in the primary winding; A L It is the ideal inductance coefficient; It is the inductance coefficient when an air gap exists; Furthermore, the design of capacitors C1, C2, and C3, and Co1 and Co2, satisfies formulas (27) and (28): (27) (28) In the formula, C This refers to the voltage ripple of the capacitor.f s The switching frequency; R For output resistance; V C This is the capacitor voltage; This is the capacitor voltage when the MOSFET is turned on; C 1 represents the capacitance value of capacitor number 1; C 2 represents the capacitance value of capacitor number 2; Co1 This is the capacitance value of output capacitor number 1; Co2 This is the capacitance value of output capacitor #2.

[0016] Furthermore, the power loss of the switch S1 satisfies formulas (29)-(36): (29) (30) (31) (32) (33) (34) (35) (36) In the formula, P S-COND This refers to the MOSFET conduction loss. I S This is the MOSFET on-state current; R S It is now at the MOSFET turn-on point; T ON This represents the on-time of the MOSFET. f S The switching frequency; P SW-ON This refers to the loss during the instant the MOSFET is turned on; V S This refers to the voltage between the source and drain terminals of the MOSFET. t a Ascending time; t b The duration of the upward trend; P SW-OFF This refers to the loss during the instantaneous turn-off of the MOSFET; t c The period of maintenance during the decline; t d For descent time; CISS For MOSFET input capacitor; R ON The on-resistance for MOSFET drive; R G Internal resistor for MOSFET drive; V GS This refers to the voltage between the gate and drain of the MOSFET. V TH This refers to the MOSFET gate threshold voltage. V GP This refers to the MOSFET gate plateau voltage. R OFF This is the MOSFET drive turn-off voltage; Q GD This refers to the gate-drain charge of the MOSFET. P S-LOSS This represents the total loss of the MOSFET.

[0017] Furthermore, the power losses of diodes D1, D2, D3, D4, D5, and D6 satisfy formulas (37)-(39): (37) (38) (39) In the formula, P Di-COND This refers to the diode's conduction loss. V FD This is the threshold voltage of the diode; I Di-AVE This represents the average diode current. I Di-rms This is the root mean square value of the diode current; r D This is the forward resistance of the diode; P SW-VDi This refers to diode switching losses; f S The switching frequency; V Di This is the diode voltage; I rr This is the reverse recovery current of the diode; t b This is the reverse recovery time of the diode; P D-LOSS This represents the total loss of the diode.

[0018] Furthermore, the power loss of the capacitor satisfies formula (40), the power loss of the magnetic element satisfies formulas (41)-(43), and the total loss and efficiency of the converter satisfy formulas (44)-(45): (40) (41) (42) (43) (44) (45) In the formula, P C-LOSS This is for capacitor losses; r Cj This is the equivalent series resistance of the capacitor; I Cj-rms This is the root mean square value of the capacitor current; P L-LOSS For input inductor power loss; r L The equivalent resistance of the input inductor; r Lj-rms This is the root mean square value of the input inductance; P L-LOSS Copper loss of the input inductor; P N-LOSS This refers to the copper loss of the coupled inductor; r N1 The resistance is the primary winding resistance of the coupled inductor; r N2 For the secondary winding resistance of the coupled inductor; I N1-rms This represents the root mean square value of the primary winding current. I N2-rms This represents the root mean square value of the secondary winding current. P MC-LOSS This represents the total power loss of the coupled inductor. P LOSS Total power loss of the converter; P S-LOSS This represents the total power loss of the MOSFET. P D-LOSS This represents the total power loss of the diode. For the efficiency of the converter; P O This refers to the output power.

[0019] Furthermore, the converter is scalable. By setting m secondary windings and inserting switched capacitor circuits into them respectively, the extended voltage gain satisfies formula (46): (46) In the formula, G m For the extended voltage gain; i This refers to the serial number of the secondary winding; m This refers to the number of secondary windings; n i This represents the ratio of the number of turns of each secondary winding to that of the primary winding.

[0020] The high-boost coupled-inductor DC-DC converter using switched capacitors and coupled inductors provided in this application has at least the following advantages: This application proposes a novel high-voltage gain converter, comprising a double-boost converter and a switched-capacitor structure. The proposed converter offers advantages such as continuous input current, shared input and output grounds, high voltage gain, and low switching stress. Simultaneously, the switched-capacitor structure utilizes leakage inductance to achieve zero-current switching (ZCS), recovering energy and reducing power loss. This converter can also be extended to achieve even higher voltage gains. This application experimentally verified the converter under conditions of 300V output voltage and 150W output power, demonstrating its feasibility. Loss analysis shows that the converter's efficiency can theoretically reach up to 95.2%. The feasibility of the aforementioned converter has been verified through experiments and analysis, demonstrating its potential application value in fields such as photovoltaic power generation and new energy vehicles. Attached Figure Description

[0021] The accompanying drawings, which are incorporated in and form part of this specification, illustrate embodiments consistent with this application and, together with the description, serve to explain the principles of this application.

[0022] Figure 1 A circuit diagram of a high boost coupled inductor DC-DC converter using switched capacitors and coupled inductors is provided for embodiments of this application. Figure 2 An equivalent circuit diagram of a high boost coupled inductor DC-DC converter using switched capacitors and coupled inductors is provided for embodiments of this application. Figure 3 A key waveform diagram of a high boost coupled inductor DC-DC converter using switched capacitors and coupled inductors under CCM operation, provided for embodiments of this application; Figure 4 The diagram illustrates the operating states of a high-boost coupled-inductor DC-DC converter employing switched capacitors and coupled inductors, as provided in this application embodiment; (a) Mode 1; (b) Mode 2; (c) Mode 3; (d) Mode 4; (e) Mode 5; Figure 5A key waveform diagram of a high boost coupled inductor DC-DC converter using switched capacitors and coupled inductors provided in this application embodiment under DCM operation; Figure 6 A schematic diagram of a high boost coupled inductor DC-DC converter using switched capacitors and coupled inductors, provided for embodiments of this application, in a special mode (fifth mode) within one switching cycle under DCM operation; Figure 7 Voltage gain provided for embodiments of this application G CCM A graph showing the relationship between duty cycle D, ratio n, and coupling coefficient k; Figure 8 A schematic diagram of the normalized boundary time constant of an inductor provided in an embodiment of this application; Figure 9 Detailed diagrams of the switch being turned on and off provided for embodiments of this application; Figure 10 An extended structure diagram of the converter provided in the embodiments of this application; Figure 11 The converter gain curves at different m values ​​when n=2 are provided in the embodiments of this application; Figure 12 A converter performance comparison diagram provided for embodiments of this application; wherein, (a), voltage gain; (b), normalized switching transistor voltage stress; (c), normalized output diode voltage stress; Figure 13 The experimental platform and prototype diagram provided in the embodiments of this application; Figure 14 Output power provided for embodiments of this application P o Output voltage at 150W V o Output current I o Input voltage V in Input current I in Experimental waveform diagram; Figure 15 Output power provided for embodiments of this application P o =150W capacitor C 1. C 2. Co1 , Co2 voltage V C1 , V C2 , V Co1 ,V Co2 Experimental waveform diagram; Figure 16 Output power provided for embodiments of this application P o Output voltage at 150W V o Diode D1 voltage V D1 Diode D1 current I D1 Experimental waveform diagram; Figure 17 Output power provided for embodiments of this application P o Output voltage at 150W V o Diode D2 voltage V D2 Diode D2 current I D2 Experimental waveform diagram; Figure 18 Output power provided for embodiments of this application P o =150W Drain voltage of switching transistor V DS Diode D3 voltage V D3 Diode D3 current I D3 Experimental waveform diagram; Figure 19 Output power provided for embodiments of this application P o =150W Drain voltage of switching transistor V D4 Diode D4 voltage V D4 Diode D4 current I D4 Experimental waveform diagram; Figure 20 Output power provided for embodiments of this application P o =150W Drain voltage of switching transistor V D6 Diode D6 voltage V D6 Diode D6 current I D6 The experimental waveform diagram.

[0023] The accompanying drawings illustrate specific embodiments of this application, which will be described in more detail below. These drawings and descriptions are not intended to limit the scope of the concept in any way, but rather to illustrate the concept of this application to those skilled in the art through reference to particular embodiments. Detailed Implementation

[0024] Exemplary embodiments will now be described in detail, examples of which are illustrated in the accompanying drawings. When the following description relates to the drawings, unless otherwise indicated, the same numbers in different drawings denote the same or similar elements. The embodiments described in the following exemplary embodiments do not represent all embodiments consistent with this application. Rather, they are merely examples of apparatuses and methods consistent with some aspects of this application as detailed in the appended claims.

[0025] The collection, storage, use, processing, transmission, provision, and disclosure of financial data or user data involved in the technical solution of this application all comply with the provisions of relevant laws and regulations and do not violate public order and good morals.

[0026] It should be noted that in the embodiments of this application, certain software, components, models and other existing solutions in the industry may be mentioned. These should be regarded as exemplary and are only intended to illustrate the feasibility of implementing the technical solution of this application. However, they do not mean that the applicant has used or necessarily used the solution.

[0027] The technical solution of this application and how the technical solution of this application solves the above-mentioned technical problems are described in detail below with specific embodiments. These specific embodiments can be combined with each other, and the same or similar concepts or processes may not be described again in some embodiments. The embodiments of this application will now be described with reference to the accompanying drawings.

[0028] This application provides a high-boost coupled-inductor DC-DC converter employing switched capacitor and coupled-inductor technology. Specifically, it is a coupled-inductor DC-DC converter that combines integrated two-stage boost, switched capacitor, and coupled-inductor technologies to achieve high voltage gain, low switching stress, and high efficiency. Figure 1 As shown, the converter utilizes the leakage inductance of the coupled inductor to enable zero-current switching (ZCS) of the diodes in the switched capacitor unit. Simultaneously, the energy in the leakage inductance can be fed back to the load, improving the converter's efficiency. The proposed converter is scalable; the voltage gain can be increased by changing the duty cycle, transformer turns ratio, and the number of secondary windings. It not only achieves high voltage gain but also maintains low voltage stress on the semiconductor devices within the converter.

[0029] The following examples are detailed in Example 1: Continuous Continuous Mode (CCM) and Discontinuous Continuous Mode (DCM) of the converter; Example 2: Steady-state analysis of the converter; Example 3: Component parameter design and power loss calculation; Example 4: Extended structure and performance comparison of the converter; Example 5: Experiments and analysis.

[0030] Example 1: In this embodiment, some assumptions are considered to simplify the circuit analysis of the proposed converter: 1) All components are ideal, and the parasitic capacitance and on-resistance of switch S1 and all capacitors are ignored. The forward voltage drop of the diodes is also ignored.

[0031] 2) All capacitors are large enough that their voltage is considered constant over one cycle.

[0032] 3) Coupled inductance is represented as magnetizing inductance L m 1. Primary leakage inductance L k And an ideal transformer. Its coupling coefficient k is approximately equal to L. m / (L m +L k The turns ratio n of the coupled inductor is equal to N2 / N1.

[0033] The equivalent circuit model of the proposed converter is as follows: Figure 2 As shown. This converter consists of a secondary boost converter as the primary section and switched capacitors as the secondary section. The secondary boost section includes two capacitors C1 and Co1, diode D3, and the primary side of an N1-turn coupled inductor. The switched capacitor section includes diodes D4, D5, and D6, and capacitors C2, C3, and Co2. These capacitors charge in parallel and discharge in series. When the secondary current of the coupled inductor becomes positive, C2 and C3 charge in parallel; conversely, when the secondary current of the coupled inductor becomes negative, C2 and C3 discharge in series. Input inductor L in The current is i in Because the input current is continuous, i in ≥0 holds true in every steady-state switching cycle.

[0034] The key waveforms of the main components of the proposed converter when it is in CCM steady-state operation are as follows: Figure 3 As shown, there are 5 operating modes within one switching cycle, and the current flow path in each mode is as follows. Figure 4 As shown in (a) - (e).

[0035] 1) Mode 1 [t0, t1]: At t=t0, switch S1 is turned on, capacitor C1 discharges, and energy is transferred from the primary winding of the coupled inductor to the secondary winding. In this transition mode, the magnetized inductor L... m It continuously releases energy, and its current iLm It continues to decrease, i Lk Increase. Current path as follows Figure 4 As shown in (a), when diode D1 is turned on, Vin charges Lin, and the current i in The current rises. This is because the primary winding current of the coupled inductor rises, and the secondary winding current rises by i... Lm As the current decreases at the same rate as / n, the charging current of diodes D3 and D4 to the switched capacitor also decreases. Output capacitors Co1 and Co2 continue to provide energy to the load. At t=t1, i D4 with i D5 When the value drops to 0, this mode ends.

[0036] 2) Mode 2 [t1, t2]: At t=t1, switch S1 is already turned on, and diodes D4 and D5 achieve zero-current soft-switching (ZCS) turn-off. Magnetizing inductor L m The primary leakage inductance Lk is connected in series with capacitor C1, and begins to release the energy of C1, with a current i Lm i Lk Increase, and i C1 Decrease. Current path as follows Figure 4 As shown in (b), the direction of the secondary winding current changes, and switched capacitors C2 and C3 switch from charging to discharging, increasing the current iD6 to charge capacitor Co2. Power supply Vin charges Lin through diode D1 and switch S1, with the current increasing linearly. Output capacitors Co1 and Co2 continuously release energy to the load. When t=t2, and the PWM signal abruptly drops to 0, switch S1 is turned off. This mode ends.

[0037] 3) Mode 3 [t2, t3]: At t=t2, switch S1 and diode D1 are off, while D2 and D3 are on. The current path is as follows: Figure 4 As shown in (c), the switched capacitor continues to discharge. The inductor Lin discharges, its voltage polarity reverses, and the current i... in Continuously decreasing. Magnetizing inductance L m Receive energy from the front end, i Lm Continue to increase. Because D3 is conducting, the leakage inductance Lk outputs energy to capacitor Co1, whose voltage is also negative, and the current i... Lk It continues to decrease. Therefore, the primary winding current decreases with i. Lk The current decreases as the leakage inductance decreases. Because the leakage inductance is relatively small, its current i... Lk The descent rate is relatively fast. The output capacitor provides energy to the load. When i decreases... Lk Become equal to the increase of i Lm When the secondary winding current is 0, the mode ends.

[0038] 4) Mode 4 [t3, t4]: The secondary winding current crosses zero at t=t3, and diode D6 is turned off. The current flow path is as follows: Figure 4 As shown in (d), the magnetizing inductor Lm releases energy in the reverse direction, which is transferred to the secondary winding via the coupling inductor to charge C2 and C3. D4 and D5 are turned on, and i Lm Decrease. Because the voltages of Lin and Lk are reversed, the current i in and i Lk As the voltage decreases linearly, diodes D2 and D3 remain conducting, while capacitors C1 and Co1 charge. The energy stored in the output capacitor is transferred to the load. When i Lk When the value decreases to 0, the mode ends.

[0039] 5) Mode 5 [t4, t5]: Current flow path as follows Figure 4 As shown in (e). During this time, the magnetized inductor Lm releases energy, i Lm The input current decreases. The induced current in the secondary winding charges the switched capacitor through diodes D4 and D5. in Both the secondary winding current and the output capacitor reduce. The output capacitor provides energy to the load. This mode ends at t=t5, which also corresponds to the start of a new switching cycle. After that, the operating mode repeats.

[0040] The proposed converter input current i in Continuous, defined in the magnetizing inductance L m Current i Lm When discontinuous switching occurs, the circuit is in Discontinuous Conversion Mode (DCM). Within one switching cycle, this mode can be divided into five modes: Mode 1, Mode 2, Mode 3, Mode 4, and Mode 5. The first three modes (Mode 1, Mode 2, and Mode 3) are identical to Modes 2, 3, and 4 in Discontinuous Conversion Mode (CCM). Mode 4 in DCM is essentially the same as Mode 5 in CCM, the difference being that in Mode 4 of DCM, the magnetized inductor Lm releases all its energy, and the current i... Lm The secondary winding current also drops to zero, diodes D4 and D5 are cut off, and the switched capacitor stops charging. The key waveforms of the main components of the proposed converter in DCM operation are as follows: Figure 5 As shown. The last mode (the fifth mode) is analyzed in detail below: Fifth mode [t4, t5]: When t=t4, the magnetizing inductance L m Energy is completely discharged. Only D2 is conducting; all other semiconductor devices are off. The current path is as follows: Figure 6 As shown. Input current i in The voltage continues to decrease. The output capacitor provides energy to the load. When t=t5, the next switching cycle begins, and this mode ends.

[0041] Example 2: To simplify the analysis, when analyzing the voltage gain of the proposed converter in CCM operation, the transient modes caused by leakage inductance and switching parasitic capacitance within one switching cycle are ignored, and only modes 2 and 4 are considered. Figure 4 (b) According to Kirchhoff's voltage law, the following equation can be obtained: (1) In the formula, V Lin The voltage at which the input inductor is applied; V in Input voltage; V Lm This is the voltage across the magnetizing inductor; V C1 The voltage across capacitor 1; V Co1 This is the voltage across output capacitor 1; V C2 The voltage across capacitor 2; V C3 The voltage across capacitor 3 is given.

[0042] like Figure 4 As shown in (d), when the converter operates in mode 4, the following equation can be obtained: (2) Based on Mode 2 and Mode 4, the following equation can also be derived: (3) In the formula, V O This is the output voltage.

[0043] Based on the volt-second balance of the inductor, substituting (1) and (2) into the equation yields the following formula: (4) (5) (6) In the formula, DT s It is the product of duty cycle and period.

[0044] The capacitor voltage can be obtained from (1), (2), (4), (5), and (6): (7) From (3), we know the output voltage V o The magnitude is the sum of the voltage across the output capacitor. Substituting (7) into (3) yields the voltage gain. G CCM : (8) In the formula, G CCM Voltage gain; D Duty cycle; n For variable ratios.

[0045] Figure 7 Demonstrated voltage gain G CCM The relationship between duty cycle D, turns ratio n, and coupling coefficient k shows that the voltage gain increases with increasing k, n, and D. When the turns ratio n is constant, different values ​​of k have little effect on the voltage gain.

[0046] Figure 8 The curves showing the converter gain at different turn ratios and different duty cycles are displayed.

[0047] For ease of analysis, let k=1, then the voltage gain... G CCM Represented as: (9) Table 1 Voltage and Current Stress

[0048] Assuming the capacitor voltage remains constant, in modes 2 and 4, Kirchhoff's voltage law is applied according to the capacitor voltage formula (7). Therefore, the voltages of the input inductor and the coupling inductor when the switch is turned on and off can be calculated as follows: (10) In the formula, V Lin_on This is the voltage across the input inductor when the MOSFET is turned on; V Lin_off This is the voltage across the input inductor when the MOSFET is turned off; V N1_on This is the voltage across the primary winding of the coupled inductor when the MOSFET is turned on; V N1_off This is the voltage across the primary winding of the coupled inductor when the MOSFET is turned off; V N2_on This is the voltage of the secondary winding of the coupled inductor when the MOSFET is turned on; V N2_off This is the voltage of the secondary winding of the coupled inductor when the MOSFET is turned off.

[0049] pass Figure 4 For (b) and (d), the capacitor current can be expressed by the following equation based on Kirchhoff's current law: (11) In the formula, I C1_on This represents the current of capacitor 1 when the MOSFET is turned on; I C1_off This represents the current in capacitor 1 when the MOSFET is turned off. I D1 This represents the current in diode 1; I S This represents the current of the MOSFET. I D2 This represents the current in diode 2; I D3 This represents the current in diode 3; I D4 This represents the current of diode 4; I D5 This represents the current of diode 5; I C2_on This represents the current flowing through capacitor 2 when the MOSFET is turned on. I C2_off This is the current in capacitor 2 when the MOSFET is turned off; I C3_on This represents the current flowing through capacitor 3 when the MOSFET is turned on. I C3_off This represents the current in capacitor 3 when the MOSFET is turned off. I Co1_on This is the current of output capacitor 1 when the MOSFET is turned on; I Co1_off This is the current of output capacitor 1 when the MOSFET is turned off; I Co2_on This is the current of output capacitor 2 when the MOSFET is turned on; I Co2_off This is the current of output capacitor 2 when the MOSFET is turned off.

[0050] Under steady state, the following formula can be established based on the principle of capacitor charge balance: (12) In the formula, I Ci_on This represents the current flowing through the capacitor when the MOSFET is turned on. I Ci_off This represents the current flowing through the capacitor when the MOSFET is turned off.

[0051] According to the law of conservation of energy, the proposed converter loop current satisfies the following relationship: (13) In the formula, IO This is the output current.

[0052] Combining (10), (11), (12), and (13), the voltage and current stresses on the capacitor, diode, and switch can be obtained. The specific stress details of the proposed converter are summarized in Table 1.

[0053] To simplify the analysis, the effect of leakage inductance is ignored and the capacitor voltage is assumed to be constant. Additionally, due to the short transition time in Mode 2, it is neglected for calculation purposes. D, D a D b like Figure 5 The figures show the duty cycles of the proposed converter in modes 1, 3, and 4 during DCM operation. Because the input current i in Continuous, applying the volt-second balance law to the input inductor Lin, the expression for the voltage across capacitor C1 is the same as under CCM. Applying the volt-second balance law to its coupled inductor as well, the output voltage can be expressed as: (14) In the formula, D a and D b This represents the duty cycle for modes 3 and 4.

[0054] Since the leakage inductance effect is ignored, the maximum current through D3 can be considered to be the same as the maximum current through Lm, and their relationship is shown in (15). Applying ampere-second balancing to the output capacitor Co1, the duty cycle D of mode 3 during DCM operation can be obtained. a As shown in (16): (15) (16) In the formula, I Lmp_DCM This represents the maximum current flowing through Lm during the DCM period; RT S It is the product of the output resistance and the period.

[0055] The average current of diode D5 is I. o At the same time I D5 The maximum value satisfies (17), therefore we can conclude that D b Value: (17) (18) In the formula, i D5p_DCM This represents the maximum current flowing through diode 5 during the DCM period.

[0056] Normalized time constant of magnetizing inductor τ Lm Defined as (19). Substituting (16), (18), and (19) into (14) yields the voltage gain G of the proposed converter under DCM. DCM for: (19) (20) In the formula, G DCM Voltage gain in discontinuous operation mode; τ Lm is the normalized time constant of the magnetizing inductance.

[0057] When the current in the magnetizing inductor Lm is in a critical continuous state, the proposed converter operates in BCM mode. In BCM mode, the voltage gain is equal in CCM and DCM modes. Based on the gain expressions shown in (9) and (20), the normalized time constant of the boundary magnetizing inductor is... It can be represented as: (twenty one) Figure 8 The normalized time constant of the boundary magnetization inductance of the proposed converter is expressed. τ LmB The curve relationship between the duty cycle D and the duty cycle. When τ Lm Greater than τ LmB At that time, the proposed converter was in CCM operation mode.

[0058] Example 3: To verify the correctness and feasibility of the theoretical analysis of the proposed converter, this embodiment provides design guidelines for the input inductor, coupling inductor, and capacitor components in the circuit.

[0059] 1) Input Inductance: To reduce the impact of input current ripple on the circuit, the input inductor current ripple is usually defined as... I L =20% I in Therefore, the formula for calculating the low-current ripple input inductance is: (twenty two) In the formula, I L For input inductor current ripple; L For input capacitance; f s The switching frequency; Vin Input voltage; This is the input inductor voltage when the MOSFET is turned on; I L This is the input inductor current.

[0060] 2) Magnetized Inductor: Similarly, under CCM operating conditions, the current ripple of the magnetized inductor is defined as... I Lm =20%I Lm Therefore, the formula for calculating magnetizing inductance can be obtained. (twenty three) In the formula, I m The current ripple of the magnetized inductor; I m The current in the magnetizing inductor; R For output resistance; This is the voltage of the magnetizing inductance when the MOSFET is turned on.

[0061] A of the magnetic core p The value can be calculated according to the following equation: (twenty four) In the formula, I peak It is the peak value of the magnetizing inductor current, which can be calculated using the following formula: I peak = I Lm + i Lm / 2. W represents energy processing capacity. Assume a window coefficient. k u = 0.4, magnetic flux density B m = 0.5T, current density J = 3 A / mm 2 According to formula (24), the following can be calculated: Ap = 2.904 cm 4 .

[0062] In high-current and high-frequency operating scenarios, an air gap is typically incorporated into coupled inductors to balance inductor performance and reliability. The air gap increases magnetic reluctance and reduces magnetic flux density, thus preventing core saturation. The size of the air gap affects A. L The value of this value, in turn, affects the number of turns in the winding.

[0063] (25) In the formula, μ eff It is the effective permeability. μ r It is the relative permeability of the magnetic core. l m It is the length of the magnetic circuit. δ It refers to the air gap thickness.

[0064] A' can be obtained by calculation according to (25). L Therefore, the number of turns of the primary winding N1 can be calculated by equation (26): (26) In the formula, A L It is the ideal inductance coefficient; It is the inductance coefficient when the air gap exists.

[0065] 3) Capacitors: Voltage stress and voltage ripple should be considered in capacitor design. The voltage ripple of a capacitor is typically... C =2% V C .

[0066] The method for selecting the capacitance value is as follows: (27) We obtain the information from Table 1 and formula (27). (28) In the formula, C This refers to the voltage ripple of the capacitor. f s The switching frequency; R For output resistance; V C This is the capacitor voltage; This is the capacitor voltage when the MOSFET is turned on; C 1 represents capacitor 1; C 2 represents capacitor 2; Co1 Output capacitor 1; Co2 The output capacitor is 2.

[0067] Table 2 Component Parameters

[0068] Detailed device parameters and definitions are summarized in Table 2. Switching losses include conduction losses and switching losses. Switching losses further include turn-on losses and turn-off losses.

[0069] (29) (30) (31) In the formula, P S-COND This refers to the MOSFET conduction loss. I S This is the MOSFET on-state current; R S It is now at the MOSFET turn-on point; T ON This represents the on-time of the MOSFET. f S The switching frequency; P SW-ON This refers to the loss during the instant the MOSFET is turned on; V S This refers to the voltage between the source and drain terminals of the MOSFET. t a Ascending time; t b The duration of the upward trend; P SW-OFF This refers to the loss during the instantaneous turn-off of the MOSFET; t c The period of maintenance during the decline; t d This refers to the descent time.

[0070] Figure 9 This indicates detailed information about the switch-on process; the specific process at the moment of turn-off is similar. Therefore, it can be known that the turn-off process involves a maintenance time tc and a fall time t. d The time t is obtained by calculating the moments when the Miller platform turns on and off. a and t d .

[0071] (32) (33) In the formula, C ISS For MOSFET input capacitor; R ON The on-resistance for MOSFET drive; R G Internal resistor for MOSFET drive; V GS This refers to the voltage between the gate and drain of the MOSFET. V TH This refers to the MOSFET gate threshold voltage. VGP This refers to the MOSFET gate plateau voltage. R OFF This is the MOSFET drive turn-off voltage.

[0072] Duration t b and t c The calculation formula is as follows: (34) (35) In the formula, Q GD This refers to the gate-drain charge of the MOSFET.

[0073] The power loss of the switch is: (36) In the formula, P S-LOSS This represents the total efficiency loss of the MOSFET.

[0074] Diode losses include conduction losses and switching losses. Since diodes D4, D5, and D6 achieve ZCS (Zero-Cyclic Switching), their switching losses are not considered.

[0075] (37) (38) In the formula, P Di-COND This refers to the diode's conduction loss. V FD This is the threshold voltage of the diode; I Di-AVE This represents the average current of the diode; I Di-rms This is the root mean square value of the diode current; r D This is the forward resistance of the diode; P SW-VDi For the switching losses of the MOSFET; f S The switching frequency; V Di Auntie is a diode; I rr This is the reverse recovery current of the diode.

[0076] Therefore, the sum of the power losses of the diodes is: (39) In the formula, P D-LOSS This represents the total power loss of the diode.

[0077] The total loss on the capacitor is: (40) In the formula, P C-LOSS This is for capacitor losses; r Cj This is the equivalent series resistance of the capacitor; I Cj-rms This is the root mean square value of the capacitor current.

[0078] Next, the losses on the input inductor and the coupled inductor are calculated. The losses of magnetic components mainly include copper losses and iron losses. For the purpose of simplifying the analysis, this embodiment ignores the analysis of iron core losses, which are difficult to measure and have relatively small losses. The calculation method for copper losses of magnetic components is as follows: (41) (42) (43) In the formula, P L-LOSS For input inductor power loss; r L The equivalent resistance of the input inductor; r Lj-rms This is the root mean square value of the input inductance; P L-LOSS Copper loss of the input inductor; P N-LOSS This refers to the copper loss of the coupled inductor; r N1 The resistance is the primary winding resistance of the coupled inductor; r N2 For the secondary winding resistance of the coupled inductor; I N1-rms This represents the root mean square value of the primary winding current. I N2-rms This represents the root mean square value of the secondary winding current. P MC-LOSS This represents the total power loss of the coupled inductor. P LOSS Total power loss of the converter; P S-LOSS This represents the total power loss of the MOSFET.

[0079] Therefore, the total loss of the proposed converter can be calculated as follows: (44) In the formula, P LOSS This represents the total power loss of the converter. P S-LOSS For MOSFET efficiency losses;P D-LOSS This represents the total power loss of the diode.

[0080] The efficiency of the proposed converter is expressed as follows: (45) In the formula, For converter efficiency; P O This refers to the output power.

[0081] Example 4: The converter proposed in this embodiment is scalable. For example... Figure 10 As shown, by setting up multiple coupled inductors, the output gain of the converter is improved by inserting each secondary winding into a switched capacitor circuit. In this case, the proposed converter has an additional degree of freedom *m* to control the converter gain. *m* is the number of secondary windings. The extended converter gain is: (46) In the formula, n i This represents the turns ratio of each secondary winding to the primary winding.

[0082] Voltage gain such as Figure 11 As shown, increasing the number of secondary windings significantly improves the converter gain. However, this also increases the number of components, converter size, and losses. Therefore, it is necessary to consider the actual situation and rationally set the number of secondary windings during application.

[0083] Table 3 Comparison of the proposed converter with other converters

[0084] It should be noted that the representations

[24] -

[32] respectively represent:

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[25] : Literature "S. Esmaeili, M. Shekari, M. Rasouli, S. Hasanpour, A. A. Khan, and H. Hafezi, ‘High Gain Magnetically Coupled Single Switch Quadratic Modified SEPIC DC-DC Converter’, IEEE Trans. Ind. Appl., vol. 59, no. 3, pp. 3593–3604, May 2023, doi: 10.1109 / TIA.2023.3250405.";

[26] : Literature "D. Sadeghpour and J. Bauman, “High-effificiency coupled-inductor switched-capacitor boost converter with improved input current ripple,” IEEE Trans. Ind. Electron., vol. 69, no. 8, pp. 7940-7951, Aug. 2022";

[27] : Literature "J. Ai and M. Lin, “Ultralarge gain step-up coupled-inductor DC–DC converter with an asymmetric voltage multiplier network for a sustainable energy system,” IEEE Trans. Power Electron., vol. 32, no. 9, pp. 6896–6903, Sep. 2017.";

[28] : Literature "M. Eskandarpour Azizkandi, F. Sedaghati, H. Shayeghi, and F. Blaabjerg, “Two-and three-winding coupled-inductor-based high step-up DC–DC converters for sustainable energy applications,” IET Power Electron., vol. 13, no. 1, pp. 144–156, 2020.";

[29] : Literature "T.-J. Liang, P. Luo, and K.-H. Chen, “A high step-up DC–DC converter with three-winding coupled inductor for sustainable energy systems,” IEEE Trans. Ind. Electron., vol. 69, no. 10, pp. 10249–10258, Oct. 2022.";

[30] : Literature "S. Gao, Y. Wang, Y. Liu, Y. Guan, and D. Xu, “A novel DCM soft-switched SEPIC-based high-frequency converter with high step-up capacity,” IEEE Trans. Power Electron., vol. 35, no. 10, pp. 10444–10454, Oct. 2020, doi: 10.1109 / TPEL.2020.2975130.";

[31] : Literature "Y. Tang, H. Tong, R. Afzal, and Y. Guo, ‘High Step-Up ZVT Converter Based on Active Switched Coupled Inductors’, IEEE Access, vol. 13, pp. 15178–15187, 2025, doi: 10.1109 / ACCESS.2020.3041004.";

[32] : Literature "S.-W. Lee and H.-L. Do, ‘Quadratic Boost DC–DC Converter With High Voltage Gain and Reduced Voltage Stresses’, IEEE Trans. Power Electron., vol. 34, no. 3, pp. 2397–2404, Mar. 2019, doi: 10.1109 / TPEL.2018.2842051.";

[0085] To illustrate the superior performance of the proposed converter, Table 3 compares it with nine other converters. Key performance parameters include the number of components and converter gain. The proposed converter boasts advantages such as high voltage gain, wide duty cycle adjustment range, low voltage stress on switches and diodes, high efficiency, continuous input current, and shared ground. However, its drawbacks include a relatively large number of components, resulting in higher overall power loss at high power output.

[0086] exist Figure 12 The image shows the voltage gain, switching transistor voltage stress, and output diode voltage stress curves for these ten converters. To ensure a fair comparison, the turns ratio n=2 for the coupled inductors with two windings in the converter. For the coupled inductors with three windings, the converter turns ratio is set to n2=0.5 and n3=1.5. After setting the coupled inductor turns ratio, Figure 12 Figure (a) shows the relationship between the converter voltage gain and duty cycle. It can be seen that the proposed converter's voltage gain is significantly higher than other converters when D > 0.4, and the voltage gain rise rate is also high. Furthermore, under the same conditions, the normalized voltage stress curves of the proposed converter and other converter switches are shown below. Figure 12 As shown in (b), the proposed converter exhibits relatively low switching stress, placing it in the upper-middle range compared to the other nine converters. Figure 12 (c) compares the voltage stress of the converter output diode. It can be seen that the proposed converter output diode has very low voltage stress. The voltage stress of its output diode decreases with increasing duty cycle. Based on this, a diode with lower on-resistance can be selected appropriately in different application environments to reduce device losses.

[0087] Example 5: To verify the superiority of the proposed converter, a 150W experimental prototype was built. The experimental platform is as follows: Figure 13 As shown in Table 4, the detailed experimental parameters and device selection are described in Table 4. The experimental waveforms for the 150W prototype at Vin=24V and Vo=300V are shown below. Figures 14 to 20 As shown. Figure 14 The input voltage Vin = 24V, the input current Iin = 6.9A, the output voltage Vo = 301.6V, and the output current Io = 0.5A. The analysis yields a boost ratio of 12.57, which is consistent with the analysis result of formula (9), and the experimental efficiency is 91%.

[0088] Figure 15 The voltages across capacitors C1, C2, Co1, and Co2 are displayed, with the voltages fluctuating within a 20% range. The experimental waveforms agree well with the theoretical analysis. Figure 16 and Figure 17This shows the voltage and current characteristics of diodes D1 and D2 when the output voltage Vo = 300V. It can be seen that their conduction status and values ​​are consistent with the analysis results.

[0089] Figure 18 The switching voltage V DS The voltage and current characteristics of diode D3 are shown. Due to its low leakage inductance, the current in diode D3 decreases rapidly, achieving ZCS (Zero-Cost Conversion), which further improves the converter's efficiency. Figure 19 and Figure 20 This represents the switching transistor voltage, and the voltage and current of diodes D4 and D6. It is clear that the current drops to zero before the diodes are turned off, achieving zero-current-slip (ZCS). Therefore, except for diodes D1 and D2, the losses of other diodes are mainly conduction losses. Because the parasitic inductance of the diode D6 circuit in the experimental prototype is relatively large, a large reverse recovery current spike will be generated when the diode is turned off. This should be considered during the prototype fabrication stage.

[0090] Table 4 Main parameters of the experimental prototype

[0091] In summary, this application proposes a novel high-voltage gain converter, which consists of a double-boost converter and a switched-capacitor structure. The proposed converter offers advantages such as continuous input current, shared input and output grounds, high voltage gain, and low switching stress. Simultaneously, the switched-capacitor structure utilizes leakage inductance to achieve zero-current switching (ZCS), recovering energy and reducing power loss. This converter can also be extended to achieve even higher voltage gains. This embodiment experimentally verifies the converter under conditions of 300V output voltage and 150W output power, demonstrating its feasibility. Loss analysis shows that the converter's efficiency can theoretically reach up to 95.2%. The feasibility of the aforementioned converter has been verified through experiments and analysis, demonstrating its potential application value in fields such as photovoltaic power generation and new energy vehicles.

[0092] Finally, it should be noted that the above embodiments are only used to illustrate the technical solutions of this application, and are not intended to limit them. Although this application has been described in detail with reference to the foregoing embodiments, those skilled in the art should understand that modifications can still be made to the technical solutions described in the foregoing embodiments, or equivalent substitutions can be made to some or all of the technical features therein. Such modifications or substitutions do not cause the essence of the corresponding technical solutions to deviate from the scope of the technical solutions of the embodiments of this application.

Claims

1. A high-boost coupled-inductor DC-DC converter employing switched capacitors and coupled inductors, characterized in that, Including input inductance L in The components include a coupling inductor, a switch S1, diodes D1, D2, D3, D4, D5 and D6, capacitors C1, C2 and C3, and output capacitors Co1 and Co2; the coupling inductor includes a magnetizing inductor L. m , primary leakage inductance L k And an ideal transformer, wherein the turns ratio of the ideal transformer is n=N2 / N1, and the coupling coefficient is k=L. m / (L m +L k The converter consists of a primary two-stage boost converter and a secondary switched capacitor structure cascaded together. The two-stage boost converter includes capacitors C1 and Co1, diode D3, and the primary side of an ideal transformer. The switched capacitor structure includes diodes D4, D5, and D6, capacitors C2 and C3, and an output capacitor Co2. Capacitors C2 and C3 are connected in parallel for charging and in series for discharging. The diodes in the switched capacitor unit achieve zero-current switching through the leakage inductance of the coupled inductor, and the leakage inductance energy is fed back to the load. The voltage gain of the converter in continuous operation mode satisfies formula (8): (8) In the formula, G CCM Voltage gain; D Duty cycle; n For variable ratios.

2. The high-boost coupled-inductor DC-DC converter using switched capacitors and coupled inductors as described in claim 1, characterized in that, In continuous operation mode, the converter includes five operating modes within one switching cycle, namely: Mode 1: At time t0, switch S1 is turned on, capacitor C1 discharges, and energy is transferred from the primary winding of the coupled inductor to the secondary winding. The input voltage V in Through diode D1, the input inductor L in During charging, output capacitors Co1 and Co2 supply power to the load, and at time t1, the current of diodes D4-D5 drops to 0. Mode 2: At time t1, switch S1 remains on, diodes D4 and D5 achieve zero-current soft switching to turn off, capacitors C2 and C3 switch from charging to discharging, charging output capacitor Co2, and switch S1 turns off at time t2. Mode 3: At time t2, switch S1 and diode D1 are off, diodes D2 and D3 are on, and input inductor L... in Discharge, the magnetizing inductor Lm receives energy, and the leakage inductance L k Energy is output to capacitor Co1, and the secondary winding current is 0 at time t3; Mode 4: At time t3, diode D6 is turned off, and magnetizing inductor L... m Energy is released, charging capacitors C2 and C3 through the coupling inductor. Diodes D4 and D5 conduct. At time t4, the leakage inductance L... k The current drops to 0; Mode 5: Magnetized Inductor L m Energy is continuously released to charge the switched capacitor, the input current and secondary winding current decrease, and the next switching cycle begins at time t5.

3. The high-boost coupled-inductor DC-DC converter using switched capacitors and coupled inductors as described in claim 2, characterized in that, The converter also includes an intermittent operating mode, when the magnetized inductor L... m When the current is discontinuous, it enters the discontinuous operating mode, and the voltage gain in the discontinuous operating mode satisfies formula (20): (20) In the formula, G DCM Voltage gain in discontinuous operation mode; τ Lm is the normalized time constant of the magnetizing inductance; The intermittent operating mode includes five operating modes within one switching cycle: mode 1, mode 2, mode 3, mode 4, and mode 5. Modes 1, 2, and 3 correspond to modes 1, 2, and 3 of the continuous operating mode, respectively. In mode 4, the magnetizing inductor L... m After the energy is released, the current i Lm Reduced to zero, in the fifth mode, the magnetizing inductance L m The energy is completely discharged, only diode D2 is turned on, the rest of the semiconductor devices are turned off, the input current continues to decrease, and the output capacitor supplies power to the load.

4. The high-boost coupled-inductor DC-DC converter using switched capacitors and coupled inductors as described in claim 1, characterized in that, The converter also includes a boundary conduction mode, when the magnetized inductor L m When the current is critically continuous, it enters the boundary conduction mode. At this time, the voltage gain of the continuous operation mode and the intermittent operation mode are equal, and the normalized time constant of the boundary magnetizing inductance satisfies formula (21): (21) In the formula, τ LmB is the normalized time constant of the boundary magnetization inductance.

5. The high-boost coupled-inductor DC-DC converter using switched capacitors and coupled inductors as described in claim 1, characterized in that, The input inductance L in The design satisfies formula (22): (22) In the formula, I L For input inductor current ripple; L For input inductance; f s To control the frequency of the PWM signal; V in Input voltage; This is the voltage across the input inductor when the switching transistor is turned on; I L For input inductor current; The magnetizing inductor L m The design satisfies formula (23): (23) In the formula, I m The current ripple of the magnetized inductor; I m The current in the magnetizing inductor; R For output resistance; This is the voltage across the magnetizing inductor when the switching transistor is turned on; The coupled inductor has an air gap, and its effective permeability and corrected inductance coefficient satisfy formula (25), while the number of turns in the primary winding satisfies formula (26). (25) (26) In the formula, μ eff It is the effective permeability; μ r It is the relative permeability of the magnetic core; l m It is the length of the magnetic circuit; δ It is the air gap thickness; N 1 represents the number of turns in the primary winding; A L It is the ideal inductance coefficient; It is the inductance coefficient when the air gap exists.

6. The high-boost coupled-inductor DC-DC converter using switched capacitors and coupled inductors as described in claim 1, characterized in that, The design of capacitors C1, C2, and C3, and Co1 and Co2, satisfies formulas (27) and (28): (27) (28) In the formula, C This refers to the voltage ripple of the capacitor. f s To control the frequency of the PWM signal; R For output inductance; V C This is the capacitor voltage; This is the capacitor voltage when the switching transistor is turned on; C 1 represents capacitor 1; C 2 represents capacitor 2; Co1 Output capacitor 1; Co2 The output capacitor is 2.

7. The high-boost coupled-inductor DC-DC converter using switched capacitors and coupled inductors as described in claim 1, characterized in that, The power loss of the switch S1 satisfies formulas (29)-(36): (29) (30) (31) (32) (33) (34) (35) (36) In the formula, P S-COND This refers to the MOSFET conduction loss. I S This is the MOSFET on-state current; R S It is now at the MOSFET turn-on point; T ON This represents the on-time of the MOSFET. f S The switching frequency; P SW-ON This refers to the loss during the instant the MOSFET is turned on; V S This refers to the voltage between the source and drain terminals of the MOSFET. t a Ascending time; t b The duration of the upward trend; P SW-OFF This refers to the loss during the instantaneous turn-off of the MOSFET; t c The period of maintenance during the decline; t d For descent time; C ISS For MOSFET input capacitor; R ON The on-resistance for MOSFET drive; R G Internal resistor for MOSFET drive; V GS This refers to the voltage between the gate and drain of the MOSFET. V TH This refers to the MOSFET gate threshold voltage. V GP This refers to the MOSFET gate plateau voltage. R OFF This is the MOSFET drive turn-off voltage; Q GD This refers to the gate-drain charge of the MOSFET. P S-LOSS This represents the total loss of the MOSFET.

8. The high-boost coupled-inductor DC-DC converter using switched capacitors and coupled inductors as described in claim 1, characterized in that, The power losses of diodes D1, D2, D3, D4, D5 and D6 satisfy formulas (37)-(39): (37) (38) (39) In the formula, P Di-COND This refers to the diode's conduction loss. V FD This is the threshold voltage of the diode; I Di-AVE This represents the average diode current. I Di-rms This is the root mean square value of the diode current; r D This is the forward resistance of the diode; P SW-VDi This refers to diode switching losses; f S The switching frequency; V Di This is the diode voltage; I rr This is the reverse recovery current of the diode; t b This is the reverse recovery time of the diode; P D-LOSS This represents the total loss of the diode.

9. The high-boost coupled-inductor DC-DC converter using switched capacitors and coupled inductors as described in claim 1, characterized in that, The power loss of the capacitor satisfies formula (40), the power loss of the magnetic element satisfies formulas (41)-(43), and the total loss and efficiency of the converter satisfy formulas (44)-(45): (40) (41) (42) (43) (44) (45) In the formula, P C-LOSS This is for capacitor losses; r Cj This is the equivalent series resistance of the capacitor; I Cj-rms This is the root mean square value of the capacitor current; P L-LOSS For input inductor power loss; r L The equivalent resistance of the input inductor; r Lj-rms This is the root mean square value of the input inductance; P L-LOSS Copper loss of the input inductor; P N-LOSS This refers to the copper loss of the coupled inductor; r N1 The resistance is the primary winding resistance of the coupled inductor; r N2 For the secondary winding resistance of the coupled inductor; I N1-rms This represents the root mean square value of the primary winding current. I N2-rms This represents the root mean square value of the secondary winding current. P MC-LOSS This represents the total power loss of the coupled inductor. P LOSS Total power loss of the converter; P S-LOSS This represents the total power loss of the MOSFET. P D-LOSS This represents the total power loss of the diode. For the efficiency of the converter; P O This refers to the output power.

10. The high-boost coupled-inductor DC-DC converter employing switched capacitors and coupled inductors according to any one of claims 1 to 9, characterized in that, The converter is scalable. By setting m secondary windings and inserting switched capacitor circuits into them respectively, the extended voltage gain satisfies formula (46): (46) In the formula, G m For the extended voltage gain; i This refers to the serial number of the secondary winding; m This refers to the number of secondary windings; n i This represents the turns ratio between the secondary and primary windings for each extended winding.

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